JPH0746839A - Two-transistor type insulated switching power source - Google Patents

Abstract

PURPOSE:To achieve zero-volt switching and to reduce a switching loss by giving a main control pulse signal to a main switching element in accordance with a terminal voltage of a load and by giving an auxiliary control pulse signal to an auxiliary switching element before the main control pulse signal is given. CONSTITUTION:A series circuit of a reactor 27 for resonance, a diode 26 for preventing a reverse current and a MOSFET 25 is connected in parallel to a first MOSFET 3, and a series circuit of a reactor 32 for resonance, a diode 31 for preventing the reverse current and a MOSFET 30 is connected in parallel to a second MOSFET 6. A terminal voltage of a load 2 is detected by a control circuit 18 and an auxiliary control pulse signal is given to the MOSFETs 25 and 30 before a main control pulse signal is given to the first and second MOSFETs 3 and 6. Thereby the first and second MOSFETs 3 and 6 are turned ON at 0V and a switching loss at the time of turning ON can be reduced.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a two-stone isolated switching power supply, and more particularly to a two-stone isolated switching power supply capable of reducing switching loss.

[0002]

2. Description of the Related Art In recent years, there has been a strict demand for miniaturization of electronic equipment, and there is also a strong demand for miniaturization of a switching power supply which is a power supply device used for the electronic equipment. Generally, a switching power supply with a higher switching frequency is used to reduce the size of a switching power supply.However, as the switching frequency increases, the switching loss of the main switching element increases and the amount of heat generated by the main switching element increases. The large size has been an obstacle to miniaturization. For this reason, miniaturization of the switching power supply requires not only high frequency but also high efficiency. For example, a first main switching element and a transformer
Connecting the secondary winding and the second main switching element in series,
A rectifying / smoothing circuit is connected between the secondary winding of the transformer and the load to turn on the first and second main switching elements simultaneously.
An AC voltage is generated from the secondary winding of the transformer by turning it off, and the AC voltage is converted to a DC voltage by the rectifying and smoothing circuit and a DC output with a constant voltage level different from the voltage of the DC power supply is supplied to the load. Since the two-stone insulation type switching power supply can use a small main switching element having a low breakdown voltage, it has been widely used as a relatively small and inexpensive switching power supply.

[0003]

By the way, in the above-mentioned Futashi type insulation type switching power supply, the current waveform and the voltage waveform overlap each other at the on-conversion period and the off-conversion period of the first and second main switching elements, There is a drawback that switching loss based on this occurs in each main switching element. Further, since this switching loss becomes Joule heat, and the amount of heat generated by each main switching element increases, the size of the fins for heat radiation and the like becomes very large, which makes it extremely difficult to downsize the entire device.

Therefore, it is an object of the present invention to provide a two-stone insulated switching power supply that can reduce switching loss.

[0005]

A two-stone insulated switching power supply according to the present invention has a first main switching element, a primary winding of a transformer, and a second main switching element connected in series at both ends of a DC power supply. And a rectifying and smoothing circuit is connected between the secondary winding of the transformer and the load, and the first and second
By simultaneously turning on and off the main switching elements of the transformer, an AC voltage is generated from the secondary winding of the transformer, the AC voltage is converted into a DC voltage by the rectifying and smoothing circuit, and the voltage of the DC power supply is used. In a two-stone insulated switching power supply that supplies direct current outputs of different constant voltage levels to the load, a first resonance reactor, a first backflow prevention rectification element and a first resonance reactor are provided in parallel with the first main switching element. A first series circuit with the first auxiliary switching element is connected, and the second resonance reactor, the second backflow preventing rectifying element, and the second auxiliary switching element are connected in parallel with the second main switching element. A second series circuit is connected to apply a main control pulse signal to each control terminal of the first and second main switching elements according to a terminal voltage of the load, and the main control pulse signal is applied. Wherein prior to applying the pulse signal first
And an auxiliary control pulse signal is applied to each control terminal of the second auxiliary switching element. Further, the tertiary winding of the transformer may be inserted in series with the first series circuit, and the quaternary winding of the transformer may be inserted in series with the second series circuit.

[0006]

The auxiliary control pulse signal is applied to each control terminal of the first and second auxiliary switching elements before the main control pulse signal is applied to each control terminal of the first and second main switching elements. When the switching elements are turned on at the same time, the current flowing through each auxiliary switching element gradually rises from zero. When the current value becomes equal to the value obtained by converting the load current value into the primary winding of the transformer, the first and second
The voltage applied to the main switching element of is gradually decreased. Then, when the voltage becomes 0 V, a main control pulse signal is applied to each control terminal of the first and second main switching elements to turn on each main switching element at the same time. It is possible to reduce the switching loss. If the third and fourth windings of the transformer are inserted,
Also, the second auxiliary switching element is turned off at zero current at the same time, and the switching loss can be further reduced.

[0007]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a two-stone insulated switching power supply according to the present invention will be described below with reference to FIGS. 1, 2 and 6 and 7. The isolated switching power supply of this embodiment is shown in FIG.
As shown in FIG. 1, a first N-channel MOSFET 3 as a first main switching element, a primary winding 5 of a transformer 4 and a second N-channel MOSFET 6 as a second main switching element are provided at both ends of the DC power supply 1. Are connected in series. The first and second MOSFETs 3 and 6 are equivalently equivalent to the switching element bodies 12 and 15 and the built-in diodes 13 and 16 connected in antiparallel between the source and drain terminals of the switching element bodies 12 and 15, respectively.
And built-in capacitors 14 and 17 connected in parallel to the built-in diodes 13 and 16. The built-in diodes 13 and 16 and the built-in capacitors 14 and 17 are a parasitic diode and a parasitic capacitance between the source and drain terminals of the MOSFETs 3 and 6, respectively. The first and second M
OSFETs 3 and 6 having the same electrical characteristics are used. Secondary winding 8 of transformer 4 and output terminals 19 and 20
A rectifying / smoothing circuit 35 including a rectifying diode 7 and a freewheeling diode 9, a smoothing reactor 10 and a smoothing capacitor 11 is connected between and. Output terminals 19, 2
A load 2 is connected between 0s. First MOSFET
Between the source and drain terminals of 3, the first resonance reactor 27, the first backflow prevention diode 26 as the first backflow prevention rectifying element, and the third N channel as the first auxiliary switching element. A series circuit with the MOSFET 25 is connected. The second resonance reactor 3 is provided between the source and drain terminals of the second MOSFET 6.
2 and a second backflow prevention diode 31 as a second backflow prevention rectifying element and a fourth N-channel MOSFET 30 as a second auxiliary switching element are connected in series. Third and fourth MOSFETs 25, 30
Are equivalently equivalent to the switching element bodies 28, 3 respectively.
3 and built-in diodes 29 and 34, and has a parasitic capacitance between the source and drain terminals like the first and second MOSFETs 3 and 6, but the first and second MOSFEs.
Since it is used for a short period of time as compared with T3 and T6 and a MOSFET having a small parasitic capacitance is used, it is omitted here. First
The second and second resonance reactors 27 and 32 are provided to moderate the increase in current at the on-turning period of the switching element body portions 28 and 33 that form the third and fourth MOSFETs 25 and 30, respectively. . In addition, the first and second resonance reactors 27 and 32 and the third and fourth MOs.
SFETs 25 and 30 having the same electrical characteristics are used. Also, the output terminals 19 and 20 and the first to fourth M
A main control pulse signal is applied to each gate terminal of the first and second MOSFETs 3 and 6 between the gate terminals of the OSFETs 3, 6, 25 and 30 according to the voltage of the output terminals 19 and 20, and A control circuit 18 for applying an auxiliary control pulse signal is connected to the gate terminals of the third and fourth MOSFETs 25, 30 before applying the main control pulse signal.

The details of the control circuit 18 are as shown in FIG.
Voltage detection circuit 5 connected to output terminals 19 and 20 of the power supply
1, a reference voltage source 53, an error amplifier 54, a PWM pulse forming circuit 52 including a PWM (pulse width modulation) control circuit 55, a delay circuit 56, an AND gate 57, a monostable multivibrator 60, and a first ~ Fourth drive circuit 5
It is composed of 8, 59, 61 and 62. The voltage detecting circuit 51 is composed of a voltage dividing circuit, and this voltage dividing point, that is, the detection line is connected to the inverting input terminal of the error amplifier 54.
The error amplifier 54 has a non-inverting input terminal connected to the reference voltage source 53, and the reference voltage of the reference voltage source 53 and the voltage detection circuit 5
A signal corresponding to the difference in the detected voltage of 1 is output. The PWM control circuit 55 connected to the output terminal of the error amplifier 54
A triangular wave generator and a voltage comparator are included, and the voltage comparator generates a square wave with a constant period. In addition, PWM
In this embodiment, a PWM control IC (integrated circuit) is used as the control circuit 55. For example, a commercially available MB3759, μPC
494 etc. can be used. One input terminal of the AND gate 57 is directly connected to the PWM control circuit 55, and the other input terminal of the AND gate 57 is PWM via the delay circuit 56.
It is connected to the control circuit 55. The monostable multivibrator 60 is directly connected to the PWM control circuit 55. The output terminal of the AND gate 57 is connected to the first and second FET control lines 21 and 22 via the first and second drive circuits 58 and 59. The monostable multivibrator 60 is connected to the third and fourth FET control lines 23 and 24 via the third and fourth drive circuits 61 and 62. First to fourth FET control lines 21 to 24
Are connected to the respective gate terminals of the first to fourth MOSFETs 3, 6, 25, 30.

FIG. 7 shows voltage waveforms at points A, B and C in FIG.
Shown in (A), (B), and (C). From the PWM control circuit 55 to FIG.
A square wave pulse (PWM pulse) having a pulse width T _ON shown in (A) is repeatedly generated at a cycle T (point A in FIG. 6). When the output voltage of the power supply becomes higher than the reference value, the pulse width T _ON becomes narrower. On the contrary, when the output voltage of the power supply becomes lower than the reference value, the pulse width T _ON becomes wider. This is the same as the operation of a general PWM-controlled switching power supply. Since the pulse of FIG. 7A and the delay pulse of delay time T ₂ for this pulse are input to the AND gate 57, the AND gate 5
The main control pulse signal shown in FIG. 7B is output from the output terminal 7 (point B in FIG. 6). On the other hand, the pulse shown in FIG. 7A is also input to the monostable multivibrator 60, and the auxiliary control pulse signal shown in FIG. 7C is output from the monostable multivibrator 60 (point C in FIG. 6). This auxiliary control pulse signal is a pulse signal with a period T having a constant pulse width T ₁ . The main control pulse signal is the first and second drive circuits 5
The auxiliary control pulse signal is applied to the respective gate terminals of the first and second MOSFETs 3 and 6 via 8, 59, and the auxiliary control pulse signal is passed through the third and fourth drive circuits 61, 62.
It is applied to each gate terminal of the OSFETs 25 and 30. Therefore, before the control circuit 18 having the above configuration detects the terminal voltage of the load 2 and applies the main control pulse signal to the gate terminals of the first and second MOSFETs 3 and 6,
Also, an auxiliary control pulse signal can be applied to each gate terminal of the fourth MOSFETs 25 and 30.

In the above structure, the first to fourth MOSs
When the FETs 3, 6, 25 and 30 are all in the off state (see FIG.
Before t ₀ ), no current flows in the primary winding 5 and the secondary winding 8 of the transformer 4. Therefore, no voltage is applied to the primary winding 5 of the transformer 4, and the first and second MOSFE
The built-in capacitors 14 and 17 in T3 and 6 are charged to a voltage V _IN / 2 which is half the power supply voltage V _IN of the DC power supply 1. Further, no voltage is induced in the secondary winding 8 of the transformer 4, and the rectifying diode 7 in the rectifying / smoothing circuit 35 is in the off state. At this time, the load 2 is connected to the smooth reactor 1
0, smoothing capacitor 11 and load 2, freewheeling diode 9
The load current I _OUT flows through the path of.

As shown in FIG. 2B, at t ₀ , an auxiliary control pulse signal is applied from the control circuit 18 to each gate terminal of the third and fourth MOSFETs 25 and 30, and each switching element body 28, 33. When the auxiliary control pulse signal voltages V _G3 and V _G4 are changed from the low level to the high level, the switching element body portions 28 and 33 are turned on at the same time. At this time, from the DC power source 1, the switching element body 28, the first backflow prevention diode 26, the first resonance reactor 27, the primary winding 5 of the transformer 4, the second resonance reactor 32, and the second resonance reactor 32. A current starts to flow in the path of the backflow prevention diode 31 and the switching element body 33. A voltage is induced in the secondary winding 8 of the transformer 4, the rectifying diode 7 in the rectifying and smoothing circuit 35 is turned on, and a current flows, but the current flowing in the rectifying diode 7 is a return current while the load current is I _OUT or less. The diode 9 holds the ON state. Therefore, while the current flowing through the rectifier diode 7 is equal to or lower than the load current I _{OUT, the} primary windings 5 and 2 of the transformer 4 are
The voltage induced in the secondary winding 8 remains at approximately 0V. Therefore, the currents I _Q3 and I _Q4 flowing through the switching element bodies 28 and 33 in the third and fourth MOSFETs 25 and 30, respectively.
Is a slope (V _IN / (L ₂₇ + L ₃₂ )) related to the substantially power supply voltage V _IN and the inductances L ₂₇ and L ₃₂ of the first and second resonance reactors 27 and 32 as shown in FIG. 2 (F). ) With 0
Gradually increases from. Also, the third and fourth MOS
Each switching element body 28 in the FET 25, 30
The voltages V _Q3 and V _Q4 applied to 33 rapidly drop to 0 V as shown in FIG.

[0012] As shown in FIG. 2 (F), at t _1, the current I _Q3, I _Q4 flowing through the third and fourth switching elements body portion 28, 33 in the MOSFET25,30 load current 1 When the value converted to the next winding is reached, the built-in capacitors 14 and 1 in the first and second MOSFETs 3 and 6 are
The electric charge of 7 starts to be discharged, and the first resonance reactor 27
And the built-in capacitor 14 and the second resonance reactor 32.
And the built-in capacitor 17 resonate respectively. For this reason,
As shown in FIG. 2G, the built-in capacitor 14, the switching element body 28, the first backflow prevention diode 26, and the first resonance reactor 27, the built-in capacitor 17, and the second resonance reactor 27. 32, the second reverse current preventing diode 31, and the switching element body 33, respectively, a resonance current flows in the circuit (I _C14 , I _C17 ). Therefore, as shown in FIG. 2 (F), the third and fourth MOSFs are
Each switching element body 28, 3 in the ET 25, 30
The currents I _Q3 and I _Q4 flowing through 3 increase sinusoidally. In addition, as shown in FIG. 2C, the first and second MOSFETs are
The voltages V _Q1 and V _Q2 applied to the respective switching element bodies 12 and 15 in 3 and 6 are the first and second MOSFETs 3,
The electric charges of the built-in capacitors 14 and 17 in 6 are discharged to 0V.

At t ₂ the first and second MOSFETs
When all the electric charges accumulated in the built-in capacitors 14 and 17 in the MOSFETs 3 and 6 are discharged, the switching element body portions 12 in the first and second MOSFETs 3 and 6 are discharged as shown in FIG. The voltages V _Q1 and V _Q2 applied to 15 become 0V. In addition, as shown in FIGS. 2F and 2G, the built-in capacitor 1
4, switching element body 28, first backflow prevention diode 26, and first resonance reactor 27 circuit, built-in capacitor 17, and second resonance reactor 3
2. The resonance currents flowing in the circuit of the second backflow prevention diode 31 and the switching element body 33 reach their maximum values. At this time, the first and second resonance reactors 2
2 (G) and (H) due to the inertia of the current in each of the resonance reactors 27, 32 due to the resonance between the internal capacitors 14, 17 in the first and second MOSFETs 3, 6.
As shown in, the discharge current I of each built-in capacitor 14, 17
_C14 and I _C17 are commutated to the built-in diodes 13 and 16 in the first and second MOSFETs 3 and 6, respectively (I _D13 and I _D16 ). Therefore, the switching element body 28, the first backflow prevention diode 26, the first resonance reactor 27 and the circuit of the built-in diode 13, the second resonance reactor 32, the second backflow prevention diode 31, Currents I _D13 and I _D16 shown in FIG. 2 (H) respectively flow in the loops of the switching element body 33 and the built-in diode 16. In addition, as shown in FIG. 2 (F), the third and fourth M
Each switching element body 2 in the OSFETs 25 and 30
The currents I _Q3 and I _Q4 flowing through the capacitors 8 and 33 are substantially constant after that because the voltage drop across the first and second resonance reactors 27 and 32 is substantially 0V.

As shown in FIG. 2A, at t ₃ , a main control pulse signal is applied from the control circuit 18 to each gate terminal of the first and second MOSFETs ₃ and 6, and each switching element body 12, 15 is supplied. Main control pulse signal voltage V _G1 ,
When V _G2 changes from the low level to the high level, the switching element body portions 12 and 15 are turned on at the same time. At this time, since the voltages V _Q1 and V _G2 applied to the switching element bodies 12 and 15 are 0 V as shown in FIG. 2C, the switching element bodies 12 and 15 are turned on at 0 V. Therefore, each switching element body 12, 15
Zero voltage switching with almost no switching loss can be realized during the on-conversion period. At this point in time, since the switching element bodies 28 and 33 are in the ON state, all the current flowing through the primary winding 5 is caused by the inertia of the current flowing through the first and second resonance reactors 27 and 32. 2 (E), currents I _Q1 and I _Q are supplied to the switching element main bodies 12 and 15 as shown in FIG.
_Q2 does not flow.

As shown in FIG. 2B, at t ₄ , the third
And the auxiliary control pulse signal voltages V _G3 and V 3 of the switching element bodies 28 and 33 of the fourth MOSFETs 25 and 30, respectively.
_{When the} level of _G4 changes from the high level to the low level, the switching element body portions 28 and 33 simultaneously turn off. At this time, as shown in FIGS. 2 (E) and 2 (F), the primary winding 5 of the transformer 4 flowing through the switching element main bodies 28 and 33.
Current (I _Q3 , I _Q4 ) of the first and second MOSFETs
The commutation is carried out to the switching element main body parts 12 and 15 in 3 and 6 (I _Q1 and I _Q2 ).

As shown in FIG. 2A, at t ₅ , the first
When the main control pulse signal voltages V _G1 and V _{G2 of the} switching element bodies 12 and 15 of the second MOSFETs 3 and 6 change from high level to low level, the switching element bodies 12 and 15 are turned off at the same time. At this time, due to the inertia of the current in the primary winding 5 of the transformer 4, a charging current flows through the built-in capacitors 14 and 17 in the first and second MOSFETs 3 and 6 as shown in FIG. The built-in capacitors 14 and 17 are charged, and the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 gradually rise from 0 V as shown in FIG. 2 (C).

As shown in FIG. 2C, at t ₆ , the first
And the built-in capacitors 1 of the second MOSFETs 3 and 6
The voltage applied to 4, 17, that is, the first and second MOSFs
When the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 of the ETs 3 and 6 reach V _IN / 2 (V _IN : the power supply voltage of the DC power supply 1), the primary winding 5 of the transformer 4
The voltage induced in the secondary winding 8 is about 0V. At the same time, the rectifying diode 7 in the rectifying / smoothing circuit 35 is turned off, the exciting inductance of the transformer 4 and the built-in capacitors 14 and 17 resonate, and the respective voltages V _Q1 and V _Q2 further rise sinusoidally. . Then, each voltage V _Q1 , V
_When the maximum value of _Q2 reaches the maximum value of the resonance voltage and the voltage V _IN / 2, the respective voltages V _Q1 and V _Q2 drop sinusoidally, and at t ₇ , the respective voltages V _Q1 and V _{Q Q2} is equal to V _IN / 2, respectively. At this time, no current flows in the primary winding 5 and the secondary winding 8 of the transformer 4 and no voltage is induced. The load current I _OUT flows through the load 2 through the smoothing reactor 10, the smoothing capacitor 9, the load 2, and the free wheeling diode 9.

As described above, in this embodiment, the first and second
Switching element body 1 of MOSFETs 3 and 6
Turn on 2 and 15 at 0V at the same time and
Since the rising of the current waveform of each switching element body 28, 33 of the fourth MOSFET 25, 30 is gentle, each switching element body 12, 15, 28,
It is possible to reduce the switching loss during the on-conversion period of 33 (at the time of turn-on).

Next, another embodiment of the two-stone insulated switching power supply according to the present invention will be described with reference to FIGS. 3 and 4. However, in FIG. 3, the same parts as those in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. The details of the control circuit 18 of FIG. 3 are the same as those of FIGS. 6 and 7 shown in the embodiment of FIG. In the circuit of the embodiment shown in FIG. 3, a tertiary winding 37 that is coupled to the primary winding 5 and the secondary winding 8 of the transformer 4 in reverse polarity is inserted between the third MOSFET 25 and one end of the DC power supply 1. And the fourth MOSFET 30
And the other end of the DC power source 1, a quaternary winding 38 that is coupled to the primary winding 5 and the secondary winding 8 of the transformer 4 in reverse polarity is inserted. That is, the tertiary windings 37 and 4 of the transformer 4
The secondary winding 38 has a direction opposite to the current flowing through the first and second resonance reactors 27 and 32 when the switching element main body portions 12 and 15 in the first and second MOSFETs 3 and 6 are in the ON state. When the switching element bodies 12 and 15 are wound so as to induce a voltage and turned on, the currents flowing in the switching element bodies 28 and 33 of the third and fourth MOSFETs 25 and 30 are gradually reduced. , For turning off the switching element bodies 28 and 33 at zero current. Incidentally, N ₃ between the number of turns N ₃ of the third winding 37 and the winding number N ₄ of 4 winding 38
There is a relationship of = N ₄ .

In the above structure, the first to fourth MOSs
When the FETs 3, 6, 25 and 30 are all in the off state (see FIG.
Before t ₀ ), the primary to quaternary windings 5, 8 of the transformer 4 are
No current flows through 37 and 38. Therefore, transformer 4
No voltage is applied to the primary winding 5 of the first and second M
The built-in capacitors 14 and 17 in the OSFETs 3 and 6 are charged to a voltage V _IN / 2 which is half the power supply voltage V _IN of the DC power supply 1. Further, no voltage is induced in the secondary to quaternary windings 8, 37, 38 of the transformer 4, and the rectifying and smoothing circuit 3
The rectifying diode 7 in 5 is off. At this time,
The load 2 includes a smoothing reactor 10 and a smoothing capacitor 11
And the load current I _OUT through the path of the load 2 and the return diode
Flows.

As shown in FIG. 4B, at t ₀ , an auxiliary control pulse signal is applied from the control circuit 18 to the gate terminals of the third and fourth MOSFETs 25 and 30, and the switching element body portions 28 and 33 are provided. When the auxiliary control pulse signal voltages V _G3 and V _G4 are changed from the low level to the high level, the switching element body portions 28 and 33 are turned on at the same time. At this time, the DC winding 1 to the tertiary winding 3 of the transformer 4
7, switching element body 28, first backflow prevention diode 26, first resonance reactor 27, primary winding 5 of transformer 4, second resonance reactor 32, second backflow prevention diode 31. , Switching element body 3
Current begins to flow in the path of the third winding and the fourth winding 38 of the transformer 4. A voltage is induced in the secondary winding 8 of the transformer 4, the rectifying diode 7 in the rectifying and smoothing circuit 35 is turned on, and a current flows, but the current flowing in the rectifying diode 7 is a return current while the load current is I _OUT or less. The diode 9 holds the ON state. Therefore, while the current flowing through the rectifying diode 7 is less than or equal to the load current I _{OUT, the} voltage induced in the primary to quaternary windings 5, 8, 37, 38 of the transformer 4 remains at approximately 0V. Therefore, the third and fourth MOSFETs 25, 3
The currents I _Q3 and I _Q4 flowing through the switching element bodies 28 and 33 in 0 are substantially the power supply voltage V _{IN as} shown in FIG.
And the inclination (V _IN / (L ₂₇ + L 2) related to the inductances L ₂₇ and L ₃₂ of the first and second resonance reactors ₂₇ and _32.
32 )) gradually increases from 0. Further, the voltages V _Q3 and V _Q4 applied to the respective switching element bodies 28 and 33 in the third and fourth MOSFETs 25 and 30 rapidly drop to 0 V as shown in FIG. 4 (D).

As shown in FIG. 4 (F), at t ₁ , the currents I _Q3 and I _Q4 flowing through the switching element bodies 28 and 33 in the third and fourth MOSFETs 25 and 30 have a load current value of 1 When the value converted to the next winding is reached, the built-in capacitors 14 and 1 in the first and second MOSFETs 3 and 6 are
The electric charge of 7 starts to be discharged, and the first resonance reactor 27
And the built-in capacitor 14 and the second resonance reactor 32.
And the built-in capacitor 17 resonate respectively. For this reason,
As shown in FIG. 4G, the built-in capacitor 14 and the transformer 4
Of the third winding 37, the switching element body 28, the first backflow prevention diode 26 and the first resonance reactor 27, the built-in capacitor 17, the second resonance reactor 32, and the second backflow. Resonant currents flow in the loops of the prevention diode 31, the switching element body 33, and the quaternary winding 38 of the transformer 4 ( _IC14 , _IC17 ). Therefore, as shown in FIG. 4 (F), the third and fourth MOS
Each switching element body 28 in the FET 25, 30
The currents I _Q3 and I _Q4 flowing through 33 increase sinusoidally.
In addition, as shown in FIG. 4C, the first and second MOSFE
The voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 in T3 and 6 are the first and second MOSFETs.
The electric charges of the built-in capacitors 14 and 17 in 3 and 6 are discharged, so that the voltage drops to 0V.

At t ₂ the first and second MOSFETs
When all the electric charges accumulated in the built-in capacitors 14 and 17 in the circuits 3 and 6 are discharged, as shown in FIG. 4C, the switching element main body 12 in the first and second MOSFETs 3 and 6, The voltages V _Q1 and V _Q2 applied to 15 become 0V. In addition, as shown in FIGS. 4 (F) and (G), the built-in capacitor 1
4, the tertiary winding 37 of the transformer 4, the switching element body 28, the first backflow prevention diode 26 and the first resonance reactor 27, the built-in capacitor 17, the second resonance reactor 32, The resonance currents flowing through the second backflow preventing diode 31, the switching element body 33, and the circuit of the quaternary winding 38 of the transformer 4 each reach the maximum value. At this time, the first and second resonance reactors 2
4 (G) and (H) due to the inertia of the current of the resonance reactors 27, 32 due to the resonance of the built-in capacitors 14, 17 in the first and second MOSFETs 3, 6 with each other.
As shown in, the discharge current I of each built-in capacitor 14, 17
_C14 and I _C17 are commutated to the built-in diodes 13 and 16 in the first and second MOSFETs _{3 and 6} , respectively (I _{D1 3} and I _D16 ). Therefore, the tertiary winding 37 of the transformer 4, the switching element body 28, the first backflow prevention diode 26, the first resonance reactor 27 and the circuit of the built-in diode 13, and the second resonance reactor 3 are provided.
2, the second backflow prevention diode 31, the switching element body 33, the quaternary winding 38 of the transformer 4 and the circuit of the built-in diode 16 current I _D13 , I shown in FIG. 4 (H) respectively.
_D16 flows. Also, the first and second MOSFETs 3,
When the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 in 6 become 0 V, the primary winding 5 of the transformer 4
The voltage applied to the DC power supply 1 becomes equal to the power supply voltage V _IN of the DC power supply 1, and the secondary winding 8 has a voltage V _N2 = (N ₂ / N ₁ ) · V _IN (N ₂ :
The number of turns of the secondary winding 8) is induced. Further, in the tertiary winding 37 and the fourth winding 38 of the transformer 4, the currents I _D13 ,
A voltage V _N3 =-(N ₃ / having a polarity opposite to that of the flowing direction of I _D16
N ₁ ) · V _IN , V _N4 = − (N ₄ / N ₁ ) · V _IN (N ₃ : the number of turns of the tertiary winding 37, N ₄ : the number of turns of the quaternary winding 38, where N ₃ =
N ₄ ) is induced. Therefore, the current flowing through the first and second resonant reactor 27 and 32, i.e. the current I _Q3 flowing through each switching device body unit 28, 33, I _Q4 and a current flowing in each of the built-in diode 13, 16 I _D13, I
_As shown in FIGS. 4 (F) and (H), _{D16 is-} (N ₃ / N ₁ ). (V _IN / L ₂₇ ),-(N ₄ /
N ₁ ). (V _IN / L ₃₂ ) (However, N ₃ = N ₄ , L ₂₇ = L ₃₂ ) Gradient gradually decreases.

As shown in FIG. 4A, at t ₃ , the main control pulse signal is applied from the control circuit 18 to the gate terminals of the first and second MOSFETs _{3 and 6} , and the switching element body portions 12 and 15 are supplied. Main control pulse signal voltage V _G1 ,
When V _G2 changes from the low level to the high level, the switching element body portions 12 and 15 are turned on at the same time. At this time, since the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 are 0 V as shown in FIG. 4C, the switching element bodies 12 and 15 are turned on at 0 V. Therefore, each switching element body 12, 15
Zero voltage switching with almost no switching loss can be realized during the on-conversion period. At this point, the third and fourth switching elements body portion 28, 33 in the MOSFET25,30 is because in the ON state, subsequently flows current I _Q3, I _Q4 to the switching device body unit 28 and 33, FIG. As shown in FIG. 4 (F),-(N ₃ / N ₁ ). (V _IN /
L ₂₇ ),-(N ₄ / N ₁ ) ・ (V _IN / L ₃₂ ) (where N ₃ =
With the slope of N ₄ , L ₂₇ = L ₃₂ ), it gradually decreases.

At t ₄ , the first and second MOSFE
When the currents I _D13 and I _D16 flowing through the respective built-in diodes 13 and 16 in T3 and T6 become 0 as shown in FIG.
And currents I _Q3 and I _Q4 flowing through the switching element bodies 28 and 33 in the fourth MOSFETs 25 and 30 are shown in FIG.
As shown in (F), the load current value reaches the value converted to the primary winding. At this time, as shown in FIG.
Switching elements 12 and 1 in the MOSFETs 3 and 6 of
The currents I _Q1 and I _Q2 start to flow in 5 and gradually increase.
At the same time, the current I flowing through each switching element body 28, 33 in the third and fourth MOSFETs 25, 30.
_Q3, I _Q4, as shown in FIG. _{4 (F) - (N 3} / N 1) · (V IN
/ L ₂₇ ),-(N ₄ / N ₁ ) ・ (V _IN / L ₃₂ ) (where N ₃ =
N ₄ and L ₂₇ = L ₃₂ ) further decreases with the inclination.

As shown in FIG. 4 (F), at t ₅ ,
The currents I _Q3 and I _Q4 flowing through the switching element bodies 28 and 33 in the fourth MOSFETs 25 and 30 are zero. All the currents of the primary winding 5 of the transformer 4 flow through the respective switching element bodies 12 and 15 in the first and second MOSFETs 3 and 6 (I _Q1 ,
I _Q2 ). At this time, as shown in FIG.
The auxiliary control pulse signal voltages V _G3 and V _{G4 of the} switching element bodies 28 and 33 in the MOSFETs 25 and 30 change from high level to low level, and the switching element bodies 28 and 33 are simultaneously turned off. Therefore, zero current switching with less switching loss can be realized even in the off-turning period of the switching element main body portions 28 and 33 in the third and fourth MOSFETs 25 and 30. Further, since the switching element bodies 28 and 33 are turned off at the same time with zero current, the surge component due to the inductance of the wiring of the circuit and the intermittent current flowing through the first and second resonance reactors 27 and 32 does not occur.

As shown in FIG. 4A, at t ₆ , the first
When the main control pulse signal voltages V _G1 and V _{G2 of the} switching element bodies 12 and 15 in the second MOSFETs 3 and 6 change from high level to low level, the switching element bodies 12 and 15 are simultaneously turned off. At this time, as shown in FIG. 4 (G), the first and second MOSFETs 3 and 6 are
A charging current flows through the internal capacitors 14 and 17 in the internal capacitors 14, 17 to charge the internal capacitors 14 and 17 and the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 are shown in FIG.
As shown in (C), it gradually rises from 0V.

As shown in FIG. 4C, at t ₇ , the first
And the built-in capacitors 1 of the second MOSFETs 3 and 6
The voltage applied to 4, 17, that is, the first and second MOSFs
When the voltages V _Q1 and V _Q2 applied to the switching element bodies 12 and 15 of the ETs 3 and 6 reach V _IN / 2 (V _IN : the power supply voltage of the DC power supply 1), the primary winding 5 of the transformer 4
The voltage induced in the secondary winding 8 is about 0V. At the same time, the rectifying diode 7 in the rectifying / smoothing circuit 35 is turned off, the exciting inductance of the transformer 4 and the built-in capacitors 14 and 17 resonate, and the respective voltages V _Q1 and V _Q2 further rise sinusoidally. . Then, each voltage V _Q1 , V
_When the maximum value of _Q2 reaches the maximum value of the resonance voltage and the voltage V _IN / 2, the respective voltages V _Q1 and V _Q2 drop sinusoidally, and at t ₈ , the respective voltages V _Q1 and V _Q2 is equal to V _IN / 2, respectively. At this time, the primary of the transformer 4 ~
No current flows in the quaternary windings 5, 8, 37, 38 and no voltage is induced. Further, the load 2 includes a smoothing reactor 10, a smoothing capacitor 9, a load 2, and a free wheeling diode 9.
The load current I _OUT flows through the path of.

As described above, also in the embodiment shown in FIG. 3, the same effect as the embodiment shown in FIG. 1 can be obtained with respect to the switching loss. Furthermore, in the embodiment shown in FIG. 3, the third and fourth
Since zero current switching with little switching loss can be realized even during the off-turning period (turn-off time) of each switching element body 28, 33 in each MOSFET 25, 30, the circuit wiring inductance and the first and second resonance No surge component is generated due to the interruption of the current flowing through the reactors 27 and 32. Therefore, the embodiment shown in FIG. 3 is more effective than the embodiment shown in FIG.

The embodiment of the present invention is not limited to the above-mentioned embodiments, and various modifications can be made. For example, the following (a)
(e) is a part of the modified example. (A) First to fourth MOSFETs 3, 6, 25, 30
Each of the built-in diodes 13, 16, 29, and 34 therein can be an independent diode instead of the built-in diode. (B) Independent capacitors can be connected to the built-in capacitors 14 and 17 in the first and second MOSFETs 3 and 6 without using the parasitic capacitance of the MOSFETs. (C) As the main switching element and the auxiliary switching element, a bipolar transistor, a thyristor or the like can be used instead of the MOSFET. When an element such as a bipolar transistor or a thyristor that does not include a diode of reverse polarity is used, the backflow preventing diodes 26 and 31 may not be inserted. (D) The circuit of the embodiment shown in FIG. 1 may be modified into the circuit shown in FIG. In addition, the diode 36 shown in FIG.
Does not have to be inserted. (E) The resonance reactors 27 and 32 may use the leakage inductance of the transformer 4.

[0031]

As described above, according to the present invention, zero voltage switching of the first and second main switching elements can be easily achieved, so that the voltage waveform and the current waveform of each main switching element overlap. Can be reduced to reduce the power loss at the turn-on time of each main switching element, that is, the switching loss at the time of turn-on of each main switching element. Therefore, the heat generation amount of each main switching element can be reduced to reduce the size of the heat dissipation fins, etc., and high-frequency switching is possible, and a small-sized two-stone insulated switching power supply can be realized.

[Brief description of drawings]

FIG. 1 is an electric circuit diagram of a two-stone insulated switching power supply showing an embodiment of the present invention.

FIG. 2 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

FIG. 3 is an electric circuit diagram of a two-stone insulation type switching power supply showing another embodiment of the present invention.

FIG. 4 is a waveform diagram showing the voltage and current of each part of the circuit of FIG.

5 is an electric circuit diagram showing a modified example of the circuit of FIG.

FIG. 6 is a block diagram showing details of the control circuit of FIGS. 1, 3 and 5.

FIG. 7 is a waveform diagram showing the voltage of each part of the circuit of FIG.

[Explanation of symbols]

1. ． ． DC power supply, 2. ． ． Load 3, 6, 25, 3
0. ． ． First to fourth N-channel MOSFETs,
4. ． ． Transformer 5, 8, 37, 38. ． ． 1st to 4th
Next winding, 7. ． ． Rectifier diode, 9. ． ． Free wheeling diode, 10. ． ． Smoothing reactor, 11. ． ． Smoothing capacitor, 18. ． ． Control circuit, 19, 20. ． ． Output terminals, 26, 31. ． ． First and second backflow preventing diodes 27, 32. ． ． The first and second resonance reactors,
35. ． ． Rectifying and smoothing circuit

Claims (2)

[Claims] 1. A first main switching element, a primary winding of a transformer, and a second main switching element are connected in series at both ends of a DC power supply, and between the secondary winding of the transformer and a load. An AC voltage is generated from the secondary winding of the transformer by connecting a rectifying / smoothing circuit and controlling ON / OFF of the first and second main switching elements simultaneously, and the AC voltage is supplied to the rectifying / smoothing circuit. In the two-stone insulation type switching power supply for converting into a direct current voltage and supplying a direct current output of a constant voltage level different from the voltage of the direct current power source to the load, a first resonance in parallel with the first main switching element. Connecting a first series circuit of the first reactor and the first backflow preventing rectifying element and the first auxiliary switching element, and connecting the second resonance reactor and the second resonance reactor in parallel with the second main switching element. A second series circuit of a flow prevention rectifying element and a second auxiliary switching element is connected, and a main control pulse signal is supplied to each control terminal of the first and second main switching elements according to the terminal voltage of the load. And the auxiliary control pulse signal is applied to each control terminal of the first and second auxiliary switching elements before applying the main control pulse signal. Type switching power supply. 2. The "claim 1" wherein the tertiary winding of the transformer is inserted in series with the first series circuit, and the quaternary winding of the transformer is inserted in series with the second series circuit. The two-stone insulated switching power supply described.