JPH0670571A - Controller for inverter device - Google Patents

Abstract

PURPOSE:To make high-responce and high-accuracy torque control possible by a method wherein the output of current control, which controls an exciting component, is divided into the voltage command of the exciting component and the voltage command of a torque component to add them, in an AC induction motor, in which flux and current are controlled separately. CONSTITUTION:An AC induction motor, in which a DC power collected from a line through a pantograph 11 is converted into an AC voltage having a variable voltage and variable frequency by a PWM inverter 1 and is impressed on an AC motor 2 for driving an electric car, is equipped with a current controller 8, inputting an exciting current command Id*, an exciting current Id, a torque current Iq and a primary angular frequency command omega1*. The exciting current component of a rotary feild coordinate system is divided into a (d) axis voltage command and a (q) axis voltage command from a voltage command operator 5 in accordance with the primary angular frequency command omega1* or the rotating speed of the motor 2 to add them. In this case, when the output frequency of the inverter 1 is lower than a predetermined value, the exciting current component is added to the (d) axis voltage command but when the same is lower than the predetermined value, the exciting current component is added to the (q) axis voltage command.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an inverter device, and more particularly to a control device for an inverter device which is suitable for independently controlling magnetic flux and current of an AC induction motor with high accuracy. Regarding

[0002]

2. Description of the Related Art As a method of controlling torque of an AC motor with high response and high accuracy, vector control has been proposed which independently controls a magnetic flux and an electric current of the AC motor. In the vector control, since the magnetic flux and the current are controlled independently, it is necessary to control the magnetic flux of the AC motor to be constant and it is necessary to know the equivalent circuit constant of the AC motor in advance.

Particularly, in the AC motor, a change in the secondary resistance due to a temperature rise on the secondary side causes a change in the slip frequency, which causes an interference between the magnetic flux of the motor and the current. Compensation law is important.

As a conventional technique for performing such control, for example, a technique described in Japanese Patent Laid-Open No. 59-156184 is known.

This prior art is to detect the output voltage of the inverter controlling the AC motor, correct the set value of the secondary resistance based on this voltage detection signal, and compensate the slip frequency. .

Regarding the change in magnetic flux, it has also been proposed to detect the output voltage of the inverter and correct the exciting current command based on this voltage detection signal.

[0007]

However, the above-mentioned prior art is provided with the current control means for controlling the magnetic flux and the current of the AC motor. This current control is performed by changing the rotational speed of the AC motor. It functions only in a part of the range, does not fully function over the entire range of rotation speed, and has a problem of unstable control.

The object of the present invention is to solve the above-mentioned problems of the prior art and to control the magnetic flux and current of an AC motor with high accuracy by current control over the entire range of change in the rotational speed of the AC motor. An object of the present invention is to provide a control device for an inverter device that can perform the operation.

[0009]

According to the present invention, the above object is achieved in vector control in which the primary current of an AC motor is separated into an excitation component (d-axis) and a torque component (q-axis) for current control. If the output frequency of the inverter is less than the specified value,
This is achieved by adding the output of the excitation current control to the voltage command of the d-axis and adding the output of the excitation current control to the voltage command of the q-axis when the output frequency of the inverter is equal to or higher than a predetermined value. Further, the above-mentioned object can be achieved by controlling the current in combination with frequency control for performing closed loop control of the torque component (q axis) or the magnitude of the primary current.

[0010]

In general, the voltage-current characteristic of an AC motor changes depending on its rotation speed. When the voltage-current characteristics of the AC motor are represented by the excitation component (d axis) and the torque current component (q axis) of the rotating magnetic field coordinate system, the inventors of the present invention can operate in the low operating speed range of the AC motor. It was discovered that the exciting current that creates a magnetic flux flows by the voltage of the d-axis, whereas the exciting current flows by the voltage of the q-axis in an operating range where the rotation speed is high.

Therefore, the present invention is based on the above findings.
Focusing on the voltage component that controls the magnetic flux, the output of the current control that controls the exciting current is continuously output to the d-axis voltage command and the q-axis voltage command according to the rotation speed of the AC motor or the magnitude of the inverter frequency. It was switched or added and distributed.

As a result, according to the present invention, the exciting current of the AC electric motor can be controlled so that it matches the exciting current command and the magnetic flux also matches the command value and becomes constant. Further, when the magnetic flux is controlled to be constant, the torque current flows in proportion to the slip frequency, so frequency control is performed by closed-loop control of the torque component (q-axis) or the magnitude of the primary current. It is possible to compensate for the change in the slip frequency due to the change in the secondary resistance due to the temperature rise on the secondary side. Accordingly, the present invention can control the torque current of the AC electric motor so as to match the torque current command.

[0013]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a control device for an inverter device according to the present invention will be described in detail below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of a first embodiment of the present invention, FIG. 2 is a block diagram showing the configuration of a voltage command calculator, and FIG. 3 is a block diagram showing the configuration of a current controller. FIG. 4 is a diagram for explaining the characteristics of the output frequency of the inverter and the voltage component that contributes to the magnetic flux. 1 to 3, 1
Is a PWM inverter, 2 is an AC motor, 3 is a PWM signal calculator, 4 and 9 are coordinate calculators, 5 is a voltage command calculator, 6
Is a control command calculator, 7 is a speed controller, 8 is a current controller,
10 is a slip frequency command calculator, 16 is a speed detector, 5
00, 502, 504, and 507 are constant units that give a gain, 801, 805, and 811 are function generators, 804 and 8
Reference numerals 07 and 810 are current regulators.

In the first embodiment of the present invention shown in FIG. 1, the pantograph 1 supplies DC power to the inverter 1.
1, an electric train driving inverter that is performed from an overhead wire via a DC reactor 12 and a filter capacitor 13, but the present invention is not limited to this, and any inverter that controls an AC motor can be used. It can be applied to anything used for any purpose.

In one embodiment of the invention shown in FIG. 1, P
The WM inverter 1 converts a DC voltage into an AC voltage having a variable voltage and a variable frequency, and supplies a three-phase AC voltage to the AC motor 2, and is composed of a self-extinguishing element such as a transistor, an IGBT, and a GTO. . And this PWM
The inverter 1 is controlled by ON / OFF pulses from the PWM signal calculator 3 which is created by comparing the output voltage commands Vu *, Vv *, Vw * with the carrier signal.

The coordinate converter 4, an exciting current component, the output voltage command Vd ** for the torque current component, Vq ** coordinate reference signal sin .omega ₁ * t, a and cosω ₁ * t, 3-phase AC voltage command Vu * , Vv *, Vw * are calculated, and these commands are given to the PWM signal calculator 3. The speed signal ωr from the speed detector 16 is input to the adders 17 and 18. Also,
The current detectors 15 u to 15 w detect the output currents iu, iv, iw of the PWM inverter 1 and give the detected values to the coordinate converter 9. The coordinate converter 9 uses the coordinate reference signal sin.
omega ₁ * t, based on cos .omega ₁ * t, the detected inverter 1 output currents iu, converts iv, exciting current component Id of the iw rotating field coordinate system, the torque current component Iq, these current components Id , Iq are input to the current controller 8.

The slip frequency command calculator 10 calculates the slip frequency ωs * from the exciting current command Id * and the torque current command Iq * of the rotating magnetic field coordinate system from the control command generator 6,
Input to the adder 18. The adder 18 adds the velocity signal ωr and the slip frequency command ωs * to add the primary angular frequency command ω _0.
* Is calculated and output to the adder 24. The adder 24 adds the given primary angular frequency command ω ₀ * and the deviation Δω ₁ * from the current controller 8 to calculate the primary angular frequency command ω ₁ *, and the result is an integrator. 20, the control command generator 6, the output voltage command calculator 5, and the current controller 8.

The integrator 20 calculates the coordinate reference signal ω ₁ * t from the primary angular frequency command ω ₁ * and outputs it to the oscillator 21.
The control command generator 6 uses the torque command T from the speed controller 7.
*, The voltage of the filter capacitor 13 of the DC power supply circuit of the inverter and the primary angular frequency command ω ₁ * from the adder 24
With this, the exciting current component command Id * and the torque current component command Iq * of the rotating magnetic field coordinate system are calculated, and the slip frequency command calculator 10, the output voltage command calculator 5, and the current controller 8 are calculated.
Output to.

The output voltage command calculator 5 includes a primary angular frequency command ω ₁ * and an exciting current component command I from the control command generator 6.
d *, the torque current component command Iq *, and the magnetic flux command φ _d * for the AC motor 2 given as a constant, the excitation output voltage command Vd * and the torque output voltage command Vq * of the inverter 1 are calculated, and the adder is calculated. It outputs to 22 and 23. The adders 22 and 23 add the output voltage commands Vd * and Vq * from the output voltage command calculator 5 and the deviations ΔVd * and ΔVq * of the output voltage command from the current controller 8, respectively, to generate an exciting current component and a torque. Output voltage commands Vd **, Vq for current components
Output as ** to the coordinate converter 4.

The adder 17 calculates the deviation between the speed command ωr * from the speed commander 19 and the speed signal ωr from the speed detector 16, and outputs the result to the speed adjuster 7. The speed controller 7 calculates the torque command T * based on the speed deviation from the adder 17, and outputs it to the control command generator 6.

The output voltage command calculator 5 is, as shown in FIG. 2, a constant device 500, 5 for giving a gain to an input signal.
02, 504, 507 and adders 501, 506, 50
8 and multipliers 505 and 508.

As shown in FIG. 2, the output voltage command calculator 5
Is input with the excitation current command Id *, the torque current command Iq *, the primary angular frequency command ω ₁ *, and the magnetic flux command φ _d *, and outputs the voltage commands Vd * and Vq * of the rotary excitation coordinate system from them. To do. The excitation current command Id * is input to the constant devices 500 and 504, and the torque current command Iq * is input to the constant devices 502 and 50.
7 is input and a predetermined gain is given. Constant number 50
The output of 2 is added to the multiplier 503, multiplied by the primary angular frequency command ω ₁ *, and output to the adder 501. Also,
The output of the constant device 504 is the magnetic flux command φd * at the adder 506.
After being added, the primary angular frequency command ω ₁ *
It is multiplied by and output to the adder 508.

The adder 501 adds the output of the constant unit 500 and the output of the multiplier 503 to generate an exciting voltage command Vd * for the rotating magnetic field coordinate system, and outputs it to the adder 22. Further, the adder 508 adds the output of the constant unit 506 and the output of the multiplier 505 to add the torque voltage command Vq * of the rotating magnetic field coordinate system.
Is generated and output to the adder 23.

As shown in FIG. 3, the current controller 8 includes function generators 801, 805, 811 and a current regulator 804,
807 and 810, multipliers 803, 806 and 809,
It is configured to include adders 802 and 808. The function generators 801, 805, 811 are configured so that the AC motor 2 can be controlled in both forward and reverse directions.

The current controller 8 receives the exciting current command Id *, the exciting current Id, the torque current command Iq *, the torque current Iq, and the primary angular frequency command ω ₁ * as inputs, and rotates based on these. Deviation ΔV of excitation voltage command in magnetic field coordinate system
d *, the deviation ΔVq * of the torque voltage command, and the deviation Δω ₁ * of the primary angular frequency command are output.

The primary angular frequency command ω ₁ * is calculated by the function generator 80.
1, 805, 811, the exciting current command Id *,
The exciting current Id is input to the adder 802. Adder 8
02 calculates the deviation between the exciting current command Id * and the exciting current Id and outputs it to the multipliers 803 and 806. Multiplier 803
Outputs the output signal of the function generator 801 and the deviation of the exciting current to the current regulator 804. Current regulator 8
04 calculates the deviation ΔVd * of the excitation voltage command of the rotating magnetic field coordinate system so as to make the output of the multiplier 803 zero, and outputs it to the adder 22. The multiplier 806 is a function generator 80.
The output signal of No. 5 and the deviation of the exciting current are multiplied and output to the current controller 807. The current regulator 807 is a multiplier 80.
The deviation ΔVq * of the torque voltage command in the rotating magnetic field coordinate system is calculated so that the output of 6 becomes zero, and the difference is output to the adder 23.

Torque current command Iq *, torque current Iq
Is input to the adder 808 and the torque current command Iq * is input.
And the torque current Iq are calculated, and the difference is input to the multiplier 809. The multiplier 809 is the function generator 8
The output signal of 11 and the deviation of the torque current are multiplied and output to the current regulator 810. The current controller 810 calculates the deviation Δω ₁ * of the primary angular frequency command so that the output of the multiplier 809 becomes zero and outputs it to the adder 24.

Next, referring to FIG. 4 for explaining the characteristics of the output frequency of the inverter and the voltage component contributing to the magnetic flux, the method of controlling the inverter according to the embodiment of the present invention configured as described above. Will be explained.

The configuration of one embodiment of the present invention shown in FIG. 1 is for controlling the magnetic flux of the AC motor 2 and the torque current component orthogonal thereto, and is used for driving a train. Therefore, the DC power supply of the inverter 1 is supplied from the overhead wire via the pantograph 11, the DC reactor 12, and the filter capacitor 13, but the voltage of this overhead wire is
Due to the power running and regeneration operations of other trains, the voltage fluctuations are large, and in the illustrated embodiment, maintaining the magnetic flux of the AC motor 2 at a predetermined value is extremely important for torque control.

In order to independently control the magnetic flux of the AC motor and the torque current component orthogonal to the magnetic flux, the magnitude of the magnetic flux is maintained at a predetermined value and an appropriate slip frequency is given to control the magnitude of the torque current. Need to control.

Therefore, in the first embodiment of the present invention, in order to control the magnitude of the magnetic flux of the AC electric motor, the relationship between the voltage component of the rotating magnetic field coordinate system and the magnetic flux is obtained with respect to the rotational frequency of the AC electric motor. The voltage component that is more effective in controlling the magnetic flux is controlled. In addition, this embodiment utilizes the fact that a change in current caused by a change in the magnitude of the magnetic flux is detected as an exciting current component Id parallel to the magnetic flux in the rotating magnetic field coordinate system. 801 to 807 constitute a current control system, and its output signals ΔVd * and ΔVq *
Is calculated according to the rotation frequency ωr or the primary angular frequency command ω ₁ * for the inverter.

Further, as a result of controlling the magnetic flux of the AC motor to a predetermined value, the change in current due to the change in slip frequency is
It is detected as a torque current component Iq perpendicular to the magnetic flux in the rotating magnetic field coordinate system. Therefore, in the first embodiment of the present invention, the frequency control system based on the torque current component Iq is used as the functional element 8.
The output signal Δω ₁ * is added to the output frequency of the inverter.

Next, the operation of the above-described embodiment of the present invention will be described using mathematical expressions.

Rotating magnetic field coordinate system of the AC motor (dq axes)
The following magnetic flux components φ _d and φ _q , the voltage components Vd and Vq, and the output signals ΔVd * and ΔVq * of the current regulators 804 and 807 have the following relationship. Voltage component Vd,
Vq is supplied from the inverter 1 as the sum of the voltage component commands Vd *, Vq * calculated and output by the voltage command calculator 5 and the output signals of the current regulators 804, 807.

Φ _d = [M / Z (s)] · (Vd + ω ₁ · ls · Iq + ω ₁ · φ _q ) (1) φ _q = [M / Z (s)] · (Vq−ω ₁ · _{ls · Id-ω 1 · φ} d -r 1 · Iq) ...... (2) Vd = Vd * + ΔVd * ...... (3) Vq = Vq * + ΔVq * ...... (4) Z (s) = r 1 · (1 + T ₂ · s) · (1 + T _sa · s) (5) where M is the excitation inductance, ls is the leakage inductance, ωr is the rotation frequency, ω ₁ is the inverter output frequency, and T ₂ is the secondary time constant. , T _sa is a leak time constant, and s is a Laplace operator.

Further, the output signal Vd of the voltage command calculator 5
* And Vq * are expressed by the following equations.

Vd * = r ₁ · Id * −ω ₁ * · ls · Iq * (6) Vq * = r ₁ · Iq * + ω ₁ * · ls · Id * + ω ₁ * · φ _d * (7) In the embodiment shown in FIG. 1, in the equilibrium state (steady state), the output signals Vd of the voltage command calculator 5 are set so that the current components Id and Iq match the command values Id * and Iq *. *,
Controlled by Vq *. At this time, ΔVd * = 0, Δ
Vq * = 0, and the magnetic flux components φ _d and φ _q of the AC motor are
From the above equations (1) to (7), it is obtained as shown in the following equation.

Φ _d = [M / Z (s)] ・ (r ₁・ Id * + ω ₁ * ・ φ _q ) ... (8) φ _q = [M / Z (s)] ・ [ω ₁ * ・(Φ _d * −φ _d )] (9) When the equations (8) and (9) are arranged with respect to φ _d , the following equation can be obtained.

Φ _d = [M · r ₁ · Id * / Z (s)] / [1+ {M · ω ₁ * / Z (s)} ² ] + [{M · ω ₁ * / Z (s)] } ² · φd *] / _{[1+ {M · ω 1 * /} Z (s)} 2 ] ... (10) this equation (10) the first term on the right side is the magnetic flux due to the voltage component of the d-axis, right side The second term is the magnetic flux due to the q-axis voltage component.

FIG. 4 shows the first right side of the equation (10) for the primary angular frequency command ω ₁ * which is the output frequency command of the inverter.
The term, the second term, and the sum of the first and second terms are shown. From this figure, it is understood that the voltage component contributing to the magnetic flux changes according to the primary angular frequency command ω ₁ * of the inverter. . That is, in the low speed range where the primary angular frequency command ω ₁ * of the inverter is small, the first term on the right side of formula (10) is dominant, and in the middle to high speed regions, the second term on the right side of formula (10) is dominant.

Therefore, in order to control the magnetic flux with high accuracy, the exciting current Id accompanying the fluctuation of the magnetic flux is detected, and the current regulators 804 and 8 are adjusted so that Id coincides with the command Id *.
07, output signals ΔVd *, ΔVq * are generated, and these output signals ΔVd *, ΔVq * are added to the d-axis or q-axis voltage component according to the rotation frequency ωr or the primary angular frequency command ω ₁ * to the inverter. do it. The function generators 801 and 805 perform this switching.

This can be expressed by the following equation.

Φ _d = [M · (r ₁ · Id * + ΔVd *) / Z (s)] / [1+ {M · ω ₁ * / Z (s)} ² ] + [M · {ω ₁ * / Z (s)} ² · (M · φd * + ΔVq *)] / [1+ {M · ω ₁ * / Z (s)} ² ] (11) Next, the torque current control operation will be described.

In the first embodiment of the present invention, the magnetic flux of the AC motor 2 can be controlled to a predetermined value φ _d * by performing the magnetic flux control as described above. When an error in the slip frequency command occurs due to a change in the secondary resistance of the AC motor 2 due to a temperature change in the AC motor 2 or a detection delay of the speed detector 16, the change is caused by a torque perpendicular to the magnetic flux. The current component Iq changes.

Therefore, in the above-described embodiment of the present invention, the change in the torque current is calculated by the adder 808 as a deviation value, and the current controller 810 sets the deviation value to 1 so that the deviation value becomes zero.
The deviation Δω ₁ * of the next angular frequency command is calculated. Adder 24
Determines the primary angular frequency command ω ₁ * by adding the deviation Δω ₁ * of the primary angular frequency command, the slip frequency command ωs *, and the rotation frequency ωr.

As a result, the first embodiment of the present invention can correct the slip frequency error of the AC motor 2.

In the first embodiment of the present invention described above, when the primary angular frequency command, which is the output frequency of the inverter, is less than or equal to a predetermined value, the output of the current control for controlling the excitation component is changed to the voltage of the excitation component. In this case, in the present invention, when the primary angular frequency, which is the output frequency of the inverter, is less than or equal to a predetermined value, the current control for controlling the excitation component is controlled by proportionality and integration. When the output frequency of is above a predetermined value, the current control for controlling the excitation component can be proportional control.

FIG. 5 is a block diagram showing the configuration of the second embodiment of the present invention, FIG. 6 is a block diagram showing the configuration of the voltage command calculator, and FIG. 7 is a block diagram showing the configuration of the current controller. 5 to 7, 5a is a voltage command calculator, 8a
Is a current controller, 509 is an adder, 807a is a magnetic flux controller, and other reference numerals are the same as those in FIGS.

A second embodiment of the present invention shown in FIG. 5 is shown in FIG.
The difference from the first embodiment shown in FIG. 3 is that a magnetic flux controller is provided so that the exciting current component Id parallel to the magnetic flux of the rotating magnetic field coordinate system matches the command Id *, and the medium to high speed AC motor is provided. In the region, the magnetic flux command φ
_The point is that _d * is modified.

Next, referring to FIGS. 6 and 7, the second embodiment of the present invention will be described.
The configurations of the voltage command calculator 5a and the current controller 8a of this embodiment will be described.

The voltage command calculator 5a shown in FIG. 6 is different from the voltage command calculator 5 of the first embodiment shown in FIG.
An adder 509 is provided to add a magnetic flux command φd * and an output signal Δφ _d * of a magnetic flux adjuster 807a described later, and adder 5
The output signal φ _d ** of 09 is input to the adder 506.

In addition, the current controller 8a shown in FIG.
The difference from the current controller 8 of the first embodiment shown in (1) is that a magnetic flux controller 807a is provided instead of the current controller 807, and the output signal Δφd * of the magnetic flux controller 807a
The point is that the magnetic flux command φd * is corrected.

Next, the operation of the second embodiment of the present invention will be described using equations.

The magnetic flux component φ _d of the AC electric motor 2 is the above-mentioned (1
The ratio of the first term on the right side and the second term on the right side of the equation (10) contributing to the magnetic flux changes according to the inverter output frequency ω ₁ *. Therefore, the second embodiment of the present invention, by .DELTA.Vd * in the low-speed range the primary angular frequency command omega ₁ * is small for the inverter and by Derutafaid * at medium to high speed range, to control the magnetic flux component phi _d I have to. That is, according to the second embodiment of the present invention, the magnetic flux component φ _d is represented by the following equation.

Φ _d = [M · (r ₁ · Id * + ΔVd *) / Z (s)] / [1+ {M · ω ₁ * / Z (s)} ² ] + [{M · ω ₁ * / Z (s)} ² · (φ _d * + Δφ _d *)] / [1+ {M · ω ₁ * / Z (s)} ² ] (12) The above is according to the second embodiment of the present invention. This is a magnetic flux control operation.

FIG. 8 is a block diagram showing the configuration of the third embodiment of the present invention. In FIG. 8, reference numeral 25 is a master controller, and other reference numerals are the same as those in FIG.

The third embodiment of the present invention is different from the first embodiment of the present invention shown in FIG. 1 in that the input signal of the control command generator 6 is supplied from the master controller 25. That is the point.

The master controller 25 outputs a notch command N * and a train traveling direction command D * to the control command generator 6. The control command generator 6 outputs the filter capacitor voltage to the voltage detector 1
4, the excitation current command Id * and the torque current command Iq * for the AC motor 2 are calculated based on the filter capacitor voltage, the inverter output frequency ω ₁ *, the notch command N *, and the train traveling direction command D *. .

According to the third embodiment of the present invention, the present invention can be applied to a train driving device which does not use a speed controller.

FIG. 9 is a block diagram showing the configuration of the fourth embodiment of the present invention. In FIG. 9, reference numeral 26 is a speed estimator, and other symbols are the same as those in FIG.

The difference of the fourth embodiment of the present invention from the first embodiment of the present invention shown in FIG. 1 is that instead of using the speed detector 16 for detecting the speed of the AC motor 2, An estimator 26 is provided, and an output signal ωr of this speed estimator 26
Based on the above, the primary angular frequency command ω ₁ * which is the inverter output frequency is calculated.

In the fourth embodiment of the present invention, the output signal Δω ₁ * of the current controller 8 is compensated for the change of the secondary resistance due to the temperature change of the AC motor 2 and the estimation error of the speed estimator 26. Signal is included. When a load is connected to the electric motor, its speed changes only according to the inertia of the load even if torque is applied. Therefore, the speed estimator 26
It can be configured by a simple first-order delay circuit, and the motor speed can be estimated by the signal from the speed commander 19.

In the fourth embodiment of the present invention, therefore, the magnetic flux and torque of the AC motor 2 can be controlled without using the speed detector 16.

In the above-described embodiment of the present invention, the voltage command of the excitation component and the torque component is controlled by the output frequency of the inverter, but the present invention controls these by the rotation speed of the AC motor. You can also do it.

[0066]

As described above, according to the present invention, the magnetic flux of the driven AC electric motor is controlled so as to match the command value even when the DC power source voltage fluctuation is large like the inverter device for electric train control. Furthermore, since the setting error of the slip frequency can be compensated by the change of the inverter output current, the torque control of the electric motor can be performed with high response and high accuracy without using a voltage detector. In particular, when applied to a train or the like, the acceleration performance can be improved.

[Brief description of drawings]

FIG. 1 is a block diagram showing a configuration of a first exemplary embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a voltage command calculator.

FIG. 3 is a block diagram showing a configuration of a current controller.

FIG. 4 is a diagram illustrating characteristics of an output frequency of an inverter and a voltage component contributing to magnetic flux.

FIG. 5 is a block diagram showing a configuration of a second exemplary embodiment of the present invention.

FIG. 6 is a block diagram showing a configuration of a voltage command calculator.

FIG. 7 is a block diagram showing a configuration of a current controller.

FIG. 8 is a block diagram showing a configuration of a third exemplary embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a fourth exemplary embodiment of the present invention.

[Explanation of symbols]

 1 PWM Inverter 2 AC Motor 3 PWM Signal Calculator 4, 9 Coordinate Converter 5 Voltage Command Calculator 6 Control Command Generator 7 Speed Controller 8, 8a Current Controller 10 Slave Frequency Command Calculator 16 Speed Detector 801, 805 , 811 Function generator 804, 807, 810 Current regulator

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Toshiaki Okuyama 4026 Kujimachi, Hitachi City, Ibaraki Prefecture Hitachi Research Laboratory, Hitachi, Ltd.

Claims (5)

[Claims] 1. A control device for an inverter device, wherein a primary current of an AC electric motor is separated into an excitation component and a torque component of a rotating magnetic field coordinate system for current control, and an AC having a variable voltage and a variable frequency is supplied to the AC electric motor. , The output of the current control for controlling the excitation component is distributed to the voltage command of the excitation component and the voltage command of the torque component at a ratio according to the output frequency of the inverter or the rotation speed of the AC motor. A control device for an inverter device, characterized by adding and adding. 2. A control device for an inverter device, wherein a primary current of an AC electric motor is separated into an excitation component and a torque component of a rotating magnetic field coordinate system for current control, and an AC having a variable voltage and a variable frequency is supplied to the AC electric motor. If the output frequency of the inverter or the rotation speed of the AC motor is less than or equal to a predetermined value, the output of the current control for controlling the excitation component is added to the voltage command of the excitation component, and the output frequency of the inverter or the rotation speed of the AC motor. Is greater than or equal to a predetermined value, the output of the current control for controlling the excitation component is added to the voltage command of the torque component. 3. A control device for an inverter device, which separates a primary current of an AC electric motor into an excitation component and a torque component of a rotating magnetic field coordinate system for current control and supplies an AC of a variable voltage variable frequency to the AC electric motor. If the output frequency of the inverter or the rotation speed of the AC motor is less than or equal to a predetermined value, the output of the current control for controlling the excitation component is added to the voltage command of the excitation component, and the output frequency of the inverter or the rotation speed of the AC motor. Is greater than or equal to a predetermined value, the magnetic flux command is added to the voltage command of the torque component. 4. When the output frequency of the inverter or the rotation speed of the AC motor is less than or equal to a predetermined value, the current control for controlling the excitation component is controlled by proportional and integral, and the output frequency of the inverter or the rotation speed of the AC motor. The control device for an inverter device according to claim 1, 2 or 3, wherein the current control for controlling the excitation component is proportional control when is greater than or equal to a predetermined value. 5. The inverter device according to claim 1, wherein a notch command and a traveling direction command from the master controller are input to a control command generator for controlling the inverter. Control device.