JP3241252B2 - AC motor control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an AC motor which performs current control on rotating coordinates.
In particular, the present invention relates to a control device suitable for controlling an AC motor that frequently performs power regeneration operation and energy saving operation.

[0002]

2. Description of the Related Art As a power control device for controlling a three-phase AC motor, a control method for feeding back current in a stationary coordinate system (the former) and a feedback control of current in a rotating coordinate system that matches the magnetic flux of the AC motor are used. There is a control method (the latter). The former method does not require complicated calculations such as coordinate conversion, and is a commonly used control method, for example, a current control method described in Japanese Patent Application Laid-Open No. 5-153705. .

[0003] On the other hand, the latter is a method suitable for a digital control device having a limited response due to sampling time because current can be handled as a DC amount.
In particular, when the primary frequency of the AC motor is several hundred Hz or more and digital operation is performed, the latter is advantageous.

As the latter control method, for example, the one described in FIG. 3 of Symposium S8-4 “Semiconductor Converter in Vector Control” of the 1983 IEEJ National Convention is cited as a first known example. be able to.

In this first known example, a primary current of an AC motor is decomposed into a d-axis which coincides with a magnetic flux and a q-axis which is orthogonal thereto, and each current component is fed back for each current command. , Proportional and integral calculations, and this method is generally used because it is easy to understand because it aims at reducing the current deviation of each axis to zero.

On the other hand, for example, Japanese Patent Application Laid-Open No. 59-169369
The latter control method is also proposed in Japanese Unexamined Patent Publication (Kokai) No. H11-27083. If this is the second known example, in this method, the d-axis current deviation is q
The axis voltage is integrated with the d-axis voltage in accordance with the q-axis current deviation in accordance with the angular frequency ω used for the coordinate conversion. As a result, higher stability is obtained in the current control than in the first known example. Can be given.

However, the second known example has a problem that since the angular frequency ω greatly changes, the width of change of the integral gain becomes large, and the control response greatly changes. For example, if the rotation speed of the motor is 1
When the speed is suddenly changed from the high speed region of 0000 RPM to the low speed region of 100 RPM, it takes 100 times as long to correct the voltage integrated value calculated in the high speed region.

[0008] Under such circumstances, the second control method capable of driving without any problem in a normal operation state is generally adopted in many products.

[0009]

In the case of a crane or hoist using such an AC motor control device, when the load is light, the rotation speed of the motor increases from the normal rotation speed. At this time, when the load is decreased, the regenerative operation state in the field weakening region is performed at a high speed.

On the other hand, in the case where the AC motor control device is used in an electric vehicle, the efficiency is given the highest priority in order to extend the round trip distance, and when the driving torque is small, the field weakening control is frequently used. At this time, when the vehicle travels on a long downhill, the regenerative operation is performed for a long time. In particular, on a steep downhill, the vehicle is driven at a high speed in a regenerative state in a weak field region.

However, when the current control method shown in the second known example is applied in the regenerative operation state in the field weakening region, the stability of the control system may be reduced.
FIG. 18 shows that the field-weakening control is performed in the regenerative operation state, and
FIG. 4 shows a voltage and current vector diagram of the induction motor during high-speed operation. Considering a vector diagram relating to the integration operation, a current deviation vector Δi1 which is a difference between the current command vector i1 * and the current vector i1 is shown. *, An integral correction voltage vector V2 * is added and applied in the same direction.

In this case, the applied voltage vector V
Since * becomes smaller than the original voltage command vector V1 *, the current vector i1 may be separated rather than approaching the current command vector i1 *.
In particular, depending on the operating conditions of the AC motor control device,
It has been found that when the three conditions of the regeneration and the field weakening control overlap, the stability is greatly reduced in some cases.

SUMMARY OF THE INVENTION An object of the present invention is to provide a current control system which is excellent in stability even in the case of load disturbance and regeneration in an AC motor control device which always requires small size and high efficiency driving. Another object of the present invention is to provide an AC motor control device which is excellent in following up an operation command such as a position command, a speed command, and a torque command.

[0014]

SUMMARY OF THE INVENTION An object of the present invention is to provide an AC motor, a power converter for generating electric power to be supplied to the AC motor, and generating a magnetic flux of the AC motor from a torque command value to be generated by the AC motor. A current command value generator for calculating a d-axis current command value and a q-axis current command value orthogonal to the d-axis current command value, and feeding back the d-axis current of the AC motor and the q-axis current orthogonal thereto, The d-axis voltage command value is obtained from the d-axis current deviation between the value and the d-axis current, and the q-axis current command is obtained from the q-axis current deviation between the q-axis current command and the q-axis current.
In an AC motor control device including a current control device that controls the power conversion device by calculating a shaft voltage command value, the current control device uses a vector phase based on a driving state of the AC motor, This is achieved by calculating a corrected q-axis voltage and a corrected q-axis voltage from the d-axis current deviation and the q-axis current deviation, and providing a voltage vector correction unit that corrects the d-axis voltage and the q-axis voltage.

Further, the other object is achieved by providing feedforward voltage means for previously calculating an applied voltage to be generated based on a d-axis current command value and a q-axis current command value.

In the AC motor control device, a d-axis current command value for generating a magnetic flux of the AC motor from a torque command value and a q-axis current command value orthogonal thereto are calculated, and the d-axis current of the AC motor is orthogonal to the d-axis current. The q-axis current is fed back, and a d-axis voltage command value and a q-axis voltage command value are calculated from the respective current deviations. Based on these values, the power converter is driven to control the current of the AC motor.

Here, the driving state is detected from the applied voltage and the primary current of the AC motor, and the vector phase is calculated based on the driving state. This vector phase is a phase of the correction voltage for stably setting the current deviation to 0 with respect to the current deviation.

Therefore, a corrected d-axis voltage and a corrected q-axis voltage are calculated so that a corrected voltage is generated in a direction advanced by the obtained vector phase with respect to the d-axis current deviation and the q-axis current deviation. At this time, it is particularly important to change the phase of the voltage to be corrected according to the driving state of the motor such as the applied voltage and the phase difference between the primary currents.

The applied voltage is corrected by adding these voltages to the d-axis voltage and the q-axis voltage, respectively. As a result, the voltage is added in a direction suitable for the driving state of the motor, so that the current deviation can be stably converged even under special load conditions such as regeneration, high-speed rotation, and field weakening. In the steady state, the primary current is made to match the command value. Therefore, it is possible to obtain an AC motor control device with more stable current control characteristics performed in the magnetic flux coordinate system.

Further, by inputting an applied voltage calculated from an equivalent circuit in a steady state using a d-axis current command value and a q-axis current command value as a feedforward signal in advance, a normal current control calculation and newly added Since the correction voltage calculation only needs to generate a voltage for compensating the current deviation, the responsiveness and the time to a steady state can be reduced. Thereby, the torque controllability of the AC motor control device can be further improved.

[0021]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, an AC motor control device according to the present invention will be described in detail with reference to the illustrated embodiments.
FIG. 1 shows an embodiment of the present invention. In the figure, reference numeral 1 denotes an induction motor, 2 denotes a three-phase power conversion circuit, and 3 denotes a DC power supply. It is configured to convert into AC power and drive the induction motor 1. The DC power supply 3 may use a battery as shown in the figure, but when driven from an AC power supply,
The AC power is rectified to form a DC power supply 3.

Reference numeral 7 denotes a control device which includes a torque command generation circuit 8, a current command generation circuit 9, a current control circuit 10, a coordinate conversion circuit 11, and various circuits such as a PWM generation circuit 12, and controls the power conversion circuit 2. Required for each phase
Signals Pu, Pv, and Pw are generated, thereby controlling the power conversion circuit 2, converting the DC power of the power supply 3 into three-phase AC power, and controlling the drive of the induction motor 1.

Next, the operation of the control device 7 will be described. A torque command value τ * is generated from the torque command generation circuit 8 and is input to a current command generation circuit 9 where
By a known vector control method, a d-axis current command value id * that matches the magnetic flux of the induction motor in the rotating coordinate system and a q-axis current command value iq * that is orthogonal thereto are calculated. Then, the product of the d-axis current command value id * and the q-axis current command value iq * becomes a value proportional to the torque command value τ *.

The current command generation circuit 9 calculates a slip angular frequency command value ωs * of the induction motor 1 from the d-axis current command value id * and the q-axis current command value iq *. The value ωs * is also the d-axis current command value id
* And the q-axis current command value iq * are output.

The d-axis current command value id * and the q-axis current command value iq * are input to the current control circuit 10,
The d-axis voltage command values Vd * and q
The shaft voltage command value Vq * is output. These various values are values in the rotating coordinate system, and the coordinate conversion circuit 11 uses the rotation angle command value θ * to output the three-phase AC current command value vu of the stationary coordinate system.
*, Vv *, vw * are determined.

Here, the rotation angle command value θ * is calculated from the rotation angle calculation circuit 14 by the following processing. That is, the rotational position of the induction motor 1 obtained from the position detector 13 is supplied to a speed calculating circuit 40, where the speed ω _M is obtained by time differentiation, and the speed ω _M and the above-mentioned slip angular frequency command value ωs The rotation angle command value θ * is obtained by calculating the primary angular frequency ω <b> 1 by adding *. Therefore, the rotation angle command value θ * means a command value of the phase of the rotating magnetic flux of the induction motor 1.

The AC current command values Vu *, Vv *, Vw * output from the coordinate conversion circuit 11 are input to a PWM generation circuit 12, where they are compared with a triangular carrier signal fc, and a PWM signal Pu of each phase is output. , Pv, Pw.

Then, these PWM signals Pu, Pv,
The switching element of the power conversion circuit 2 is controlled by Pw, and the DC power from the DC power supply 3 is converted into three-phase AC power, supplied to the induction motor 1, and drive-controlled.

On the other hand, as a result, the current flowing in each phase of the primary winding of the induction motor 1 is detected by the current sensor 15, and the coordinate conversion circuit 16 changes the stationary coordinate system to the rotating coordinate system based on the rotation angle command value θ *. And the d-axis current i
These are output as d and q-axis currents iq, which are input to the current control circuit 10.

Next, the current control circuit 1 which is a feature of the present invention
0 will be described in detail. The d-axis current command value id * output from the current command generation circuit 9 and the q-axis current command value iq
* Denotes d output from the coordinate conversion circuit 16
The d-axis current deviation Δid * and the q-axis current deviation Δiq * which are compared with the axis current id and the q-axis current iq are obtained.

Therefore, the q-axis current control circuit 17 calculates and outputs a first q-axis voltage command value Vq1 * from the q-axis current deviation Δiq * by proportional operation or proportional integral operation, and similarly outputs d The shaft current control circuit 18 calculates and outputs a first d-axis voltage command value Vd1 * from the d-axis current deviation Δid *.

Next, in the voltage / current phase difference calculation circuit 19,
The voltage-current phase difference θc is calculated from the d-axis voltage command value Vd *, the q-axis voltage command value Vq *, the d-axis current id, and the q-axis current iq.

FIG. 2 is a vector diagram of the applied voltage vector V1 and the primary current vector i1 in the induction motor 1. At this time, the applied voltage vector V1 substantially coincides with the first voltage command vector V1 *. D
It is represented using the shaft voltage command value Vd * and the q-axis voltage command value Vq *.

Next, the voltage-current phase difference θc is calculated according to the flowchart of FIG. That is, first, as is clear from the vector diagram of FIG. 2, the voltage phase θv is calculated in step 101 from the d-axis voltage command value Vd * and the q-axis voltage command value Vq * using a trigonometric function. Similarly, in step 102, the current phase θi of the primary current vector i1 becomes d
It is obtained from the axis current id and the q-axis current iq. Therefore, the voltage-current phase difference θc can be calculated in step 103 from the difference between the voltage phase θv and the current phase θi.

Next, in step 104, it is determined whether the phase difference θc between the voltage and the current is in the range of negative 0 ° to 90 ° or exceeds 90 °. First, when the phase difference θc is negative, step 1
05, θc = 0 °, while the phase difference θc is 90 °.
Is exceeded, in step 106, θc is set to 90 °. That is, the phase difference θc always takes a value in the range of 0 to 90 °. Here, this restricted phase difference θc is defined as a vector phase.

The vector phase θc is input to the second voltage command circuit 20, where the vector phase θc is used together with the d-axis current deviation Δid * and the q-axis current deviation Δiq * as shown in the flowchart of FIG. Perform processing.

First, in steps 107 and 108, a d-axis integral input value ΔVd2 * and a q-axis integral input value ΔVq2 * are calculated by the illustrated operations. Next, step 10
In steps 9 and 110, the d-axis integral input value ΔVd2 * and the q-axis integral input value ΔVq2 * are multiplied by gains kd and kq, respectively, and integrated to obtain a second d-axis voltage command value Vd2 * and a second Calculated as q-axis voltage command value Vq2 *.

These operations are performed by calculating the current deviation vector Δi1
Means that the second voltage command vector V2 * is integrated in a phase direction in which * is advanced by the vector phase θc,
Therefore, this relationship is as shown in the vector diagram of FIG.

As is apparent from the vector of FIG.
In the case where the second voltage command circuit 20 is not used, that is, in the related art, 1 is applied to the first voltage command vector V1 *.
Since the secondary current vector i1 is generated, the primary current vector i1 is delayed from the current command vector i1 *.
The phase relationship of the load of the induction motor 1 is shifted.

However, in the present embodiment, the second voltage command circuit 20 is provided, so that the primary voltage vector i1 matches the current command vector i1 *. Is calculated and added to the first voltage command vector V1 * to become the voltage command vector V *.
The next current vector i1 can be made to match the current command vector i1 *, and control can be performed without breaking the phase relationship of the induction motor 1.

Therefore, by providing the second voltage command circuit 20, it is possible to easily obtain a voltage command vector V1 * in which the primary current vector matches the current command vector in a steady state. On the other hand, at the time of regeneration, the phase difference between the applied voltage and the primary current may exceed 90 °. In this case, however, it is possible to obtain a stable current control system by setting the vector phase θc to 90 °. Can be. Therefore, according to this embodiment, it is possible to provide an AC motor control device having always stable current control in any operation state.

Next, another embodiment of the present invention will be described. In the following embodiments, only the control device 7 is described for simplification.

FIG. 6 shows another embodiment of the present invention.
The embodiment of FIG. 6 also has the same basic configuration as the embodiment of FIG. 1, but differs from the embodiment of FIG. 1 in that non-interference control circuits 25 and 26 are added, The two points are that a sine wave generation circuit 24 is provided instead of the phase difference calculation circuit 19, and therefore, these differences will be mainly described below.

First, the non-interference control circuits 25 and 26 calculate in advance the voltage vector V3 * that is originally required for flowing the d-axis current command value id * and the q-axis current command value iq * by feedforward. It functions to apply the voltages Vd1 * and Vq1 *.

Specifically, the non-interference control circuit 25,
In 26, each _{Vq3 * = k3ω M · id *} Vd3 * = calculates the k2ω _M · iq *, d-axis voltage command value is to add to the q-axis voltage command value. Here, the motor speed omega _M no may be used primary angular frequency ω1 of the power converter 2.

Next, the sine wave generation circuit 24 uses the d-axis voltage command value Vd * and the q-axis voltage command value Vq * output from the current command generation circuit 9 and uses them together with the d-axis current command value id.
*, Using the q-axis current command value iq *,
The cosine wave signal Vc * proportional to c and the sine wave signal Vs * proportional to Sin θc are calculated.

Vc * = vd * · id * + vq * · iq * Vs * = vq * · id * −vd * · iq * As apparent from these equations, in the embodiment of FIG. D-axis current id and q obtained from 15 detected values
In contrast to the calculation using the axis current iq, in this embodiment, the d-axis current command value id * and the q-axis current command value iq *
Therefore, according to this embodiment, there is an advantage that it is not affected by the current pulsation or the current sensor.

In the second voltage command circuit 20, unlike the embodiment of FIG. 1, these cosine wave signals Vc
Using * and the sine wave signal Vs *, an operation corresponding to FIG. 4 is performed. That is, in steps 107 and 108 of FIG. 4, Vc * and Vs * are used instead of Cos θc and Sin θc.
It is calculated using.

When the calculation is performed by this method, simpler calculation processing can be performed as compared with the calculation method of the embodiment shown in FIG. 1, so that there is a feature that the calculation time can be reduced when a single-chip microcomputer is used.

In this embodiment, since the non-interference control is added to the current control system, the applied voltage obtained from the equivalent circuit must be input in advance, and the current control is performed by the error of the equivalent circuit and the load fluctuation. It works only for the influence. Therefore, the stability of the current control system can be further improved. Further, since a stable current control system can be configured by a simpler calculation method, a highly reliable AC motor control device can be obtained.

Next, FIG. 7 shows another embodiment in which the method of obtaining the vector phase θc is different, in which a speed phase circuit 27 is used instead of the voltage / current phase difference calculation circuit 19 in the embodiment of FIG. is there. Rotation angle calculation circuit 1 in embodiment of FIG.
4, the primary angular frequency ω1 is calculated as described above. Therefore, in the embodiment of FIG. 7, this primary angular frequency ω1 is input to the speed phase circuit 27, and the vector phase θ
c is calculated and output to the second voltage command circuit 20.

The processing contents of the speed phase circuit 27 are as follows.
FIG. 8 shows the relationship between the secondary angular frequency ω1 and the vector phase θc in advance.
Is a table search process in which a table is prepared as shown in FIG.

As is apparent from FIG. 8, when the primary angular frequency ω1 is 0, the vector phase θ
This is a table obtained by calculating c so that the vector phase θc approaches 90 degrees (deg) when c becomes 0 and the primary angular frequency ω1 becomes a high frequency.

Then, the second voltage command value vd for the current deviations Δid * and Δiq * is determined by the vector phase θc.
Since 2 * and vq2 * are integrated by advancing the phase in a vector manner, the second voltage command value is increased in the direction substantially equal to the current deviation at low speed, and the second voltage command value is increased by 90 degrees from the current deviation at high speed. The voltage command value will be increased.

Therefore, according to this embodiment, a stable current control system can be constructed regardless of whether the rotation speed of the motor is low or high.

Next, FIG. 9 shows an embodiment of the present invention in which the integral gain Gc of the second voltage command value is made variable by the vector phase. The difference lies in that a second d-axis voltage command circuit 29 and a second q-axis voltage command circuit 28 are separately used instead of the voltage command circuit 20, and that an integral gain operation circuit 30 is provided.

First, in the integral gain calculation circuit 30, the d-axis voltage command value vd *, the q-axis voltage command value vq *, and the d-axis current i
d, and the vector phase θc is calculated from the q-axis current iq. The calculation method at this time is the same as that in the embodiment of FIG. Further, this integral gain operation circuit 3
At 0, the integral gain Gc is calculated based on the vector phase θc using the table shown in FIG.

Next, the second d-axis voltage command circuit 29 and the second
In the q-axis voltage command circuit 28, the second d-axis voltage command value Vd2 * and the second q-axis voltage command Vq
2 * is calculated.

Vd2 * = Gc.Δiq * + Vd2 * Vq2 * = Gc.Δid * + Vq2 * As a result, when the vector phase θc is 0 degrees, as shown in FIG. Therefore, the outputs from the second d-axis voltage command circuit 29 and the second q-axis voltage command circuit 28 are made substantially 0, and in this case, the q-axis current control circuit 17 and the d-axis Current control circuit 18
Is controlled only by the first voltage command values Vq1 * and Vd1 * obtained in the above.

For this reason, in this embodiment, the q-axis current control circuit 17 and the d-axis current control circuit 18 are more suitable for circuits that operate not by proportional operation but by proportional integral operation.
The maximum integral gain Gcm is obtained when the vector phase θc is 90 degrees.

As a result of this calculation, when the vector phase θc is large, the second voltage command values Vq2 *, Vd2 *
By increasing these effects, the stability of current control can be improved. Therefore, the embodiment of FIG. 9 has a feature that equivalent current control characteristics can be obtained by a simpler calculation method than the embodiment of FIG.

Next, FIG. 11 shows another embodiment of the present invention. This embodiment is basically the same as the embodiment of FIG. 9 except that the influence of the second voltage command value depends on the primary angular frequency ω1. Is different from the embodiment of FIG.

For this reason, as shown in FIG. 11, the primary angular frequency ω1 calculated by the rotation angle calculation circuit 14 is input to the second d-axis voltage command circuit 29 and the second q-axis voltage command circuit 28. , Each of which performs the following operation. Vd2 * = k4 · ω1 (ΣΔiq *) Vq2 * = k4 · ω1 (ΣΔid *) As a result of this operation, when the primary angular frequency ω1 is large,
The current control system includes a second d-axis voltage command value vd2 * and a second q
It is strongly influenced by the shaft voltage command vq2 *.

As a result, when the primary angular frequency ω1 is reduced, the second d-axis voltage command value Vd2 * and the second q-axis voltage command Vq2 * are directly reduced. D-axis voltage command value Vd1 * and first q-axis voltage command Vq
The effect of 1 * increases.

Therefore, according to the embodiment, the motor speed ω _M
When the speed is suddenly decelerated from a high speed, the second voltage command value also decreases immediately, so that the time required for the current to settle in a steady state at that time can be shortened, and the responsiveness improves.

FIG. 12 shows another embodiment of the present invention in which the stability of the current control system can be ensured by a simpler operation. Instead, a permanent magnet field type synchronous motor is used.

The use of this permanent magnet synchronous motor makes the control system different in that the magnetic pole position θ _M obtained from the magnetic pole position detector is used instead of the rotation angle command value θ *.
In that it is configured to use

In the case of such a control of the permanent magnet type synchronous motor, the d-axis current command id * is usually controlled to 0. Can be changed to effect weak field.

In FIG. 12, the current control operation is performed by the q-axis current control circuit 17 and the d-axis current control circuit 1.
8, which is the same as the conventional current control method of performing the proportional-integral operation. However, in this embodiment, a mode regeneration determination circuit 33 is added to this control system. It is a feature.

The mode regeneration determining circuit 33 receives the motor speed ω _M and the q-axis current command value iq *, and first determines whether or not the q-axis current command value iq * <0. next,
Determining whether more than a fast determination speed omega _M0 the motor speed omega _M is set in advance.

Then, the result is that iq * <0 and ω _M
> Ω _M0 , it means that the synchronous motor is rotating at high speed in the regenerative state, and the integration switching signal SW
Is set to 1; otherwise, the integration switching signal SW is set to 0.
To The integration switching signal SW is input to a q-axis current control circuit 17 and a d-axis current control circuit 18 as shown in the figure.
Here, when the integration switching signal SW is 0, a normal proportional integration operation is performed to control the primary current of the synchronous motor. On the other hand, when the integration switching signal SW is 1, the integration calculation is stopped, and only the proportional calculation is performed.

By this processing, the integral calculation is stopped only when the motor is rotating at high speed in the regenerative state, thereby preventing the stability of the current control system from being lowered by the integral calculation. An AC motor control device having good torque control can be obtained by a simple current control operation.

FIG. 13 shows an integration switching method, that is, a processing method of the mode determination circuit 34 in the control circuit 10 is as follows.
This is an embodiment of the present invention which is different from the mode determination circuit 33 in the embodiment of FIG. q-axis current control circuit 17
And the d-axis current control circuit 18 performs current feedback control by proportional integral calculation. The mode determination circuit 34
D-axis current command value id * obtained from current command generating circuit 6
And the q-axis current command value iq *, and in the regenerative state,
It is determined whether or not the field is weak, and the integration switching signal S
W is determined and output.

FIG. 14 is a flowchart showing the contents of the operation performed by the mode determination circuit 34. First, these signals are input in step 111, and in the next step 112, it is determined whether or not the q-axis current command value iq * is negative.

First, when the q-axis current command value iq * is negative, it is determined that the vehicle is in the regenerative state, and the process proceeds to step 113. On the other hand, when iq * is positive or 0, it is determined that the vehicle is not in the regenerative state, and in step 115, the integration switching signal SW is set to 0.

In step 113, the d-axis current command value id
* Is compared with a preset weak field determination value id0. If the d-axis current command value id * is equal to or greater than id0, it is determined that the field is not in the field weakening state and step 1
Proceeding to step 15, the integration switching signal SW is set to 0.

On the other hand, if the d-axis current command value id * is less than id0, it is determined that the field is in a weak field state and step 11
At 4, the integration switching signal SW is set to 1. That is, the integration switching signal SW is set to 1 when it is determined that the current operation mode is regenerative and the field is weak.

The integration switching signal SW is, as shown in FIG.
The signal is output from the mode determination circuit 34 to the q-axis current control circuit 17, the d-axis current control circuit 18, the second d-axis voltage command circuit 29, and the second q-axis voltage command circuit 28.

When the integration switching signal SW is 0, the q-axis current control circuit 17 and the d-axis current control circuit 18 perform an integration operation, and the second d-axis voltage command circuit 29
And the integration calculation in the second q-axis voltage command circuit 28 is stopped.

On the other hand, when the integration switching signal SW is 1, the integration calculation in the q-axis current control circuit 17 and the d-axis current control circuit 18 is stopped, and the second d-axis voltage command circuit 29 and the second The q-axis voltage command circuit 28 performs an integration operation.

As described above, by using the integration switching signal SW, only in the regenerative state and the weak field state,
An integration operation is performed by the second d-axis voltage command circuit 29 and the second q-axis voltage command circuit 28 to prevent the stability of the current control system from deteriorating. The burden can be reduced.

Therefore, according to the embodiment of FIG. 13, the stability of the current control system can be sufficiently ensured and the load on the microcomputer can be reduced, so that the processing can be used for diagnosis of the control system. There is an advantage that the reliability of the system can be further improved.

FIG. 15 shows still another embodiment of the present invention.
In the above-described embodiment, the integration operation is stopped only when the integration switching signal SW is 1, but in this embodiment, the integration operation is always stopped. That is, in this embodiment, only the proportional operation is performed in the current control circuit. However, since a steady-state error occurs in this case, in this embodiment, the voltage drop of the resistor is added by feedforward.

FIG. 15 shows an embodiment in which only the control for the d-axis is performed as an example. As shown in FIG. 15, the d-axis current control circuit includes a d-axis proportional control calculator 41. And a d-axis resistance voltage calculation circuit 4
The voltage drop Vdr calculated in 2 is expressed by the following equation:
It is configured to feed forward the command value Vd1 * calculated by the d-axis proportional control calculator 41.

Vdr = R · id * Note that in this embodiment, the d-axis current command id * is used as shown in the above equation, but the actual current i
It can also be realized by using d.

Vdr = R · id Also, as described above, in this embodiment, the present invention is applied only to the d-axis, but the q-axis may be added to this, or may be applied to only the q-axis. Good.

As described above, the present invention has been described in some embodiments, and in the case of an induction motor and a synchronous motor in different systems. However, the present invention may be implemented by combining these embodiments. . In the above embodiment,
Several methods have been proposed for changing the direction of the applied voltage to be added with respect to the current deviation, but any method that changes the direction of the applied voltage based on information on the driving state of the AC motor is proposed. The same effect will be produced.

Next, an application example of the AC motor control device according to the present invention will be shown, and its effectiveness will be described. First, Figure 1
6 is a system configuration diagram in the case where, for example, the AC motor control device according to the present invention shown in FIG. 1 is applied to a crane. Although not shown in FIG. 16, the control device 7 is configured as shown in FIG. Have been. And with this crane,
The driver operates the operation lever 201 to give an instruction to raise or lower the luggage 204. With this operation, the operation command generator 202 causes the control device 7 to rotate forward (FWD) / reverse.
(REV) and velocity signals (CF1 to CF4).

In the control device 7, these signals are received by the torque command generation device 8 to generate a torque command τ *, which is input to the current command generation circuit 9 in the control device 7 and guided by the control device 7 described above. The motor 1 is controlled, whereby the winding drum 203 is driven via a power transmission mechanism 5a such as a gear, and the load 204 is lifted and suspended.

At this time, speed control is necessary for such a crane (the same applies to a hoist). Therefore, the torque command generator 8 outputs the input speed signals (CF1 to CF1).
A speed command is generated according to F4), a speed control calculation is performed, and a torque command τ * is output.

By the way, in this crane (the same applies to a hoist), when the load 204 has a rated load, the induction motor 1 and the power conversion circuit 3 are selected so that the load can be lifted at the rated speed. Is required to be able to operate at a speed higher than the rated speed, and in this case, it is operated in a high-speed field-weakening state.

In this case, when the load 204 is lifted, the power running operation is performed. Therefore, there is no particular problem. However, when the load 204 is suspended, the regenerative operation is performed. When the load 204 is light and suspended, a high-speed weak field is generated. If the current control of the induction motor 1 becomes unstable in the high-speed field-weakening state and the regenerative state as in the related art, the load This leads to the falling of the battery 204, which is extremely dangerous for safety. As in the prior art, during power running operation, that is, during lifting,
The current control does not become unstable, so it can be lifted normally.However, if it becomes unstable during regenerative operation, it will not be possible to lower the lifted luggage, which will cause a considerable problem in use. It is understood that the present invention is effective in this respect.

Next, FIG. 17 shows an example of a system configuration in which the present invention is applied to an electric vehicle. In such an electric vehicle, as shown, the tire 6a is driven by the induction motor 1 via the drive shaft 5b. , 6b to cause the vehicle body 4 to travel. The driver has the accelerator 21 and the brake 2
2. By operating the changeover switch 23, the traveling of the automobile is controlled.
a, the depression amount xb of the brake 22, forward, backward, switch signal SDR of the changeover switch 23 for instructing stopping, and takes in the motor speed omega _M by signal theta _M from the position detector 13 in the control device 7, thereby Torque command generator 8
a calculates a torque command value τ * to be output by the induction motor 1.

The advantage of applying the current control method according to the present invention to such an electric vehicle is obtained when the vehicle travels on a long downhill. That is, in this case, normally, regeneration control is performed in which kinetic energy is regenerated by electric power and returned to the battery. Therefore, in the torque command generator 8a,
The torque command value τ * required for this regenerative control is calculated.

Further, in the electric vehicle, the field-weakening control is actively performed according to the motor torque in order to reduce the energy loss as much as possible. In such a case, in the conventional current control method, the stability may be reduced. On the other hand, according to the embodiment of the present invention, high stability can be maintained as in the case of power running. Therefore, it can be understood from this that the present invention is a very effective method at the time of high-speed running and at the time of regeneration.

In the above embodiment, the internal elements of the control device 7 are represented as circuits for simplicity of explanation, but can be realized by analog circuits or digital circuits.
Needless to say, it can be realized by processing by software of a microprocessor.

Here, the embodiments of the present invention are listed and listed as follows. Embodiment 1 The AC motor control device according to claim 1, wherein the vector phase is changed by a primary frequency of the power conversion device. Embodiment 2 The AC motor control device according to claim 1, wherein the vector phase is obtained by a product of d and q axis components of an applied voltage of the AC motor and components of the d and q axis currents. Embodiment 3 The AC motor control device according to claim 1, wherein the vector phase is obtained from a phase difference between a voltage applied to the AC motor and a primary current.

Fourth Embodiment In the second or third embodiment, the primary current is estimated from the d-axis current command value and the q-axis current command value. .

Embodiment 5 The AC motor control device according to claim 1, wherein the corrected d-axis voltage and the corrected q-axis voltage are calculated by an integral operation. Embodiment 6 The vector phase according to claim 1, wherein the vector phase is set to 0 degree to 90 degrees.
An AC motor control device characterized by being set within a range of degrees. Embodiment 7 In Claim 1, the reference q-axis voltage is calculated by the d-axis current command value, and the reference d-axis voltage is calculated by the q-axis current command value, and the q-axis voltage command value and the d-axis voltage command value are respectively calculated. AC motor control device characterized by adding

Embodiment 8 An AC motor control apparatus according to claim 2, wherein the corrected q-axis voltage and the d-axis voltage are obtained by integrating the d-axis current deviation and the q-axis current deviation, respectively. . Ninth Embodiment An AC motor control device according to the eighth embodiment, wherein the gain of the integration is changed by a phase difference between a voltage applied to the AC motor and a primary current. Embodiment 10 In Embodiment 8, the corrected q-axis voltage and the corrected d-axis voltage are obtained by integrating the d-axis current deviation and the q-axis current deviation, respectively, and multiplying each of the primary frequencies of the power converter. An AC motor control device characterized by being obtained by: Embodiment 11 In claim 2, the reference q-axis voltage is calculated based on the d-axis current command value, and the reference d-axis voltage is calculated based on the q-axis current command value. AC motor control device characterized by adding

Embodiment 12 The AC motor control device according to claim 3, wherein the operation mode is switched to the first or second calculation method when the operation state of the AC motor is regenerative. Embodiment 13 According to claim 3, when it is determined that the operating state of the AC motor is regenerative and high-speed rotation, the first,
Alternatively, the AC motor control device is switched to a second calculation method. Embodiment 14 The method according to claim 3, wherein when the operation state of the AC motor is regenerative and the d-axis current command value is reduced to perform the field weakening control, the method is switched to the first or second calculation method. An AC motor control device characterized by the above-mentioned.

(Embodiment 15) In Claim 4, the control operation of the d-axis is limited to the proportional operation of the current deviation, and the means for correcting the voltage based on the result of the operation and the resistance voltage drop of the d-axis from the feedforward amount is provided. An AC motor control device, characterized in that: Embodiment 16 The AC motor control device according to claim 4 or 15, wherein the resistance voltage drop is calculated from the voltage command. Seventeenth Embodiment An AC motor control device according to claim 4 or 15, wherein the resistance voltage drop is calculated from the detected current.

[0103]

According to the present invention, stable current control can be obtained even during regeneration or field weakening control.
There is an effect that an AC motor control device that can always operate with high efficiency can be provided. As a result, the present invention is particularly effective for a crane (hoist) that uses powering / regeneration, and is also effective for an electric vehicle that needs to drive downhill for a long time.

Further, according to the present invention, an applied voltage to be generated for each current can be input in a feedforward manner, so that the responsiveness of the AC motor control device can be improved.

[Brief description of the drawings]

FIG. 1 is a configuration diagram showing one embodiment of an AC motor control device according to the present invention when an induction motor is used.

FIG. 2 is a vector diagram when a relationship between an applied voltage of an induction motor and a primary current according to the present invention is viewed from a dq coordinate system.

FIG. 3 is a flowchart illustrating a calculation process of a vector phase θc by a voltage / current phase difference calculation circuit according to the embodiment of the present invention.

FIG. 4 is a flowchart illustrating a process for obtaining a second voltage command value from a vector phase θc by a second voltage command circuit in the embodiment of the present invention.

FIG. 5 is a vector diagram showing a relationship between a second voltage command value and a current deviation in the embodiment of the present invention.

FIG. 6 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention when a sine wave generation circuit is used.

FIG. 7 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention in which a vector phase θc is calculated based on a primary angular frequency.

FIG. 8 is a characteristic diagram of a table used when calculating a vector phase θc from a primary angular frequency in the embodiment of the present invention.

FIG. 9 shows an embodiment of an AC motor control device according to the present invention in which a second d-axis voltage command value is obtained from a q-axis current deviation, and a second q-axis voltage command value is obtained from the d-axis current deviation. It is a block diagram.

FIG. 10 is a characteristic diagram of a table used when calculating an integral gain Gc from a vector phase θc in the embodiment of the present invention.

FIG. 11 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention in which a second d-axis voltage command value is obtained based on a primary angular frequency and a second q-axis voltage command value is changed. .

FIG. 12 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention in which integration is stopped at a high speed in a regenerative state using a permanent magnet type motor.

FIG. 13 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention in which an integration calculation method is switched during field-weakening control in a regenerative state.

FIG. 14 is a flowchart illustrating a mode determination process according to the embodiment of the present invention.

FIG. 15 is a configuration diagram showing an embodiment of an AC motor control device according to the present invention in which current control calculation is always performed as a proportional calculation, and a voltage drop of a resistor is fed forward.

FIG. 16 is a system configuration diagram when the AC motor control device according to the present invention is applied to a crane.

FIG. 17 is a system configuration diagram when the AC motor control device according to the present invention is applied to an electric vehicle.

FIG. 18 is a vector diagram showing a phase relationship between a voltage and a current when regeneration is performed by an AC motor control device according to the related art.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Induction motor 2 Power conversion circuit 3 DC power supply 7 Control device 8, 8a Torque command generation device 9 Current command generation circuit 10 Current control circuit 11, 16 Coordinate conversion circuit 12 PWM generation circuit 13 Position detector 14 Rotation angle calculation circuit 15 Current Sensor 17 q-axis current control circuit 18 d-axis current control circuit 19 voltage-current phase difference calculation circuit 20 second voltage command circuit 21 accelerator 22 brake 23 change-over switch 24 sine wave generation circuit 25, 26 non-interference control circuit 27 speed phase circuit 28 second q-axis voltage command circuit 29 second d-axis voltage command circuit 30 integral gain calculation circuit 33 mode regeneration determination circuit 34 mode determination circuit 40 speed calculation circuit 41 d-axis proportional current control circuit 42 d-axis voltage drop calculation circuit

──────────────────────────────────────────────────続 き Continuing from the front page (72) Inventor Hiroyuki Tomita 7-1-1 Higashi Narashino, Narashino City, Chiba Prefecture Hitachi, Ltd. Industrial Equipment Division (72) Inventor Ryozo Masaki 7-1-1 Omikacho, Hitachi City, Ibaraki Prefecture No. 1 Hitachi, Ltd. Hitachi Laboratory (72) Inventor Yusuke Takamoto 1-1-1, Omika-cho, Hitachi City, Ibaraki Prefecture Hitachi, Ltd. Hitachi Laboratory (56) References JP-A-59-169369 (JP, A) JP-A-5-292777 (JP, A) JP-A-63-245291 (JP, A) JP-A-2-197284 (JP, A) JP-A-58-106610 (JP, A) JP-A-63 JP-A-6-250740 (JP, A) JP-A-4-289904 (JP, A) JP-A-4-109305 (JP, A) JP-A-2-96203 (JP, A) ) JP 9 74606 (JP, A) JitsuHiraku Akira 60-44103 (JP, U) (58 ) investigated the field (Int.Cl. ^7, DB name) H02P 5/408 - 5/412 H02P 7/628 - 7/632 H02P 21/00 G05B 1/00-7/04 G05B 11/00-11/60 G05B 13/00-13/04 G05B 17/00-17/02 G05B 21/00-21/02

Claims (1)

(57) [Claims] 1. An AC motor, a power converter for generating power to be supplied to the AC motor, and a d-axis current command for generating a magnetic flux of the AC motor from a torque command value to be generated by the AC motor. A current command value generator for calculating a value and a q-axis current command value orthogonal thereto, and calculating a d-axis voltage command value from a d-axis current deviation between the d-axis current command value and the d-axis current of the AC motor. Controlling the power converter.
D-axis current controller, the q-axis current command value and the AC
Q-axis voltage command value from q-axis current deviation from q-axis current of motor
In the AC motor control device and a q-axis current control unit calculates and controls the power conversion device, and the d-axis current controller and one of the q-axis current controller
Correction is made by adding the resistance voltage drop of the axis to the axis voltage command value calculated by one axis current control device of the other.
Shaft resistance calculating means is provided, and the one shaft current control device performs the voltage
And the other shaft current control device is proportional.
The voltage command value is calculated by integral calculation.
And the voltage command value by the one shaft current control device is
An AC motor control device characterized in that the AC motor control device is obtained from the shaft resistance calculating means .