JPH0646031A - Spread spectrum demodulator - Google Patents

Abstract

PURPOSE:To prevent time degradation in error rate characteristic of demodulated data with a small hardware scale configuration. CONSTITUTION:Inphase orthogonal base band signals 102 and 103 generated from a reception signal 101 by a complex base band signal generating means 1 are sampled by a sampling means 2 to obtain a sampled complex base band signal 104, and this signal 104 and a pseudo noise signal are subjected to mutual correlation operation by a complex correlation signal generating means 3 to obtain a complex correlation signal 105, and this signal 105 is subjected to polar coordinate transform to generate a phase signal theta and an amplitude signal R. A phase difference signal DELTAtheta is generated by a phase delay device 5 and a subtractor 6, and the signal R is multiplied by the delay amplitude signal by an amplitude delay device 7 and a multiplier 8 to obtain a weighting coefficient (c), and it is corrected in accordance with relations to a prescribed threshold (h) by a correcting circuit 9 to generate a corrected weighting coefficient (w). A weighted inphase base band signal 106 obtained by subjecting the signal DELTAthetaand the coefficient (w) to rectangular coordinate transform by a rectangular coordinate transforming means 10 is discriminated in every symbol period by a discriminating means 12 in accordance with the value of an actual number synthesis signal 107 accumulated by a signal synthesizing means 11 and is outputted as a demodulated data signal 108.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a spread spectrum demodulation device for improving the characteristic of demodulating a differentially encoded data signal from a direct spread modulation spread spectrum received signal.

[0002]

2. Description of the Related Art For example, literature (Yokoyama: spread spectrum communication system, Science and Technology Publishing Company, 1988 or Proaks: D
igital Communications, Second Edition, McGraw-Hil
1, 1989), the conventional spread spectrum demodulator has a complex baseband signal generating means 1 as shown in FIG.
Is a data signal that is differentially encoded into a carrier wave on the transmission side.
Phase-phase shift keying (PSK) modulation is performed, and the spread spectrum signal received signal 101 that is directly spread-modulated with a pseudo noise (PN) signal is in-phase baseband signal 102 in quadrature relationship with local carrier and quadrature baseband in quadrature relationship. Signal 103 to generate real and imaginary components of the complex baseband signal. The sampling means 2 includes the in-phase and quadrature baseband signals 10 from the complex baseband signal generation means 1.
2 and 103 are sampled for each chip cycle T _c (PN signal repetition cycle per chip) to generate a sampled complex baseband signal 104. The complex correlation signal generation means 3 performs cross-correlation calculation with the PN signal on the sampled complex baseband signal 104 from the sampling means 2 to generate a complex correlation signal 105. The coefficient weighting unit 13 delays the complex correlation signal 105 from the complex correlation signal generation unit 3 by a symbol period T _d (repetition period of the data signal. When the number of integer M chips is 2 or more, T _d = MT _c ), the delay complex. The weighting coefficient, which is the complex conjugate number of the correlation signal, is modified according to the magnitude relationship between the absolute square value and the predetermined threshold value h, and the modified weighting coefficient w is multiplied by the complex correlation signal 105 to obtain the weighted complex correlation signal 106a. To generate. The signal synthesizing means 11a cumulatively adds the weighted complex correlation signals 106a from the coefficient weighting means 13, and outputs the complex synthesized signal 1 for each T _d.
07a is generated. The data judging means 12a makes a judgment according to the value of the complex combined signal 107a from the signal combining means 11a and outputs it as a demodulated data signal 108.

The above-described spread spectrum demodulator of the prior art example uses a rake for performing weighted signal processing by complex operation.
e) method (a signal combining method that uses the characteristics of a spread spectrum signal that has been subjected to direct spread modulation to combine the energy of delayed waves generated in a selective fading channel).

As shown in FIG. 8, the complex baseband signal generating means 1 first shifts the phase of the in-phase local carrier or the local carrier from the oscillator 1c which oscillates the received signal 101 by the multiplier 1a by π / 2 radians. The orthogonal local carriers from the corresponding phase shifters 1d are respectively multiplied. Next, each high-frequency component is removed by the filter 1b, and the in-phase baseband signal 102 (baseband signal in phase with the local carrier) and the quadrature baseband signal 103 (quadrature with the local carrier as a complex baseband signal) are real components. Baseband signal) and

The sampling means 2 has a time kT _c (k is an integer).
In-phase and quadrature baseband signals y _pk and y _{qk sampled at}
To generate a sampled complex vector signal y _k 104. y _k = y _pk + jy _qk where j represents an imaginary unit

The complex correlation signal generating means 3 is operated at time (nM +
i) T _c = nT _d + iT _c (n is an integer, i = 0, ...
, M-1) sampled complex vector signals y _k 104 and P
A complex correlation signal z _{nM + i} 105 is generated by cross-correlation calculation with the N signal u _m (m = 1, ..., M). _{z nM + i = Σ m y} (n-1) M + i + m u m m = 1~M = P nM + i + jQ nM + i proviso _{P nM + i = Σ m y} p (n-1) M _{+ i + m u m m =} 1~M Q nM + i = Σ m y q (n-1) M + i + m u m m = 1~M or _{z nM + i = R nM +} i exp (jθ _{nM +} i) provided that the energy of the ^{_{R nM + i = (P 2}} nM + i + Q 2 nM + i) 1/2 θ nM + i = tan -1 (Q nM + i / P nM + i) complex correlation signal 105 Since | z _{nM + i} | ² is generally selected as a PN signal, a sequence in which the autocorrelation coefficient has an impulse shape, such as an M sequence, so that the preceding wave (reception of the shortest arrival time) is determined by the autocorrelation characteristic of the PN signal. Signal wave) and each delayed wave (received signal wave other than the preceding wave) are separated into energies, and a remarkable maximum appears at the arrival time of the preceding wave or the delayed wave.

As shown in FIG. 9, the coefficient weighting means 13 first outputs the complex conjugate number of the delayed complex correlation signal z obtained by delaying the complex correlation signal 105 by T _d by the delay device 7a as the weighting coefficient z ^{* in the} conjugate circuit 93. . Generally, the rake method uses a complex conjugate number of a complex correlation signal delayed by T _{d with} respect to an antipodal signal (two-phase PSK modulation signal or the like) as a weighting coefficient. Next, the absolute square value of the weighting coefficient z ^* input by the correction circuit 9a and the predetermined threshold h
The weighting coefficient z ^* is corrected according to the magnitude relationship with. Further, the complex correlation signal 105 is multiplied by the correction weighting coefficient w from the correction circuit 9a in the multiplier 8b to generate the weighted complex correlation signal 106a. Time (nM + i) T _c = nT _d +
The delayed complex correlation signal z _{(n-1) M + i} and the weighting coefficient z are added to iT _c.
^* _{(n-1) M + i} (* represents a complex conjugate) and a modified weighting coefficient w _{nM + i} 106a are generated. w _{nM + i} = z ^* _{(n-1) M + i} = R _{(n-1) M + i} exp (-jθ _{(n-1) M + i} ) (| z ^* _{(n-1) M + i} │ ² ≧ h) = 0 (| z ^* _{(n-1) M + i} │ ² <h) If the threshold value h is appropriately selected, only T _d can be applied to the preceding wave and the delayed wave. The time at which the energy of the weighting coefficient, which is the complex conjugate number of the delayed complex correlation signal | z ^* _{(n-1) M + i} | ² , does not exist, that is, it can be estimated that only the noise component exists in the delayed complex correlation signal. To the modified weighting factor w
_{nM + i} can be set to 0, and the reliability of the complex synthetic signal 107a generated by the signal synthesizing means 11a can be improved.

The signal synthesizing means 11a detects the time nMT _c = n.
The complex combined signal d _n-1 107a is generated by cumulative addition of each weighted complex correlation signal 106a to T _d, and path diversity (signal combining) by the rake method is realized. d _n-1 = Σ _k w _{(n-1) M + k} z _{(n-1) M + k} k = 0 to L (L is an integer of 1 or more and less than M) = Σ _k R _{(n-2) M + k} R _{(n-1) M + k} exp [j (θ _{(n-1) M + k} −θ _{(n-2) M + k} )] k = 0 to L

The data judging means 12 determines the time nMT _c = n.
The polarity of the complex composite signal d _n-1 107a of the preceding wave is determined at T _d, and the demodulation data a ^ _n-1 ε {-1,1} corresponding to the (n-1) th data signal a _n-1 is determined. To get The same can be applied when not only the preceding wave but also the delayed wave exists. d _n-1 = R _{(n-2) M} R _{(n-1) M} exp [j (θ _{(n-1) M} −θ _{(n-2) M} )] a ^ _n-1 = 1 (0 ≦ θ _{(n-1) M} -θ _{(n-2) M} <π / 2, or 3π / 2 ≤ θ _{(n-1) M} -θ _{(n-2) M} <2π, that is, Re [d _n-1 ] ≧ 0) = -1 (π / 2 ≤ θ _{(n-1) M} -θ _{(n-2) M} <3π / 2, that is, when Re [dn _-1 ] <0) where R _e [•] represents the real part of a complex number

[0010]

In the conventional spread spectrum demodulation device as described above, since the complex conjugate number of the complex correlation signal delayed by the symbol period is used as the weighting coefficient, signal processing by complex operation is required and hardware is required. Wear scale increases. In addition, when the carrier-to-noise power ratio of the received signal is low, the signal-to-noise power ratio of the complex correlation signal is also low and the noise error of the weighting coefficient is large, so that the accurately weighted signal cannot be synthesized and the error rate of the demodulated data There was a problem that the characteristics were deteriorated.

The problem to be solved by the present invention is to provide a rake system for performing weighted signal processing only by real number arithmetic so as not to deteriorate the error rate characteristic of demodulated data in a spread spectrum demodulation device having a small hardware scale. To do.

[0012]

In order to solve the above-mentioned problems, the spread spectrum demodulation device of the present invention uses a complex base band signal generating means to generate a real number and an imaginary number of a complex base band signal from a spread spectrum received signal which has been subjected to direct spread modulation. In-phase and quadrature baseband signals in-phase and in quadrature relationship with the local carrier are generated as components, and the complex-correlation signal generation means compares the sampled complex base-band signals sampled by the normalization means for each natural multiple of the chip speed. A composite signal in which a cross correlation calculation with a pseudo noise signal is performed to generate a complex correlation signal, the coefficient weighting means further performs coefficient weighting signal processing, and the generated weighting signals are cumulatively added by the signal combining means and output for each symbol period. According to the value of, the data determination means determines and outputs a demodulated data signal,
The coefficient weighting means is provided with the following means, and a rake method for performing weighted signal processing by real number calculation is adopted.

The polar coordinate transforming means polarizes the complex correlation signal from the complex correlation signal generating means to generate a phase signal and an amplitude signal.

The phase delay device and the subtracter subtract the phase signal from the polar coordinate conversion means with the phase signal delayed by the symbol period to generate a phase difference signal.

The amplitude delay device delays the amplitude signal from the polar coordinate conversion means by a symbol period to generate a delayed amplitude signal.

The multiplier multiplies the amplitude signal from the polar coordinate conversion means with the delayed amplitude signal from the amplitude delay device or the correction coefficient from the correction circuit to generate a weighting coefficient or a correction weighting coefficient, respectively.

The correction circuit comprises a weighting factor from the multiplier,
Or for the moving average weighting coefficient or the weighted average weighting coefficient from the moving average circuit or the weighted average circuit,
The correction weighting coefficient or the correction coefficient is generated according to the magnitude relationship with the predetermined threshold value.

The moving average circuit multiplies the weighting coefficient from the multiplier or the moving average value of the symbol interval N times (N is an integer of 2 or more) of the delayed amplitude signal from the amplitude delayer by N times to obtain the moving average weighting coefficient. To generate.

The weighted averaging circuit adds a weighting coefficient from the multiplier or a forgetting coefficient (0
A weighted average is applied to the weighted average weighting coefficient to generate a weighted average weighting coefficient.

The Cartesian coordinate transformation means performs Cartesian coordinate transformation on the phase difference signal from the subtractor and the modified weighting coefficient from the correction circuit or multiplier to generate a weighted signal.

[0021]

In the spread spectrum demodulation device of the present invention, polar coordinate conversion is first performed on the complex correlation signal generated from the spread spectrum received signal by the above means to generate a phase signal and an amplitude signal. Next, a phase difference signal is generated from the phase signal and the phase signal delayed by the symbol period. In addition, the product of the amplitude signal and the amplitude signal delayed by the symbol period is directly or moving averaged or weighted averaged, and is corrected according to the magnitude relationship with the predetermined threshold, or the amplitude delayed by the symbol period is used. The signal is subjected to moving average or weighted average to be corrected according to the magnitude relation with a predetermined threshold value, and is multiplied by the amplitude signal to generate a modified weighting coefficient. Further, Cartesian coordinate transformation is performed on the generated phase difference signal and the modified weighting coefficient to generate a weighting signal. The weighted signals are cumulatively added, determined for each symbol period, and output as a demodulated data signal.

[0022]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS As shown in FIG. 1, a spread spectrum demodulating apparatus according to an embodiment of the present invention has a complex baseband signal generating means 1, a sampling means 2 and a complex correlation signal generating means 3 as shown in FIG. Corresponding to. The polar coordinate transforming means 4 polarizes the complex correlation signal 105 from the complex correlation signal generating means 3 to generate a phase signal θ and an amplitude signal R. The phase delay device 5 and the subtracter 6 subtract the phase signal θ from the polar coordinate conversion means 4 with the phase signal delayed by the symbol period T _d to generate the phase difference signal Δθ. The amplitude delay unit 7, the multiplier 8, and the correction circuit 9 multiply the amplitude signal R from the polar coordinate conversion unit 4 by the amplitude signal delayed by T _d , and the generated weighting coefficient c is determined according to the magnitude relationship with the predetermined threshold value h. It is corrected and output as a modified weighting coefficient w. The Cartesian coordinate transformation means 10 performs Cartesian coordinate transformation on the phase difference signal Δθ from the subtractor 6 and the modified weighting coefficient w from the modification circuit 9 to generate a weighted in-phase baseband signal 106. The signal synthesizing means 11 is
The weighted in-phase baseband signal 106 is cumulatively added from the Cartesian coordinate transformation means 10, and the real number composite signal 107 is obtained for each T _d.
To generate. The data determining means 12 is the signal synthesizing means 11
Is determined according to the value of the real number composite signal 107 and output as a demodulated data signal 108.

The spread spectrum demodulation device of the above embodiment adopts a rake system in which weighted signal processing is performed by real number calculation.

The polar coordinate transformation means 4 is operated at time (nM + i) T.
_The phase signal θ _{nM + i} and the amplitude signal R _{nM + i} are generated by polar coordinate calculation of the complex correlation signal z _{nM + i} 105 at _c = nT _d + iT _c . θ _{nM + i} = tan ^-1 (Q _{nM + i} / P _{nM + i} ) R _{nM + i} = (P ² _{nM + i} + Q ² _{nM + i} ) ^1/2

The phase delay unit 5 and the subtractor 6 are operated at the time (nM +
i) The delayed phase signal θ _{(n-1) M + i} and the phase difference signal Δθ _{nM + i} are generated at T _c = nT _d + iT _c . Δθ _{nM + i} = θ _{nM + i} −θ _{(n-1) M + i}

The amplitude delay unit 7, the multiplier 8 and the correction circuit 9 are
Time _{(nM + i) T c =} nT d + delay iT _c amplitude signal _{R (n-1) M +} i and the weighting factor c _{nM + i} and correction weighting coefficient w
Generate _{nM + i} and. c _{nM + i} = R _{nM + i} R _{(n-1) M + i} w _{nM + i} = c _{nM + i} (when c _{nM + i} ≧ h) = 0 (when c _{nM + i} <h)

The signal synthesizing means 11 detects the time nMT _c = nT.
_A real number composite signal d _n-1 107 is generated by cumulative addition of each weighted in-phase baseband signal 106 to d to realize bus diversity by the rake method. d _n-1 = Σ _k w _{(n-1) M + k} cos Δθ _{(n-1) M + k} k = 0 to L

The data judging means 12 determines the time nMT _c = n.
The polarity of the real wave composite signal d _n-1 107 of the preceding wave is determined at T _d, and the demodulation data a ^ _n-1 ε {-1,1} corresponding to the (n-1) th data signal a _n-1 is determined. To get The same can be applied when not only the preceding wave but also the delayed wave exists. _{d n-1 = w (n} -1) M cosΔθ (n-1) M a ^ n-1 = 1 (0 ≦ Δθ (n-1) M <π / 2,3π / 2 ≦ Δθ (n-1 _{) M} <2π, that is, when d _n-1 ≧ 0) −1 (when π / 2 ≦ Δθ _{(n-1) M} <3π / 2, that is, d _n-1 <0) Complex as in the conventional example There is an effect that the same result as the conventional example can be obtained by a simple configuration of only real number multiplication, not multiplication.

In the above embodiment, the correction circuit 9 includes the multiplier 8
Although the weighting coefficient c is directly input, the moving average circuit 91 is provided as shown in FIG. 2 to multiply the moving average value of the weighting coefficient c N times (N is an integer of 2 or more) by N times. You may input as a moving average weighting coefficient v. The moving average weighting coefficient v can improve the signal-to-noise power ratio more than the weighting coefficient c. Time (nM + i) T
_The moving average weighting coefficient v _{nM + i} and the modified moving average weighting coefficient w _{nM + i} are generated by _c = nT _d + iT _c only by real number calculation. v _{nM + i} = Σ _k c _{(nk) M + i} k = 0 to N w _{nM + i} = v _{nM + i} (when v _{nM + i} ≧ h) = 0 (when v _{nM + i} <h)

Further, in the above embodiment, the correction circuit 9 is provided with the weighted average circuit 92 as shown in FIG. 3, and the weighted coefficient c from the multiplier 8 is weighted averaged by the forgetting coefficient (coefficient of 0 or more and less than 1) λ. You may input as an average weighting coefficient x. The weighted average weighting coefficient x can improve the signal-to-noise power ratio more than the weighting coefficient c, and does not diverge because the forgetting coefficient λ is 0 or more and less than 1. As shown in FIG. 4, the weighted average circuit 92 adds the weighting coefficient c to the delayed weighted average weighting coefficient obtained by multiplying the forgetting coefficient λ from the multiplier 92c by the adder 92a, and outputs the weighted average weighting coefficient x. The delayer 92b delays the weighted average weighting coefficient x from the adder 92a by T _d , and the delayed weighted average weighting coefficient is multiplied by the multiplier 92c.
Is multiplied by a predetermined forgetting factor λ (0 ≦ λ <1). Time _{(nM + i) T c =} nT d + iT c to produce a weighted average with weighting coefficients x _{nM + i} and modified weighted average weighting factor w _{nM + i} only real number operation. _{x nM + i = c nM +} i + λx (n-1) ( recurrence formula regarding _{x nM + i) M + i} x nM + i = Σ k λ k c (n + k) M + i k = -∞ ~ 0 w _{nM + i} = x _{nM + i} (when x _{nM + i} ≥h) = 0 (when x _{nM + i} <h)

In the above embodiment, as shown in FIG. 5, the amplitude signal c obtained by delaying the amplitude signal R from the polar coordinate converting means 4 by the amplitude delay device 7 by T _d is used as the moving average circuit 91 and the correction circuit 9 shown in FIG. And a modified moving average weighting coefficient w from a multiplier 8a for multiplying the moving average weighting coefficient v obtained by multiplying the moving average value N times the symbol interval by a correction coefficient u.
Alternatively, it may be input to the orthogonal coordinate conversion means 11. The moving average weighting coefficient v can improve the signal-to-noise power ratio compared to the delay amplitude signal c, and the correction coefficient u for v can be generated only by a real number calculation. Time _{(nM + i) T c =} nT d + iT c
To generate a delay amplitude signal c _{nM + i} , a moving average weighting coefficient v _{nM + i} , a correction coefficient u _{nM + i} for v, and a modified moving average weighting coefficient w _{nM + i} . c _{nM + i} = R _{(n-1) M + i} v _{nM + i} = Σ _k c _{(nk) M + i} k = 0 to N u _{nM + i} = v _{nM + i} (v ² _{nM + i} ≧ h = 0 (when v ² _{nM + i} <h) w _{nM + i} = R _{nM + i} u _{nM + i}

In the above embodiment, as shown in FIG. 6, the amplitude signal c obtained by delaying the amplitude signal R from the polar coordinate conversion means 4 by the amplitude delay device 7 by T _d is used as the weighted average circuit 92 and the correction circuit 9 shown in FIG. And a multiplier 8a for multiplying the correction coefficient u with respect to the weighted average weighting coefficient x weighted by the forgetting coefficient λ.
Therefore, the modified weighted average weighting coefficient w may be input to the orthogonal coordinate transformation means 11. Weighted average weighting coefficient x
Can improve the signal-to-noise power ratio over the delayed amplitude signal c,
Since the forgetting factor λ is set to 0 or more and less than 1, it does not diverge. Further, the correction coefficient u for x can be generated only by a real number operation. At time (nM + i) T _c = nT _d + iT _c , the delay amplitude signal c
_{nM + i} , a weighted average weighting coefficient x _{nM + i} , a correction coefficient u _{nM + i} for x, and a modified weighted average weighting coefficient w _{nM + i} are generated. _{c nM + i = R (n} -1) M + i x nM + i = c nM + i + λx (n-1) M + i ( recurrence formulas regarding _{x nM + i) x nM +} i = Σ k λ ^k c _{(n + k) + i} k = −∞ to 0 u _{nM + i} = x _{nM + i} (when x ² _{nM + i} ≧ h) = 0 (when x ² _{nM + i} <h) w _{nM + i} = R _{nM + i} u _{nM + i}

In the above embodiment, the carrier wave modulation method has been described as the two-phase PSK modulation method, but it goes without saying that another multi-phase PSK modulation method (four-phase PSK modulation method or the like) may be used. Further, although the sampling interval of the sampling means 2 is described as being equal to the chip period, it is 1 / K (K
Is a natural number) (for example, 1/2 of the chip period)
Or 1/4). That is, it may be a natural multiple of the chip speed.

[0034]

The spread spectrum demodulator of the present invention as described above adopts the rake system in which the weighted signal processing is performed by the real number operation, and therefore the hardware configuration can be made smaller than the conventional rake system by the complex operation. effective. Further, even when the carrier-to-noise power ratio of the received signal is low, a weighting coefficient with a small noise error can be obtained, so that there is an effect that accurate weighted signal synthesis can be performed and the error rate characteristic deterioration of demodulated data can be prevented.

[Brief description of drawings]

FIG. 1 is a functional block diagram of a spread spectrum demodulator according to an embodiment of the present invention.

FIG. 2 is a diagram illustrating addition of a moving average function of the correction circuit shown in FIG.

FIG. 3 is a diagram for explaining addition of a weighted average function of the correction circuit shown in FIG.

FIG. 4 is a functional block diagram of the weighted average circuit shown in FIG.

5 is a functional block diagram of another embodiment of the amplitude signal system shown in FIG.

6 is a functional block diagram of another embodiment of the amplitude signal system shown in FIG.

FIG. 7 is a functional block diagram of a conventional spread spectrum demodulation device.

FIG. 8 is a functional block diagram of the complex baseband signal generation means shown in FIG.

9 is a functional block diagram of the coefficient weighting unit shown in FIG. 7.

[Explanation of symbols]

 1 Complex Baseband Signal Generation Means 2 Sampling Means 3 Complex Correlation Signal Generation Means 4 Polar Coordinate Transformation Means 5 Phase Delayers 6 Subtractors 7 Amplitude Delayers 8, 8a Multipliers 9 Correction Circuits 91 Moving Average Circuits 92 Weighted Average Circuits 10 Orthogonal Coordinate transforming means 11 Signal combining means 12 Data judging means 101 Received signal 102 In-phase baseband signal 103 Quadrature baseband signal 104 Sampling complex baseband signal 105 Complex correlation signal 106 Weighted in-phase baseband signal 107 Real number composite signal 108 Multitone data signal In the drawings, the same reference numerals indicate the same or corresponding parts.

Claims (5)

[Claims] 1. An in-phase and quadrature baseband signal having an in-phase and quadrature relationship with a local carrier, which is generated as a real number and an imaginary number component of a complex baseband signal from a spread-spectrum received signal subjected to direct sequence modulation, at every natural multiple of the chip rate. A complex baseband signal generating means, a sampling means, a complex correlation signal generating means, and a complex correlation A coefficient weighting means for subjecting the complex correlation signal to the coefficient weighting signal processing from the signal generating means, and a weighting signal from the coefficient weighting means are cumulatively added and determined according to the value of the combined signal output for each symbol period, and a demodulated data signal is output. In the spread spectrum demodulating device including a signal synthesizing means and a data judging means, A polar coordinate conversion unit that performs a polar coordinate conversion on the complex correlation signal from the complex correlation signal generation unit by the number weighting unit to generate a phase and amplitude signal, and a phase signal obtained by delaying the phase signal by a symbol period from the polar coordinate conversion unit A phase delay unit and a subtracter that generate a phase difference signal, and a weighting coefficient obtained by multiplying the amplitude signal obtained by delaying the amplitude signal from the polar coordinate conversion unit by a symbol period are corrected according to the magnitude relationship with a predetermined threshold value. The amplitude delay unit, the multiplier, and the correction circuit that output as the modified weighting coefficient, the phase difference signal from the subtractor and the modified weighting coefficient from the modification circuit are subjected to Cartesian coordinate transformation to generate the weighted signal. A spread spectrum demodulation device, comprising: a rectangular coordinate transformation means for outputting to the signal synthesizing means. 2. A moving average circuit is provided in the correction circuit, and a moving average weighting coefficient obtained by multiplying a moving average value N times (N is an integer of 2 or more) of symbol intervals of the weighting coefficient from the multiplier is input to the correction circuit. The spread spectrum demodulation device according to claim 1, wherein 3. A weighted average circuit is provided in the correction circuit, and a weighted average weighting coefficient obtained by weighting the weighting coefficient with a forgetting coefficient (a coefficient of 0 or more and less than 1) is input from the multiplier to the correction circuit. The spread spectrum demodulation device according to claim 1. 4. A moving average circuit for multiplying a moving average value N times (N is an integer of 2 or more) of symbol intervals of an amplitude signal obtained by delaying the amplitude signal from the polar coordinate conversion means by a symbol period by an amplitude delayer by a moving average circuit. The average weighting coefficient is output to the orthogonal coordinate conversion means as a modified moving average weighting coefficient from a multiplier that multiplies the amplitude signal by the correction coefficient from the correction circuit that corrects the average weighting coefficient according to the magnitude relation with a predetermined threshold value. The spread spectrum demodulation device according to claim 1. 5. The forgetting factor (0) is added to the amplitude signal obtained by delaying the amplitude signal from the polar coordinate conversion means by a symbol period by an amplitude delay device.
A multiplier for multiplying the amplitude signal by a correction coefficient from a correction circuit that corrects a weighted average weighting coefficient from a weighted average circuit that performs a weighted average with a coefficient less than 1). 2. The spread spectrum demodulation device according to claim 1, wherein the modified weighted average weighting coefficient is output to the Cartesian coordinate transformation means.