JPH06245577A - Pwm control device of ac motor - Google Patents

Abstract

PURPOSE:To improve a low-frequency current carrying capacity of an inverter by alleviating the concentration of generation loss in a low frequency range to one element. CONSTITUTION:A PWM control device is provided with a voltage reference correction means 30. The voltage reference correction means respectively corrects three phase voltage reference signals Vu*, Vv* and Vw* using a voltage reference correction signal Vc having a polarity opposite to one voltage reference signal having the maximum phase, whereby new three phase voltage reference signals Vu**, Vv** and Vw** are produced. Thereby, a conduction ratio of a maximum current carrying element is reduced without affecting the control characteristics of a line voltage, and hence it is possible to improve a current carrying capacity of the element in a low output frequency range.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a PWM controller for an AC motor, and more particularly to a controller for calculating a three-phase voltage reference signal for zeroing the deviation of the speed feedback signal from the speed reference signal. The present invention relates to a PWM control device for an AC electric motor, which is provided with a PWM control circuit that PWM-controls an inverter based on a phase voltage reference signal, and controls the AC electric motor at a variable speed by the output of the inverter.

[0002]

2. Description of the Related Art Since a voltage type PWM inverter device can control both output voltage and output frequency with high precision and high speed by a PWM inverter, it is widely used for variable speed driving of an AC motor.

In this type of inverter device, when the output frequency of the inverter is continuously changed in a wide frequency range including direct current (0 Hz), the continuous energization time of a specific arm becomes long in the low frequency output region, so that the energization is performed. Losses were concentrated on the arm elements, and as a result, the current carrying capacity had to be reduced and used.

FIG. 11 shows an example of the configuration of a control unit 1A according to the prior art. The controller 20 included in the CPU unit 10 inputs the speed reference signal ω _r ^* , the speed feedback signal ω _r and the current feedback signal I, performs well-known speed control and vector control calculations, and outputs the voltage reference signal V ^* . . The voltage reference signal V ^* is converted into a PWM pulse train by the carrier signal CR in the well-known PWM control circuit 33 based on the principle of the comparator and outputs the gate pulse signal GP. The gate pulse signal GP drives and controls the AC electric motor via a voltage source inverter (not shown).

FIG. 12 shows a principle main circuit waveform when an inverter is driven by such a PWM control device. Here, as a typical case of driving in the low frequency region, which is a problem of the prior art, a case of direct current (0 Hz) output and maximum Z arm current will be described. The three-phase voltage reference signals V _u ^* , V _v ^* , V _w ^* are carrier signals C
Pulse train (gate pulse) V _{AK (U)} corresponding to R,
V _{AK (V)} and V _{AK (W)} are modulated, and the constituent elements of the inverter, such as the GTO element (large power transistor), are on / off controlled based on this gate pulse. In the case of DC (0 Hz) output, the induced voltage of the AC motor is almost zero, each voltage reference signal has a very small value corresponding to only the voltage drop, and the inverter main circuit 100 connected in a three-phase bridge type U, V, W, X, Y, which compose
Of the six convenient arms of Z, the so-called chopper operation is performed in which the positive two arms U and V and the negative one arm Z are fixed and energized. Set the U, V, Z arm voltages to V _{AK (U)} , V
_{AK (V)} , V _{AK (Z)} , the frequency of carrier signal CR is f
_c (therefore, the period of the carrier signal CR 1 / f _c)
As a result, FIG. 12 shows the waveforms of the current and voltage of each part defined as shown in FIG.

In FIG. 13, the inverter 100 is a GT.
It is assumed that the O element is configured in a three-phase bridge type, and a freewheeling diode is connected in antiparallel to each arm element. The currents flowing through the GTO elements of the U, V and Z arms are respectively represented by I _{GTO (U)} ,
I _{GTO (V)} and I _{GTO (Z),} and the currents flowing through the diodes of the W arm and the X and Y arms are I _{D (W)} and
I _{D (X)} and I _{D (Y),} and U and W of the AC motor 101.
The interphase voltage and the V and W interphase voltages are V _uw and V _vw , respectively, and the W phase line current is I _LW .

In the case of FIGS. 12 and 13, G of the negative Z arm
Unlike the case where the conduction current I _{GTO (Z) of the} TO element is maximum and the AC output is different, it is not alternately shared with the GTO element of the positive W arm every half cycle, and the waveform is as shown in the figure. It must be noted that it flows continuously.

FIG. 14 is a conceptual diagram of the energizing current I _am of one arm and the generated loss P _am (which is regarded as an equivalent sine wave by smoothing the switching operation). For AC output, only half the cycle needs to be shared. However, for example, when it becomes 0 Hz at time t ₁ , a current of π times the average value in the case of AC output is continuously supplied, resulting in loss. P _am also has a continuous heat generation waveform in a form corresponding to almost the current.

From the viewpoint of generated loss, 0 Hz is the most severe condition, but in the case of extremely low frequencies close to this, a large current reduction is necessary from the viewpoint of the temperature rise of the element as shown in FIGS. 15 and 16. Become. FIG. 15 shows a heat transfer model of the element and the cooler, which is represented by a cascade network of generated loss P, thermal resistance R _thx and thermal capacity C _thx (X = 1, 2, 3). T _j is the junction temperature of the element, T _c is the element case temperature, T _f is the cooler temperature, T
_a is the ambient air temperature. In practice, when the output frequency is sufficiently high with respect to the thermal time constant of the device (generally on the order of about 1 second), the loss P is averaged and the junction temperature of the device is averaged as shown in FIG. 16 (A). T _j and element case temperature T
_Although it appears as an increase in _c , the same figure (B), (C)
As the output frequency becomes lower, the temperature rise follows the loss P and greatly fluctuates, and the maximum reaching point of the junction temperature T _j of the element becomes higher.

FIG. 17 shows the generated loss P according to the output frequency f.
It is the figure which showed the example of _LOSS and the temperature rise _Tr of an element. The horizontal axis represents frequency f, and the vertical axis represents, in order from the top, the inverter output voltage V, the average conduction current I _{AV of} one element, and the element generated loss P.
_LOSS , current limit value I _LIM , and element temperature rise T _r are shown.

The conduction ratio of the GTO element is roughly expressed by the following equation.

(Conduction rate) = 1- {1- (modulation rate) × (load power factor)} / 2 (1) Here, the modulation rate is approximately represented by the following equation.

(Modulation rate) = K × f / f _B (2) where K: constant f: output frequency f _B : base frequency Below the base frequency f _B , the output frequency decreases and GT
Although the conduction loss of the O element is slightly smaller, the temperature rise of the element increases sharply at the frequency (f ₂ or less) at which it starts to interfere with the element thermal time constant described above, and the current concentrates near 0 Hz (f ₁ or less). The temperature rise in is significantly increased.

[0014]

Based on the above examination,
An object of the present invention is to provide a PWM control device capable of mitigating the situation of loss concentration on a specific element of an inverter in a low frequency region and avoiding reduction of the low frequency current carrying capacity of the inverter.

[0015]

In order to achieve the above object, the present invention provides a common voltage reference having a polarity opposite to that of a phase having a maximum absolute value among three-phase voltage reference signals in a low frequency output region. The present invention is characterized in that a voltage reference correction means for correcting each of the three-phase voltage reference signals using the correction signal to obtain a new three-phase voltage reference signal is provided.

[0016]

[Function] In the low frequency output region, the line voltage of the inverter output is corrected by correcting each phase voltage reference signal by the voltage reference correction signal of the polarity opposite to the polarity of the maximum amplitude phase among the voltage reference signals of each phase. It is possible to reduce the conduction ratio of the maximum current energizing arm and to disperse the generated loss without changing the current, thereby improving the energizing ability in the low frequency output region.

[0017]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 3 is an overall conceptual diagram showing a PWM control device according to the present invention together with a main circuit configuration. AC motor (M)
The control device 1 is provided for an inverter main circuit 100 that drives 101 at a variable speed. The control device 1 controls speed or torque. The speed of the AC motor 101 is detected by the speed detector (SS) 102 and introduced into the control device 1 as a speed feedback signal ω _r . The speed feedback signal ω _r is compared with a separately given speed reference signal ω _r ^*, and a gate pulse signal GP for making the deviation (speed deviation) zero is output to control the inverter 5. The motor current is detected by the current detector 7 and used as a current feedback signal I of a current control minor loop (not shown).

The detailed structure of the control device 1 is shown in FIG. The control device 1 outputs a manipulated variable (here, voltage reference signal V ^** ) obtained as a result of control calculation, and a PWM that modulates the voltage reference signal V ^** into a gate pulse train GP by a carrier signal CR. It comprises a control circuit 33.
The CPU unit 10 is actually composed of software, and is provided with a voltage reference correction unit 30, a mode determination unit 31, and a function generation unit 32, in addition to the controller 20 that performs speed control and vector control calculations.

The controller 20 has a well-known structure for performing calculations for speed control and vector control,
As shown in FIG. 2, a speed control unit 21 that inputs a speed reference signal ω _r ^* and a speed feedback signal ω _r and outputs a torque current reference signal τ ^* , and a voltage reference signal V ^* that is connected to the output side of the speed control unit 21 ^. And a vector control unit 22 for outputting the frequency reference signal F ^* . The vector control unit 22 inputs the velocity feedback signal ω _r and the torque current reference signal τ ^*, and receives the current reference vector I _dq ^* and the slip velocity reference signal ω _s.
^The vector operation unit 23 that outputs ^* , and the voltage reference vector V _dq for making the difference (current deviation) between the current feedback vector I _dq and the current reference vector I _dq ^* obtained by coordinate conversion from the current feedback signal I zero. Current control unit 24 that calculates ^*
And the phase reference signal θ _o ^* , the voltage reference vector V _dq
A coordinate converter 25 for coordinate-converting ^* into a voltage reference signal V ^* and converting a current feedback signal I into a current feedback vector I _dq . In addition to the above circuit parts, the vector control unit 22 adds the slip velocity reference signal ω _s ^* to the velocity feedback signal ω _r to obtain the primary angular frequency reference signal ω _o ^*. It includes means for integrating the frequency reference signal ω _o ^* to obtain the phase reference signal θ _o ^* , and means for obtaining the frequency reference signal F ^* from the primary angular frequency reference signal ω _o ^* .

Referring again to FIG. Three-phase voltage reference signal V output from the controller 20 having the above configuration
_{^{* (V * = (V u}} *, V v *, V w *)) is facilities in the voltage reference correction unit 30, for each energization mode, the correction as the current duty ratio of the maximum current supply arm is reduced And converted into a new voltage reference signal V ^** . That is, in the mode determining unit 31, the mode is switched at intervals of about 60 ° at the voltage reference phase, and the correction signal V _c of the polarity having the maximum amplitude phase and the reverse polarity of the three phases of the voltage reference signal V ^* is _supplied to each of the three phases. By adding to, the conduction ratio of the maximum current carrying arm is reduced without changing the line voltage, and the generated loss is dispersed. Function generator 32 is intended to determine the value of the voltage reference correction signal V _C is added frequency range to function the working voltage reference correction signal V _C, for a voltage reference correction the frequency reference signal F ^* as an input Voltage reference correction signal V _C
Is calculated and output.

Next, based on the voltage reference signal V ^* which is the output of the controller 20, the mode determining section 31 operates as shown in FIG.
The generation of the mode signal MD that is switched at intervals of approximately 60 ° will be described with the logical procedure shown in FIG. Voltage reference signal V ^*
Is actually a three-phase voltage reference signal V _u ^* , V _v ^* , V
_w ^* (see Figures 7 and 8)
By multiplying the voltage reference signals of the phases (step 50), a signal P _fl representing a sign that changes every approximately 60 ° in the phase of the voltage reference is obtained, and this signal P _fl is non-negative (P _fl ≧
It is classified into two cases of 0) and negative (P _fl <0) (step 51). If P _fl <0, check which one of the three-phase voltage reference signals has the smallest value (the one with the largest absolute value), set the mode signal MD for the U phase to MD = 1, and MD for the V phase. = 2, if W phase, MD = 3 (step 5)
2). Similarly, in the case of P _fl ≧ 0, which of the three-phase voltage reference signals has the maximum value is checked, and in the case of the U phase, MD = −
1, MD = -2 for V phase and MD = -3 for W phase (step 53).

As described above, since the continuous conduction time of one element becomes longer and the resulting loss, and hence the concentration of the temperature rise, is adversely affected in the low frequency region, the function generator 32 5 or 6 forms a voltage reference correction signal V _C as a function of the frequency reference signal F ^* obtained by the controller 20.

[0023] In Figure 5, the limit frequency -F _L, as F _L that are considered low-frequency ^{_{region, -F L ≦ F * ≦ F}} L
In the frequency range, F ^* ≦ -F _L or F ^* ≧ F _L
In this case, V _C = 0, but a voltage reference correction signal V _C that changes so as to increase in the form of a linear function as F ^* = 0 approaches is output.

In FIG. 6, the voltage reference correction signal V _C has a constant value in the frequency range regarded as the low frequency region, but the frequency reference signal F is taken into consideration from the viewpoint of control stability.
^* When increase of the have to have a hysteresis characteristic take different elapsed between the time of reduction, F _L2 <as F _L1, when increasing frequency reference signal F ^* is the voltage reference correction by -F _L2 ≦ F ^* ≦ F _L1 The signal V _C has a constant value, and when the frequency reference signal F ^* decreases, −F _L1 ≦ F ^* ≦ F _L2 and the voltage reference correction signal V _C has a constant value. In other frequency regions, V _C = 0.

Correction limit frequency F in FIGS. 5 to 6
_L to F _L1, F _L2 may be determined as a measure of the frequency f ₂ in FIG. 17.

The voltage reference correction unit 30 according to the first embodiment uses the voltage reference correction signal V _C according to the sign of the signal P _fl to correct the original voltage reference signal V ^* according to the following equation, and a new voltage reference is obtained. Generate signal V ^** . [P when the ^{_{fl ≧ 0] V u ** =}} V u * -V C ... (3) V v ** = V v * -V C ... (4) V w ** = V w * -V C ... (5) [when the ^{_{P fl <0] V u **}} = V u * + V C ... (6) V v ** = V v * + V C ... (7) V w ** = V w * + V C ... (8) FIG. 7 shows the contents of the above equation. The signal P _fl , which is obtained by multiplying the voltage reference signals V _u ^* , V _v ^* , and V _w ^* of each phase before correction, and which is alternately switched between positive and negative every 60 °, reverses the signal P _fl. The voltage reference correction signal V _C of the polarity is used as the voltage reference signals V _u ^* , V _v ^* , V _w.
^{* And} new voltage reference signals V _u ^** , V _v
^** , V _w ^** are generated. When such operation is applied, the phase voltage (signal V _u ^** ,
V _v ^** , V _w ^** ) are discontinuous, but the voltage reference correction signal V is used as the line voltage expressed as the difference between the phase voltages of the two phases.
Original voltage reference signals V _u ^* , V _v ^* , V even if _C is canceled
It is possible to obtain a continuous sinusoidal output voltage with a waveform corresponding to _w ^* .

By applying the above-described operation to the voltage reference signal V ^* , the conduction ratio of the maximum current conducting arm can be reduced as shown in FIG. FIG. 9 is shown correspondingly to FIGS. 12 and 13 shown as the prior art, in which DC (0 Hz) output and Z arm current maximum, P _fl <0, M
The principle main circuit waveform at D = 3 is shown. In addition,
Here, each signal is used according to FIGS. 12 and 13. The voltage reference signal V ^* is biased and modulated on the positive side by the voltage reference correction signal V _C , and as a result, FIG.
In comparison with the case of, the energizing current I of the Z-arm GTO element
It can be seen that _{GTO (Z)} has almost the same size and the flow period is much shorter.

With the above arrangement, as shown in FIG. 10, the generated loss P _LOSS in the low frequency region is _increased under the same output voltage V and 1 element average conduction current I _AV as in the case of FIG. The concentration is alleviated, the maximum value of the element temperature rise _Tr is suppressed to a low value, and the low-frequency current carrying capacity of the inverter device as a whole can be improved.

Next, a second embodiment of the present invention will be described.

The mode determining unit 31 shown in FIG. 1 obtains the mode signal MD that switches every 60 ° in the procedure shown in FIG. 4 in the same manner as described above. The voltage reference correction unit 30 follows the mode signal MD obtained by the mode determination unit 31, and determines the three-phase voltage reference signal V _u ^* ,
For the phase having the maximum absolute value of V _v ^* and V _w ^* , a voltage reference correction signal V _{C having a} polarity opposite to that of the voltage reference signal.
(Or −V _C ), and for other phases, a voltage reference correction signal V _{C having a} polarity opposite to that of the voltage reference signal of that phase.
(Or -V _C ) is corrected by adding the voltage reference signal of the phase minus the voltage reference signal of the phase having the maximum absolute value, and the new voltage reference signals V _u ^** , V _v ^** are added. ，
Form V _w ^** . For example, the U-phase voltage reference signal V _u ^*
The maximum absolute value of (MD = -1 or MD = 1
Case) is as follows. [In the case of ^{_{MD = -1] V u ** =}} -V C ... (9) V v ** = -V C + (V v * -V u *) ... (10) V w ** = -V C ^{_{+ (V w * -V u *}} ) ... (11) [ in the case of ^{_{MD = 1] V u ** =}} V C ... (12) V v ** = V C + (V v * -V u *) ... (13) in ^{_{V w ** = V C + (}} V w * -V u *) ... (14) the above equation, equation (10), (11), (13), the second term on the right-hand side of (14) It should be noted that each is a term expressing the line voltage using the phase voltage signal and has a sinusoidal waveform. Therefore, also in this case, the phase voltages (signals V _u ^** , V _v
^** , V _w ^** ) becomes discontinuous, but as the line voltage represented as the difference between the phase voltages of the two phases, the original voltage reference signal V _u ^* , even if the voltage reference correction signal V _C is canceled V _v ^* , V _w ^*
It is possible to obtain a continuous sine wave output voltage having a waveform corresponding to.

The contents of the above equation are illustrated in FIG. The voltage reference signals V _u ^* , V _v ^* , and V _w ^* are converted into new voltage reference signals V _u ^** , V _v ^** , and V _w ^** , respectively, by the mode signal MD that switches every approximately 60 °. There is.
Even if operated in this manner, the output phase voltage (reference voltage signal) becomes discontinuous as shown in the figure, but as described above, a continuous sine wave voltage can be obtained as the line voltage.

As described above, the conduction ratio of the maximum current conducting arm is reduced as in the first embodiment, and as shown in FIG.
_{It is possible} to avoid the concentration of _LOSS , and consequently the element temperature rise _Tr , and improve the low-frequency current carrying capacity of the inverter device as a whole.

Although the inverter 5 is composed of the GTO element for the sake of convenience in the above description, the present invention can be applied to any PWM inverter as long as it is a PWM inverter using a self-turn-off switching element. is there.

[0034]

As described above, according to the present invention, P
In the WM-controlled inverter, the conduction ratio of the maximum current conducting element in the low frequency region is reduced without affecting the line voltage, that is, without affecting the control characteristics, and thereby the generated loss. The situation in which one element is concentrated is greatly alleviated, and the low-frequency current carrying capacity of the inverter device can be greatly improved.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of a PWM control device according to the present invention.

FIG. 2 is a block diagram showing a configuration example of a controller inside a control device according to the present invention.

FIG. 3 is a connection diagram showing a relationship between a control device according to the present invention and a main circuit.

FIG. 4 is a flowchart showing a mode determination procedure of a mode determination unit in FIG.

5 is a characteristic diagram showing a first example of a function of a function generator in FIG.

FIG. 6 is a characteristic diagram showing a second function example of the function generating section in FIG.

FIG. 7 is an explanatory view for explaining the operation of the first embodiment of the voltage reference correction unit in FIG.

FIG. 8 is an explanatory view for explaining the operation of the second embodiment of the voltage reference correction unit in FIG.

FIG. 9 is an operation waveform diagram of each part of the main circuit for explaining the operation and effect of the present invention.

FIG. 10 is a characteristic diagram with respect to an output frequency for explaining the operation and effect of the present invention.

FIG. 11 is a block diagram showing an outline of a general PWM control device according to a conventional technique.

FIG. 12 is an operation waveform diagram of each part of the main circuit corresponding to FIG. 9 for explaining the operation of the conventional device.

FIG. 13 is a main circuit connection diagram shown for explaining the reference numerals in FIG.

FIG. 14 is a conceptual waveform diagram for explaining the relationship between the current flow mode of one arm and the generated loss.

FIG. 15 is an equivalent circuit diagram showing a heat transfer circuit model of an arm element.

FIG. 16 is an explanatory diagram showing the difference between the generated loss and the temperature rise of the element due to the difference in frequency.

FIG. 17 shows various characteristics of a conventional device with respect to output frequency.
The figure shown corresponding to 0.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Control device 10 CPU part 20 Controller 30 Voltage reference correction part 31 Mode determination part 32 Function generation part 33 PWM control circuit 100 Inverter main circuit 101 AC electric motor 102 Speed detector

Claims (7)

[Claims] 1. A controller for calculating a three-phase voltage reference signal for zeroing the deviation of the speed feedback signal from the speed reference signal, and an inverter PW based on the three-phase voltage reference signal.
A PW of an AC motor, which is provided with a PWM control circuit for M control and which controls the AC motor at a variable speed by the output of an inverter.
In the M control device, in the low frequency output region, the three-phase voltage reference signals are respectively corrected by using the common voltage reference correction signal having the polarity opposite to the polarity of the phase having the maximum absolute value among the three-phase voltage reference signals. A PWM control device for an AC motor, characterized by comprising voltage reference correction means for providing a three-phase voltage reference signal. 2. The control device according to claim 1, wherein the voltage reference correction means is a common voltage having a polarity opposite to a polarity of a phase having a maximum absolute value among three-phase voltage reference signals in a low frequency output region. A PWM control device for an AC motor, wherein a new three-phase voltage reference signal is obtained by adding a reference correction signal to each of the three-phase voltage reference signals. 3. The control device according to claim 1, wherein the voltage reference correction means sets a polarity of the voltage reference signal for a phase having a maximum absolute value among the three-phase voltage reference signals in a low frequency output region. Is replaced with a voltage reference correction signal of the opposite polarity, and for other phases, a voltage reference correction signal of the opposite polarity to the polarity of the voltage reference signal of that phase is used. A PWM control device for an AC motor, which is a new three-phase voltage reference signal by adding a voltage reference signal. 4. The control device according to claim 1, wherein a three-phase voltage reference signal is input to the voltage reference correction means, and a phase having a maximum absolute value that changes every 60 ° is set. A PWM control device for an AC motor, characterized in that a mode determining means for determining its polarity and outputting a mode signal is additionally provided. 5. The control device according to claim 1, wherein the speed reference signal is input to the voltage reference correction means, and a voltage reference correction signal according to a predetermined function in a low frequency output region is supplied. A PWM control device for an AC motor, further comprising a function generating means for outputting. 6. The PWM control device for an AC motor according to claim 5, wherein the function generating means outputs a voltage reference correction signal which changes in the form of a substantially linear function in the low frequency output region. 7. The PW of an AC motor according to claim 5, wherein the function generating means outputs a voltage reference correction signal having a constant value in a low frequency output region.
M control device.