JPH061903B2 - Signal transmission device - Google Patents

Description

Description: TECHNICAL FIELD The present invention relates to a signal transmission device for transmitting a signal obtained by digitizing an analog signal such as a PCM signal by dividing it into blocks of a certain number of words, and in particular, each block. The present invention relates to a signal transmission device that reduces the transmission bit rate by performing companding processing every time.

[Conventional technology]

In recent years, analog audio signals, video signals, etc. are often sampled (quantized) for quantization and coding processing, and are often transmitted or recorded / reproduced as so-called PCM (pulse code modulation) signals. There is.

When transmitting or recording / reproducing such a PCM signal, the sampling frequency f _s is set in order to obtain a band of about 20 kHz and an S / N ratio of about 90 dB or more.
A linear quantization of 44.1 kHz and 1 word 16 bits is generally adopted, but the transmission rate in this case is 700K.
It is extremely high, reaching BPS (700 Kbits per second) or more.

By the way, in a digital signal obtained by A / D converting an analog signal such as an audio signal or a video signal as described above, the statistical property thereof is biased, and a portion of low importance in view of an audiovisual phenomenon. It is possible to compress the amount of information by utilizing the fact that there is, for example, difference / union processing and compression / expansion processing (companding processing).
It is known that signal quality deterioration is extremely small even if the above is performed.

In consideration of such a point, the applicant of the present invention first divides a digital PCM signal into blocks, for example, in a fixed time unit or in a fixed number of words, and performs prediction processing such as difference processing and companding for each block. It is disclosed in Japanese Patent Application No. Sho 58-97 that processing, transmission or recording / reproduction is performed.
687-9, Japanese Patent Application No. 58-163054, Japanese Patent Application No. 5
No. 8-166267 or Japanese Patent Application No. 58-210382
No., etc.

At least one for each block in these techniques
Word reference data, for example, straight PCM data, is provided, and based on this reference data, by performing arithmetic processing such as sequentially adding difference data, for example, all the original sampling data (straight PCM data) in the block can be obtained. Restorable. This is mainly to minimize the error propagation that occurs when transmitting differential processing data, and also to prevent errors such as overshoot at block boundaries when switching prediction processing and companding processing for each block. This is to prevent it.

[Problems to be solved by the invention]

By the way, when the bit compression rate is extremely high, for example, when compressing 1 word 16-bit data to 1 word 4 bits, 1 word of the reference data is converted into 4 words of the compressed data. Therefore, in order to improve the transmission efficiency or the recording efficiency, it is desirable to transmit or record / reproduce without providing the reference data in the block.

However, if only the differential output is to be transmitted, then S
/ N improvement is insufficient or S for high frequency input signals
There is a problem that / N is deteriorated, or abnormal noise occurs when a code error occurs on the transmission path or during recording / reproducing.

In view of such circumstances, the present invention can extremely reduce the transmission bit rate, can compensate for the lack of the S / N improvement amount, and can suppress the S / N deterioration at the time of inputting a high frequency signal and the occurrence of a code error. An object of the present invention is to provide a signal transmission device capable of preventing generation of abnormal noise.

[Means for Solving the Problems] In order to solve the above-mentioned problems, the signal transmission device of the present invention divides the input digital signal into blocks every constant word number along the time axis, and outputs the signal of each block. A plurality of difference processing filters that are transmitted through a filter for obtaining a prediction error, that is, a so-called difference processing filter, and that use N-order predictors and N-order or less predictors as the difference processing filters. The maximum absolute value (peak value) or the maximum absolute value (peak value) in the block of the output from each of these difference processing filters is multiplied by a coefficient and compared with each other, and the value is determined to be the minimum value. The difference processing filter is selected for the block.

Here, the plurality of difference processing filters may include a filter (flat path) that outputs the input digital signal as it is.

[Action]

A difference processing filter that outputs the smallest peak value of the respective peak values (maximum absolute value within a block) of each prediction error output for one block from the plurality of difference processing filters outputs the input digital signal within the block. This is the most efficient differential processing, and highly efficient transmission can be performed by selecting this differential processing filter. Also, by multiplying each peak value in the block by a predetermined coefficient according to the order of the difference processing filter (so-called weighting) and comparing, a low-order filter and an input signal are compared with a high-order filter. It is possible to preferentially or more frequently select a filter (flat path) to be output as it is, improve S / N deterioration at the time of inputting a high frequency signal, and prevent abnormal noise when a code error occurs.

〔Example〕

Schematic Configuration First, the overall schematic configuration of an audio bit rate reduction system as an example of a signal transmission device to which the present invention is applied will be described with reference to FIG.

The system shown in FIG. 1 comprises an encoder 10 on the transmitting side (or recording side) and a decoder 30 on the receiving side (or reproducing side). An analog audio signal having a frequency f is input to an input terminal 11 of the encoder 10. _An audio PCM signal x (n) obtained by sampling at _s , quantizing and encoding is supplied. This input signal x
(n) is sent to the predictor 12 and the adder 13, respectively, and is the prediction signal from the predictor 12. Is sent to the adder 13 as a subtraction signal. Therefore, in the adder 13, the predicted signal is calculated from the input signal x (n). Is subtracted to obtain a prediction error signal or (broadly defined) difference output d (n), that is, Is output.

Here, the predictor 12 generally uses the past p inputs x (n
-P), x (n-p + 1), ..., x (n-1) primary combination predicted value To calculate However, α _k (k = 1, 2, ... P) is a coefficient. Therefore, the prediction error output or (broadly defined) difference output d (n) is Can be expressed as

Further, according to the present invention, the input digital signal is divided into data within a fixed time, that is, every fixed word number l of the input data, and the coefficient α _k is set so that an optimum prediction filter characteristic is obtained for each block. Is selected. As will be described later, this can be regarded as providing a plurality of filters for obtaining a difference output (prediction error output) including a predictor having different characteristics or an adder. The optimum filter among the processing filters is selected for each block. The selection of this optimum filter is performed by the prediction / range adaptation circuit 21 using the maximum absolute value (peak value) in the block of the output from each of the plurality of difference processing filters or the value obtained by multiplying the maximum absolute value (peak value) by a coefficient. This is performed by comparing, and specifically, the difference processing filter that minimizes the maximum absolute value (or its coefficient multiplication value) is selected as the optimum filter for the block. The optimum filter selection information at this time is output from the prediction / range adaptation circuit 21 as mode selection information and sent to the predictor 12.

Next, the difference output d (n) as the prediction error is sent to the bit compressing means including the shifter 15 of the gain G and the quantizer 16 via the adder 14, and, for example, a floating point (floating point). In the display form, the exponent part is subjected to the gain G, and the mantissa part is subjected to the compression process or the ranging process so as to correspond to the output from the quantizer 16, respectively. That is, the shifter 15 switches the so-called range by shifting the digital binary data by the number of bits corresponding to the gain G (arithmetic shift), and the quantizer 16 changes the range of the bit-shifted data. Requantization is performed so as to extract a certain number of bits. Next, the noise shaping circuit (noise shaper) 17 obtains an error amount between the output and the input of the quantizer 16, that is, a so-called quantization error, by the adder 18, and the quantization error of the gain G ^-1 is obtained. The so-called error feedback is carried out by sending the prediction signal of the quantization error to the adder 14 as an existing signal by sending it to the predictor 20 via the shifter 19. At this time, the prediction / range adaptation circuit 21 outputs the range information based on the maximum absolute value in the block of the difference output from the filter of the selected mode, and sends this range information to the shifters 15 and 19 for each block. The gains G and G ^-1 are determined. Further, the predictor 20 is adapted to determine its characteristics by sending the mode information from the prediction / range adaptation circuit 21.

Therefore, the output d '(n) from the adder 14 is obtained by subtracting the prediction signal (n) of the quantization error from the noise shaper 17 from the difference output d (n) d' (n) = d (n )-(N) ... And the output d ″ (n) from the gain G shifter is d ″ (n) = G · d ′ (n). Also, the output from the quantizer 16 Let e (n) be the quantization error in the quantization process, Therefore, the quantization error e (n) is taken out by the adder 18 of the noise shaper 17, and the shift 1 of the gain G ^-1 is obtained.
Predictor 2 that takes a linear combination of past r inputs via 9
The prediction signal (n) of the quantization error obtained through 0 is Becomes This equation has the same form as the above equation, and the predictors 12 and 20 are FIR (finite impulse response) filter.

From these expressions, the output from the quantizer 16 Is Substituting the above formula into d (n) of this formula, And this output Are taken out through the output terminal 22. Where above Z transformation of Then, Becomes

The range information from the prediction / range adaptation circuit 21 is taken out from the output terminal 23, and the mode selection information is taken out from the output terminal 24.

Next, the output from the output terminal 22 of the encoder 10 is input to the input terminal 31 of the decoder 30 on the receiving side or the reproducing side. The signal obtained by transmitting, recording, or reproducing Is being supplied. This input signal Is sent to the adder 33 via the shifter 32 having a gain G ⁻¹ . Output from adder 33 Is sent to the predictor 34 and becomes the prediction signal (n), and this prediction signal (n) is sent to the adder 33 and is sent to the shifter 32.
Output from Is added. This added output is the decoded output Is output from the output terminal 35.

The range information and the mode selection information transmitted or recorded / reproduced from the output terminals 23 and 24 of the encoder 10 are input to the input terminals 36 and 37 of the decoder 30, respectively. Then, the range information from the input terminal 36 is sent to the shifter 32 to determine the gain G ⁻¹ , and the mode selection information from the input terminal 37 is sent to the predictor 34 to determine the prediction characteristic. The prediction characteristic of the predictor 34 is selected to be equal to the characteristic of the predictor 12 of the encoder 10.

In the decoder 30 having such a configuration, the shift Becomes Here, since the predictor 34 selects a characteristic equal to that of the predictor 12 of the encoder 10, Therefore, from the formula, Therefore, Becomes Here, assuming that there is no error in the transmission path or recording medium, Then, from the above formula and formula, Becomes

From this equation, it is clear that the noise reduction effect of G ⁻¹ is obtained with respect to the quantization error E (z). At this time, if the spectral distribution of noise appearing in the decoder output is N (z), Becomes

Further, in such a system, the G acts to normalize with a value related to the maximum absolute value in the block, but the G has a frequency characteristic. Here, in order to simplify the description, the above G is represented as a product of two elements G _p and G _f such as G = G _p · G _f . Of these two elements, G _p means the prediction gain, that is, the improvement amount of the instantaneous S / N by the prediction filter processing,
G _f means the amount of gain control by the above ranging processing, that is, the amount of expansion of the dynamic range. Therefore, G _p depends on the input signal frequency and does not depend on the input signal level, whereas G _f does not depend on the input signal frequency and depends on the input signal level. Also, G _p is Has a S / N improvement amount of, and has a frequency characteristic that is an inverse function of the transfer function 1-P (z) of the difference processing filter for obtaining the prediction error. Becomes like the above formula. Regarding G _f , it corresponds to quasi-instantaneous companding which is normalized by the maximum absolute value in the block in the mode selected for each block.

In a system having such a schematic configuration, the gist of the present invention is to use an Nth-order predictor and Nth order or less as a difference processing filter for obtaining a prediction error output or a difference output including a predictor 12 and an adder 13. , A plurality of filters using the predictor of each of the filters are provided, and the maximum absolute value in the block of the output from each filter or the product of the maximum absolute value multiplied by a coefficient is compared with each other, and the difference processing that minimizes the value is performed. That is, the filter is selected. Among the plurality of difference processing filters, a filter that directly outputs straight PCM data is also included.

Specific Configuration Example Next, FIG. 2 shows a more specific configuration example of the encoder 10 of the audio bit rate reduction system shown in FIG. Have the same reference numbers.

In FIG. 2, as the predictor 12, a plurality of, for example, four predictors 12A, 12B, 12C, 12 are used.
D is provided. Predicted outputs from these predictors 12A to 12D are respectively added by adders 13A as subtraction signals.
~ 13D and subtracted from the original input signal. That is, four predictors 12A to 12D and four adders 13A to 13D respectively form four difference processing filters for obtaining prediction errors of four systems. Here, each of the predictors 12A to 12D apparently has a quadratic configuration, and the predictor 12A uses the coefficients α ₁ and α ₂ as the coefficients.
K ₁ , K ₂ , 12B are K ₃ , K ₄ , 12C are K ₅ , K ₆ , 12D
Although K ₇ and K ₈ are provided, a prediction characteristic of a first order or lower can be obtained by setting 0 of at least one coefficient of a desired predictor. Therefore, although the four difference processing filters are apparently configured to take the second-order differences, one of them is used for the desired difference processing filter.
It is possible to obtain the characteristics that take the difference below (including those that output straight PCM data).

The output from each of the difference processing filters, that is, the output from each of the adders 13A to 13D is the l-word delay circuits 41A to 41D and the maximum absolute value hold circuit 4, respectively.
2A to 42D, the 1-word delay circuit 41A
Each output from ~ 41D is a mode changeover switch circuit 43.
Of the selected terminals a to d. That is, since one block has 1 word, the 1-word delay circuit 4
In 1A to 41D, a delay of 1 block is performed,
Each maximum absolute value (peak) during this delay
The maximum absolute value in the block is detected in the hold circuits 42A to 42D. The maximum absolute values in these blocks are sent to the prediction / range adaptation circuit 21 and compared with each other, and the one having the minimum value is selected.
At this time, the maximum absolute value in each block may be multiplied by a predetermined coefficient to perform so-called weighting and then compared. The prediction / range adaptation circuit 21 outputs mode selection information for selecting one block of data from which the minimum in-block maximum absolute value is obtained among the one block of data from each of the difference processing filters. Then, by transmitting the mode selection information to the changeover switch circuit 43, the changeover connection to the delay circuit for outputting the selected one block of data is performed. The output from the changeover switch circuit 43 is sent to the adder 14. Also,
The mode selection information from the prediction / range adaptation circuit 21 is also sent to the predictor 20 and the output terminal 24.
Here, the predictor 20 selects one having a characteristic equal to the selected one of the predictors 12A to 12D in order to make noise (see the above equation) appearing in the decoder output white, for example. That is, the predictor 20 also apparently has a quadratic predictor configuration, and the coefficients Ka and Kb corresponding to the coefficients β ₁ and β ₂ are the coefficient sets K ₁ and K ₂ of the predictors 12A to 12D. It is selected equal to the coefficient predictor of differential processing filter specified by the mode selection of the ~K _7, K _8.

Further, in the specific example of the decoder 30 shown in FIG. 3, the predictor 34 has an apparent quadratic structure corresponding to the predictors 12A to 12D in FIG. 2, and each coefficient Kc,
Kd is a set of coefficients of the predictors 12A to 12D.
Any one of K ₁ , K _{2 to} K ₇ , K ₈ is selected according to the mode selection information from the input terminal 37.

The other configurations of FIGS. 2 and 3 are the same as those of FIG. 1 described above, and thus the description thereof will be omitted.

It should be noted that the encoder 10 having the specific configuration as described above
As a hardware configuration of the decoder 30 and the decoder 30, for example, it is not necessary to actually provide a plurality of predictors 12A to 12D and the like, and the coefficient of one predictor may be used by switching in a time division manner. It is needless to say that the entire 10 and the decoder 30 can be realized by software by a system including a DSP (digital signal processor) and a memory.

First Embodiment Here, as the first embodiment of the present invention, the sampling frequency f _s of the audio PCM signal supplied to the input terminal 11 is set to 18.9 kHz, and the word number l of one block is 16 bits per word. The case where the number of words is 28 and the quantizer 16 requantizes one word into four bits of data will be described in detail. The secondary predictor 12 at this time
Examples of the coefficient sets K ₁ , K _{2 to} K ₇ , K ₈ of A to 12D are, for example, 12A: K ₁ = 1.8426, K ₂ = −0.8649 12B: K ₃ = 0.875, K ₄ = 012C: K _{5 = 1.5155, K 6 = -0.81} 12D: preset as _{K 7 = 0, K 8 =} 0. The transfer function 1-P (z) of the differential processing filter in each mode at this time is: A: 1-1.8426z ^-1 + 0.8649z ^-2 B: 1-0.875z ^-1 C: 1-1.5155z ^-1 + 0.81z ^-2 D: 1, and the frequency characteristics of each of these difference processing filters are
The curves A to D in FIG. 4 are obtained.

That is, the difference processing filter (comprising the predictor 12A and the adder 13A) corresponding to the characteristic curve A is the secondary difference PC.
It is a filter corresponding to the M mode, and has a large improvement amount in the low-frequency prediction gain, that is, the instantaneous S / N. The difference processing filter (the predictor 12B and the adder 1 corresponding to the characteristic curve B
3B) is a filter corresponding to the first-order difference PMC mode, and the difference processing filter (the predictor 12C and the adder 13C) corresponding to the characteristic curve C has a large prediction gain in the middle band. Differential processing filter consisting of predictor 12D and the adder 13D, by the coefficient K _7, K ₈ are both 0, as shown in a characteristic curve D of FIG. 4, the reference gain 1 without a frequency characteristic so-called It has a mere flat path characteristic and is equivalent to the straight PCM mode.

FIG. 5 shows a specific example of the word structure transmitted for each block. From re-quantized 1 word 4 bits, 28 words of audio data words W _{0 to} W ₂₇ and 1 word 16 bits are shown. mode selection information indicating 4 so-called range information word W _R that indicates the 4 bits of the take-out position when the re-quantization for each block to the bit, either has been selected out of four modes corresponding to the four filters a word W _M is transmitted block by block. Therefore,
The average number of bits per transmitted audio data word is (4 × 28 + 4 + 2) ÷ 28≈4.214 [bit].

In FIG. 4, when a single sine wave signal is input, the filter having the characteristic curve A is used when the input signal frequency is 0 to f ₁ and the characteristic curve C is used when the frequency is f ₁ to f ₂ . A filter and a filter having a characteristic curve D are selected for frequencies up to f ₂ to f _s / 2. It is needless to say that the frequency f _s / 2 or more of the input signal is previously removed by an LPF (low pass filter) before A / D conversion in order to prevent so-called aliasing. The frequency response of each filter selected in this way becomes the predicted gain at that frequency, that is, the improvement amount of the instantaneous S / N, and the improvement amount of the instantaneous S / N with respect to the frequency is as shown by the hatched portion in FIG.

However, since the actual audio input signal is a composite spectrum signal, the selection based on the above-described clear boundary is not performed, and the filter of the characteristic curve B is also used relatively frequently.

Further, the above-mentioned filter selection is performed when the maximum absolute value (peak value) in the block from the filter of each mode is directly compared, but a predetermined coefficient is set for each peak value in the block of each mode. Low-order filter or straight PC by multiplying by and comparing
You may make it select more filters which output M data. In this case, as an example of the coefficient, the peak value of the second-order differential PCM data from the filter of the characteristic A is multiplied by the coefficient 2, and the peak value of the straight PCM data from the filter of the characteristic D is multiplied by the coefficient 0.7. Then, the peak value of the data from the other filter is multiplied by the coefficient 1 (or not multiplied by the coefficient), and the respective values are compared with each other.
Select a filter that minimizes its value. The multiplication of such a coefficient corresponds to translating the corresponding characteristic curve in the graph of FIG. 4 in the vertical axis direction (changing the response value). For example, the peak value from the filter of the characteristic curve A is calculated as a coefficient. Multiplying by 2 gives the characteristic curve A a fourth
This corresponds to a parallel translation of about 6 dB upward in the figure.
Therefore, as a result, the filter switching frequencies f ₁ and f ₂ are shifted to the low frequency side, the characteristic C filter is more frequent than the characteristic A filter, and the characteristic C filter is more frequent than the characteristic C filter. The filter of characteristic D will be selected more often. By thus increasing the frequency of use of the low-order filter, it is possible to suppress the influence of code errors on the transmission line.

In the decoder side, the audio data words W ₀ to _W-27 of one block which is input to the contrary to the expansion processing or the encoder side in block units by the shifter 32 based on the data of the range information word W _R ranging process is performed, the filter next made of the adder 33 and predictor 34, inverse predictive processing in blocks on the basis of the data of the mode selection information word W _M is performed, the original straight PCM data is restored It

According to the system of the first embodiment, it is possible to transmit a sound quality of low to medium fidelity, and a normal voice signal or the like is transmitted at an extremely low bit rate (4.214 bits per word, 1
Transmission bit rate per channel About 79.6 kbit /
Seconds) can be transmitted.

Second Embodiment Next, as an example of a system capable of transmitting a music signal with sound quality of middle to high fidelity (middle to high fidelity), predictions corresponding to the characteristic curves A, B and D of FIG. A case will be described in which three types of filters having coefficients are used and the sampling frequency f _{s is set} to 37.8 kHz, which is double the frequency of the first embodiment. Other specific numerical values and specific configurations are the same as those in the first embodiment.

In this case, the frequency characteristics of the above three types of filters are as shown by characteristic curves A, B and D in FIG. 6 as the sampling frequency f _s is doubled. That is, the characteristic curve D corresponds to the straight PCM mode, the characteristic curve B corresponds to the first-order difference mode, and the characteristic curve A corresponds to the second-order difference mode.

By the way, when the quantizer 16 requantizes one word from 16 bits to 4 bits, 4 bits at a predetermined position are taken out. At this time, in the shifter 15,
Since the range (bit shift amount) is determined based on the data before the noise shaping process, that is, the peak value in the block of the selected mode, the noise shaper 17
Since the feedback error from the above is superposed by the adder 14, there is a possibility that the data may overflow, especially at the time of inputting data close to the maximum value of the bits taken out during the ranging processing.

In order to prevent this, a coefficient γ equal to or greater than a predetermined value 1 with respect to the peak value (maximum absolute value) in the block of the selected mode
Is preliminarily multiplied, and the multiplication result is regarded as a peak value to determine the ranging position, that is, the 4-bit extraction position out of 16 bits. In this way, since the ranging position is determined by the predetermined number γ times the true peak value,
Even if the error from the noise shaper 17 is returned, the above overflow is less likely to occur. In this case, the coefficient γ is preferably set according to the predictor characteristic of the selected mode.

Here, as the predictor 20 in the noise shaper 17, a predictor 20 having a characteristic equal to that of the predictors 12A to 12D for difference processing is selected according to the mode selection, so that the straight P
In the CM mode, the coefficients Ka and Kb in FIG. 2 are both 0, the error feedback amount is 0, and the noise
There is no influence of shaping, and the coefficient γ may be set to 1.
In the first-order difference mode, γ≈, considering that the noise spectrum at the encoder output after the noise shaping process is equal to the characteristic curve B in FIG.
In the second-order difference mode, γ≈1.33 may be set in consideration of the characteristic curve A in FIG.

That is, when the first-order difference mode is selected, the noise spectrum at the encoder output is as shown in FIG. This is the instantaneous S / when requantizing with 4 bits.
Since N is about 24 dB, full scale (4
When the maximum level that can be expressed in bits is set to 0 dB, the noise level before noise shaping is -2.
The noise level becomes 4 dB, and the noise shaping process by the error feedback of the primary prediction is applied to this, and the noise level of the spectrum distribution as shown by the hatched portion in FIG. 7 is obtained. Therefore, the noise level near the frequency f _s / 2 is about 6 dB compared to the level before noise shaping.
It will be lifted and this will cause the above-mentioned overflow. This is because the amplitude distribution of the quantization noise before noise shaping is the requantized 4-bit LSB of Since it is random within the bit, the maximum amplitude of noise when taking the first-order difference of this is Becomes ± 1LSB, which is about 6 dB near f _s / 2.
Corresponding to the lifting of. Therefore, +6 even when there is no input
4 dB full,
This means that a noise peak value may exist at a position of about -18 dB near f _s / 2 with respect to 0 dB on the scale. Here, since -18 dB is about 0.125, if the peak value of the signal is suppressed to 1-0.125, that is, 0.875 times, overflow due to noise shaping can be prevented in advance. Therefore, the multiplication coefficient γ for the peak value for determining the ranging position may be 1 / 0.875≈1.14.

Next, regarding the multiplication coefficient γ with respect to the peak value for determining the ranging position when the second-order difference mode is selected, since the rise in the vicinity of f _s / 2 of the characteristic curve A of FIG. 6 is about +12 dB, 4-bit requantization full scale 0d
The peak value of the noise level for B would be located at about -12 dB. Since -12 dB is about 0.25, by suppressing the peak value of the signal by 1-0.25 = 0.75 times, overflow due to noise shaping can be prevented, and the coefficient γ becomes about 1.33 from 1 / 0.75.

By the way, when the ranging position at the time of requantization changes abruptly due to the rapid change of the signal, the noise of the previous block may be carried over to the next block, which may cause overflow. . This is especially likely to occur when the signal level sharply drops near the block boundary, and at this time, the ranging position, that is, the 4-bit extraction position for requantization, is as shown in FIGS. 8A and 8B, for example. , LS from MSB side in 16 bits
Although it moves abruptly to the B side, 4 bits extracted by requantization are obtained by directly returning the error amount of the data of the previous block (FIG. 8A) to the data of the next block (FIG. 8B). Data or overflow. This corresponds to the gain G of the shifter 15 which was small in the previous block but sharply increased in the next block.

Therefore, when overflow is allowed, and when overflow occurs, it is fixed to the maximum positive or negative value that can be expressed by 4 bits of requantization and so-called clipping is performed to prevent the inversion of the sign and to prevent abnormal noise. It is desirable to minimize the occurrence.

Prior to this clipping processing, the movement of the bit extraction position when the range becomes small (gain G becomes large) is limited, and, for example, as shown in FIG. The position moved to the side is defined as the bit extraction position or the ranging position.

Thus, by limiting the amount of movement when the ranging position moves from the MSB side of the original 16-bit data to the LSB side, it is possible to prevent a rapid range change,
The size of overflow due to noise shaping can be reduced.

Here, the limiting action of the ranging position movement when the above-mentioned secondary difference mode in which the overflow is most likely to occur, that is, the steepest noise shaping process is performed will be described.

The noise spectrum distribution at the encoder output when the second-order difference mode is selected appears in the same curve shape as the characteristic curve A in FIG. 6, and is larger than that before the noise shaping.
The noise peak value around f _s / 2 is about four times, that is, about +1
It can be lifted by 2 dB. Therefore, when the full scale of 4 bits of the requantized data is 0 dB, the peak value of noise can exist at a position of -12 dB, and noise occupies 0.25. next,
As to the movement of the ranging position, as described above, there is a limitation that the LSB side is within 1 bit per block. Therefore, the magnitude of the noise fed back in the next block is 0 dB at the full scale at this time. As
The maximum is -6 dB for one bit, that is, 0.5. If the above limitation is not set, the range change step has a maximum of 12 bits, so -12 + 6 x 1
2 = 60, that is, about 60 dB of noise is propagated. On the other hand, when the above restrictions are set,
Since noise occupies 0.5 at the maximum in the transient part, the above multiplication coefficient γ is not set for the signal (γ = 1)
Even if the maximum value of the signal is 1 and the maximum value of the noise is 0.5
Overflow of 1.5, or about +3.5 dB, is enough.

Next, after limiting the size of such overflow, the clipping processing as described above is performed.

Here, the clipping process means that when the above-mentioned overflow occurs, the data is fixed or clipped to the maximum positive or negative value of 4 bits extracted by requantization, thereby preventing the inversion of the sign and thus the error. However, the error that occurs at this time, that is, the difference between the true value and the clipped positive or negative maximum value is fed back (error feedback) and carried over to the next time, that is, the normal value. It has been confirmed that the adverse effects such as the distortion due to the clip hardly occur by directly performing the error feedback.

On the other hand, if the error feedback is stopped at the time of the above clipping, noise shaping is temporarily lost, and the decoder output has a large level of low-frequency noise, that is, noise according to the characteristics of the decoding filter. Occurs, and the distortion propagates, which is very problematic in hearing.

Differences in operation due to the presence / absence of error feedback during such clipping processing will be described below.

Here, in order to simplify the explanation, as shown in FIG.
Consider the following case of noise shaping.

In FIG. 9, the output d ″ (n) from the shifter 15
Is requantized from 16 bits to 4 bits in the quantizer 16 and becomes the output d (n), which is output via the clipping circuit (clipper) 26 at the time of overflow. Shall be In addition, the input d ″ to the quantizer 16
(n) and output from clipper 26 The quantization noise or error e (n) which is the error component of
In the noise shaper 17, it is supplied as a subtraction signal to the adder 14 via the shifter 19 and the 1-word delay means 27.

It should be noted that, in contrast to the encoder having the configuration of FIG. 9 as a main portion, the main portion of the decoder is configured as shown in FIG. 10, and the predictor 34 is composed of the 1-word delay means 28.

Now, when shifting from one arbitrary block (first block) to the next block (second block) with the passage of time, the peak value of the signal becomes small, and the gain G of the shifter 15 becomes small.
When G changes from G to G · g (where g> 1), the final encoder output data of the first block is Output from the decoder shifter 32 Becomes Next, the encoder output data at the beginning of the second block Becomes Where the decoder output When thinking about Then, from the formula, Also, from this formula and the formula, Becomes As shown in these equations, when there is no overflow, no interference occurs between blocks, and a large quantization error in the previous block does not cause a trail in the subsequent block.

Next, when overflow occurs in the first word of the second block, the term e (n) · g in the above equation causes overflow. E when the error amount when
Rewriting the formula as Therefore, the above equation becomes Becomes Also, the decoder output of the above equation is The difference between this equation and the above equation is E · G ⁻¹ · g ⁻¹ …………, which is the distortion that appears in the decoder output due to clipping.

Next, regarding the error propagation by the clip, when the error generated in the above formula is fed back as usual, the encoded output corresponding to n + 2 is From this equation and the equation, the decoder output is And the effect of E disappears. On the other hand, if you do not feed back the error that occurred in the above formula, From this equation and the above equation, the decoder output is Therefore, as a result, the effect of not returning E + e (n + 1) is propagated.

As is clear from the results of the above consideration, it is better to feed back the error due to clipping as well as the normal quantization error. In this case, only the word that caused the overflow is E · G ⁻¹ · g ^{−1 in the} above equation.
Error is not propagated to subsequent words.

Next, the effect of limiting the amount of movement when the ranging position, that is, the requantization bit extraction position moves to the LSB side will be described.

First, feeding back the error due to clipping to the next word is shown in the above equation. The error returned in this equation is {e (n + 1) +
E} · G ⁻¹ · g ⁻¹ , which can cause overflow in n + 2 words. That is, when the requantization bit extraction position (ranging position) is rapidly moved to the LSB side as shown in FIG. 8B, However, Fs may have a full-scale size that can be represented by the quantized bits of the current block, and the error E in the word n + 1 cannot be completely sucked in the word n + 2, causing continuous overflow. It will end up. In this way, errors due to overflow propagate one after another, resulting in a large distortion. This state is shown in FIG. In FIG. 11, the first word in the block when the ranging position is rapidly moved to the LSB side is W _0, and the previous word (the last word of the previous block) is added to the data D ₀ of this word W _0. The quantization error (error) is superposed to obtain the true value P _0. This true value P ₀ exceeds the range of the full scale FS that can be represented by the requantization bit of the current block. Therefore, the output value Q ₀ is pasted to the positive maximum value by performing the clipping process. The quantization error E _{0 at} this time is the output value Q ₀ −the true value P ₀ . An inverted version of the sign of this error E ₀ , namely −
E ₀ is superposed on the data D ₁ of the next word W ₁ and becomes the true value P ₁ , but since this true value P ₁ also exceeds the full scale FS and becomes overflow, the output value Q ₁ becomes Clipped to maximum positive value. In this way, the error propagates. Note that, in FIG. 11, the true value P ₀ of the block head word W ₀ with error superposition is set within several times the full scale for the sake of illustration, but it is actually fed back to W _0. The error can be about 1000 times full scale (about 60 dB), and the error propagation can connect for long periods of time.

On the other hand, as described above, when the amount of movement when the ranging position (requantization bit extraction position) moves to the LSB side is limited to, for example, 1 bit, as shown in FIG. The error fed back at the first word W ₀ is suppressed to a small level, and error propagation is eliminated in a short time. Therefore, the distortion of the output signal from the decoder can be suppressed to a small level, and in combination with the temporal masking effect, it is possible to perform signal transmission without any trouble in hearing.

Third Embodiment Next, a third embodiment capable of high-fidelity transmission with higher quality than the second embodiment will be described.

In the third embodiment, the requantization bit number in the quantizer 16 is set to 8 bits, and the difference processing filter outputs one of primary difference PCM data and one of straight PCM data. The type is used. The sampling frequency f _s is 3 as in the second embodiment.
The characteristic curve of the first-order differential filter is equal to the curve B in FIG. Other conditions and configurations are
This is similar to the second embodiment described above.

In the case of the third embodiment, since the requantized audio data word is 8 bits, higher quality signal transmission can be performed as compared with the second embodiment. The average bit rate of transmitted data is 8.
With 214 bits, approximately 310.5 kbit / s per channel.

By the way, in the present invention, only a filter that does not make the prediction gain G _{p in} the low band of f _s / 6 or less negative in dB display is used, and a so-called summing filter is not used. This is because it is very rare that only a high frequency signal exists in a wide band music signal, and it is always low and
This is because a difference processing filter that considers the presence of the mid-range signal is suitable.

The present invention is not limited to the above embodiment, but the sampling frequency of the input digital signal, the number of bits of one word, the number of words in one block, the highest order N and the number of types of filters, or requantization bits. Of course, the number and the like can be set arbitrarily.

〔The invention's effect〕

According to the signal transmission device of the present invention, a plurality of difference processing filters that output each data of high-order differential PCM, first-order differential PCM, and straight PCM are used, and these are adaptively switched and selected. Bit rate can be reduced,
Signal transmission at an extremely low bit rate is possible without deteriorating the signal quality. Also, straight PCM
Since the data output mode can be switched and selected, it is possible to solve the S / N deterioration at the time of inputting a high frequency signal and the generation of excessive error power at the time of error occurrence.

Also, the peak value (maximum absolute value) in the block of the output from the selected filter is multiplied by the coefficient γ (γ ≧ 1), and the ranging position, that is, the requantization bit extraction position is based on this multiplied value. By determining, the overflow is less likely to occur even if the feedback error due to the noise shaping process is superimposed.

Furthermore, when the input signal level sharply drops near the block boundary and the ranging position, that is, the requantized bit extraction position is about to suddenly move to the LSB side, by limiting the moving amount, It is possible to suppress the error due to the overflow of the quantization bit. At this time, when the requantization bit overflows, clipping is performed at the maximum positive or negative value, and the error at this time is fed back to perform noise shaping processing to suppress error propagation and generate abnormal noise. It is possible to effectively prevent the adverse effects of the above.

[Brief description of drawings]

FIG. 1 is a block circuit diagram showing a schematic configuration of the entire system to which the signal transmission device according to the present invention is applied, and FIGS. 2 and 3 are more specific configuration examples of the encoder and decoder of FIG. FIG. 4 is a block circuit diagram showing each of them, FIG. 4 is a graph showing frequency characteristics of a plurality of difference processing filters, and FIG. 5 is a diagram showing an example of a transmission word configuration in one block.
FIG. 6 is a graph showing the frequency characteristics of a plurality of difference processing filters used in another specific example, FIG. 7 is a graph showing the spectral distribution of noise subjected to noise shaping processing, and FIG. 8 is a graph of requantization. FIG. 9 is a block circuit diagram showing the main part of the encoder, FIG. 10 is a block circuit diagram showing the main part of the decoder, and FIGS. 11 and 12 are requantization diagrams. It is a figure for explaining error propagation by overflow at the time of conversion. 10 ............... Encoder 12, 12A to 12D, 20, 34 ... Predictor 15, 19, 32 ... Shifter 16 ……………… Quantizer 17 ……………… …… Noise shaper 21 …………………… Prediction / range adaptation circuit 30 …………………… Decoder

Claims (2)

[Claims] 1. An input digital signal is divided into blocks by a certain number of words along a time axis, and is transmitted through a filter for obtaining a prediction error with respect to the signal of each block. Multiple predictors and a predictor of order N or less are provided, and the maximum absolute value of the output from each filter in the above block or the product obtained by multiplying the maximum absolute value by a coefficient is compared with each other, and the values are compared. A signal transmission device characterized by selecting a filter with a minimum value. 2. The signal transmission device according to claim 1, wherein at least one of the plurality of filters is provided with a filter for outputting substantially straight PCM data.