JPH0642631B2 - Signal transmission device - Google Patents

Description

Description: TECHNICAL FIELD The present invention relates to a signal transmission device for transmitting a signal obtained by digitizing an analog signal such as a PCM signal by dividing it into blocks of a certain number of words, and in particular, each block. The present invention relates to a signal transmission device that reduces the transmission bit rate by performing companding processing every time.

[Conventional technology]

In recent years, sampling (sampling) of analog audio signals, video signals, etc., performing quantization and coding processing, and transmitting or recording / reproducing as so-called PCM (pulse coat modulation) signals has become more common. There is.

When transmitting or recording / reproducing such a PCM signal, the sampling frequency s is set to 4 in order to obtain a band of about 20 kHz and an S / N of about 90 dB or more.
The linear quantization of 1 word and 16 bits is generally adopted at 4.1 kHz, but the transmission rate in this case is 700
It is extremely high, reaching over KBPS (700 Kbits per second).

By the way, in a digital signal obtained by A / D converting an analog signal such as an audio signal or a video signal as described above, there is a bias in the statistical properties, or a portion of low importance from the viewpoint of audiovisual development. It is possible to compress the amount of information by utilizing the fact that there is, for example, difference / union processing and compression / expansion processing (companding processing).
It is known that signal quality deterioration is extremely small even if the above is performed.

In consideration of such a point, the applicant of the present invention first divides a digital PCM signal into blocks, for example, in a fixed time unit or in a fixed number of words, and performs prediction processing such as difference processing and companding for each block. It is disclosed in Japanese Patent Application No. Sho 58-97 that processing, transmission or recording / reproduction is performed.
687-9, Japanese Patent Application No. 58-163054, Japanese Patent Application No. 5
No. 8-166267 or Japanese Patent Application No. 58-210382
No., etc.

At least one for each block in these techniques
Word reference data, for example, straight PCM data, is provided, and based on this reference data, by performing arithmetic processing such as sequentially adding difference data, for example, all the original sampling data (straight PCM data) in the block can be obtained. Restorable. As the companding process, it has been proposed to perform re-quantization of input data and perform noise shaping processing by feeding back a prediction value of a quantization error at this time (so-called error feedback). It is desirable that the quantization error prediction process is performed separately from the signal prediction process so as not to deteriorate the instantaneous S / N. In this case, the original data of the requantized bits during the above requantization
The extraction position for the bit, so-called ranging position is determined based on the metadata before the noise shaping processing.

[Problems to be solved by the invention]

However, when the signal prediction process and the noise prediction process are separated, if higher-order noise shaping comes to be performed, it will be over-quantized when requantizing the signal on which the feedback error is superimposed. There is a risk of flow,
This overflow causes adverse effects such as abnormal noise.

In view of such a situation, the present invention makes it difficult to cause overflow at the time of requantization due to error feedback while performing the signal and noise prediction processing separately, and prevents adverse effects such as generation of abnormal noise. It is an object of the present invention to provide such a signal transmission device.

[Means for solving problems]

In order to solve the above-mentioned problems, the signal transmission device of the present invention comprises means for dividing an input digital signal into blocks by a certain number of words along a time axis, and performing a prediction process on the signal of each block, A means for requantizing the prediction processed signal and feeding back a quantization error so-called error to perform noise shaping processing; and a coefficient γ of 1 or more for the maximum absolute value in the block of the prediction processed signal. Means for determining a requantized bit extraction position (ranging position) in the requantization based on the result of the multiplication.

[Action]

For the maximum absolute value within the block of the prediction processed signal,
In consideration of the amount of error to be fed back, a coefficient γ of 1 or more is multiplied in advance, and the ranging position at the time of requantization is determined based on the multiplication result. Is less likely to cause overflow when requantized.

〔Example〕

Schematic Configuration First, the overall schematic configuration of an audio bit rate reduction system as an example of a signal transmission device to which the present invention is applied will be described with reference to FIG.

The system shown in FIG. 1 comprises an encoder 10 on the transmitting side (or recording side) and a decoder 30 on the receiving side (or reproducing side). An analog audio signal of frequency s is input to an input terminal 11 of the encoder 10. An audio PCM signal x (n) obtained by sampling, quantizing, and coding is supplied. This input signal x
(n) is sent to the predictor 12 and the adder 13, respectively, and is the prediction signal from the predictor 12. Is sent to the adder 13 as a subtraction signal. Therefore, in the adder 13, the predicted signal is calculated from the input signal x (n). Is subtracted to obtain a prediction error signal or (broadly defined) difference output d (n), that is, Is output.

Here, the predictor 12 generally uses the past p inputs x (n
-P), x (n-p + 1), ..., x (n-1) primary combination predicted value To calculate However, α _k (k = 1, 2, ..., P) is a coefficient. Therefore, the prediction error output or (broadly defined) difference output d (n) is Can be expressed as

Further, according to the present invention, the input digital signal is divided into data within a fixed time, that is, every fixed word number l of the input data, and the coefficient α _k is set so that an optimum prediction filter characteristic is obtained for each block. Is selected. As will be described later, this can be regarded as providing a plurality of filters for obtaining a difference output (prediction error output) including a predictor having different characteristics or an adder. The optimum filter among the processing filters is selected for each block. The selection of this optimum filter is performed by the prediction / range adaptation circuit 21 using the maximum absolute value (peak value) in the block of the output from each of the plurality of difference processing filters or the value obtained by multiplying the maximum absolute value (peak value) by a coefficient. This is performed by comparing, and specifically, the difference processing filter that minimizes the maximum absolute value (or its coefficient multiplication value) is selected as the optimum filter for the block. The optimum filter selection information at this time is output from the prediction / range adaptation circuit 21 as mode selection information and sent to the predictor 12.

Next, the difference output d (n) as the prediction error is sent to the bit compression means composed of the shifter 15 of the gain G and the quantization 16 via the adder 14 to display, for example, a floating point (floating point). A compression process or a ranging process is performed such that the exponent part in the form corresponds to the gain G and the mantissa part corresponds to the output from the quantizer 16. That is, the shifter 15 shifts the digital binary data by the number of bits according to the gain G (arithmetic shift).
By doing so, the so-called range is switched, and the quantizer 16 performs requantization so as to take out a fixed number of bits of this bit-shifted data. Next, the noise shaping circuit (noise shaper) 17 obtains a so-called quantization error for the error between the output and the input of the quantizer 16 by the adder 18, and this quantization error is shifted by a gain G ^-1 . The so-called error feedback is performed by sending the prediction signal of the quantization error to the adder 14 as a subtraction signal by sending it to the predictor 20 via 19. At this time, the prediction / range adaptation circuit 21 outputs lens information based on a value obtained by multiplying the maximum absolute value in the block of the difference output from the filter of the selected mode by a coefficient γ of 1 or more, and this range information is output. The gains G and G ⁻¹ are sent to the shifters 15 and 19 and determined for each block. The characteristics of the predictor 20 are determined by sending the mode information from the prediction / lens adaptation circuit 21.

Therefore, the output d '(n) from the adder 14 is the prediction signal of the quantization error from the noise shaper 17 from the difference output d (n). Subtracted Therefore, the output d ″ (n) from the shifter of gain G is d ″ (n) = G · d ′ (n). Also, the output from the quantizer 16 Let e (n) be the quantization error in the quantization process, Therefore, the quantization error e (n) is taken out by the adder 18 of the noise shaper 17, and the shifter 1 having the gain G ^-1 is obtained.
Predictor 2 that takes a linear combination of past r inputs via 9
Prediction signal of quantization error obtained through 0 Is Becomes This equation has the same form as the above equation, and the predictors 12 and 20 are FIR (finite impulse response) filter.

From these expressions, the output from the quantizer 16 Is Substituting the above formula into d (n) of this formula, And this output Are taken out through the output terminal 22. Where above Z transformation of Then, Becomes

The range information from the prediction / lens adaptation circuit 21 is taken out from the output terminal 23, and the mode selection information is taken out from the output terminal 24.

Next, the input terminal 3 of the decoder 30 on the receiving side or the reproducing side
1, the output from the output terminal 22 of the encoder 10 The signal obtained by transmitting, recording, or reproducing Is being supplied. This input signal Is sent to the adder 33 via the shifter 32 having a gain G ⁻¹ . Output from adder 33 Is sent to the predictor 34 and the prediction signal And this predicted signal Is sent to the adder 33 and output from the shifter 32 Is added. This added output is the decoded output Is output from the output terminal 35.

The range information and the mode selection information transmitted or recorded / reproduced from the output terminals 23 and 24 of the encoder 10 are input to the input terminals 36 and 37 of the decoder 30, respectively. Then, the range information from the input terminal 36 is sent to the shifter 32 to determine the gain G ⁻¹ , and the mode selection information from the input terminal 37 is sent to the predictor 34 to determine the prediction characteristic. The prediction characteristic of the predictor 34 is selected to be equal to the characteristic of the predictor 12 of the encoder 10.

In the decoder 30 having such a configuration, the output from the shifter 32 Is And the output of the adder 33 Is Becomes Here, since the predictor 34 selects a characteristic equal to that of the predictor 12 of the encoder 10, Therefore, from the formula, Becomes next, Z transformation of Then, Therefore, Becomes Here, assuming that there is no error in the transmission path or recording medium, Then, from the above formula and formula, Becomes

From this equation, it is clear that the noise reduction effect of G ⁻¹ is obtained with respect to the quantization error E (z). At this time, if the spectral distribution of noise appearing in the decoder output is N (z), Becomes

Further, in such a system, the G acts to normalize with a value related to the maximum absolute value in the block, but the G has a frequency characteristic. Here, in order to simplify the description, the above G is represented as a product of two elements G _p and G f such as G = G _p · Gf. Of these two elements, G _p means the prediction gain by the above prediction filtering, that is, the improvement amount of the instantaneous S / N,
Gf means the amount of gain control by the above ranging processing, that is, the amount of expansion of the dynamic range. Therefore, G _p depends on the input signal frequency and does not depend on the input signal level, whereas G f does not depend on the input signal frequency but depends on the input signal level. Also, G _p is Has a S / N improvement amount of, and has a frequency characteristic that is an inverse function of the transfer function 1-P (z) of the difference processing filter for obtaining the prediction error. Becomes like the above formula. Regarding Gf, it corresponds to quasi-instantaneous companding which is normalized by the maximum absolute value in the block in the mode selected for each block.

In a system having such an overall configuration, the gist of the present invention is that the difference selected in the prediction / range adaptation circuit 21 for determining the gain G of the shifter 15, the gain G _{−1 of the} shifter 19 and the like. Output d from the processing filter
For the maximum absolute value (peak value) in the block of (n),
This is to multiply one or more coefficients γ in consideration of the amount of feedback error and to determine the gains of the shifters 15 and 19 based on the multiplied value.

Specific Configuration Example Next, FIG. 2 shows a more specific configuration example of the encoder 10 of the audio bit rate reduction system shown in FIG. Have the same reference numbers.

In FIG. 2, as the predictor 12, a plurality of, for example, four predictors 12A, 12B, 12C, 12 are used.
D is provided. Predicted outputs from these predictors 12A to 12D are respectively added by adders 13A as subtraction signals.
~ 13D and subtracted from the original input signal. That is, four predictors 12A to 12D and four adders 13A to 13D respectively form four difference processing filters for obtaining prediction errors of four systems. Here, each of the predictors 12A to 12D apparently has a quadratic configuration, and the predictor 12A uses the coefficients α ₁ and α ₂ as the coefficients.
K ₁ , K ₂ , 12B are K ₃ , K ₄ , 12C are K ₅ , K ₆ , 12D
Has K ₇ and K ₈ , but by setting at least one of the coefficients of the desired predictor to 0, it is possible to obtain a prediction characteristic of order 1 or lower. Therefore, the four difference processing filters described above are also configured to take the quadratic difference in appearance, but the desired difference processing filter has the characteristic of taking the difference of not more than the first order (that outputs straight PCM data. It is also possible to obtain).

The output from each of the difference processing filters, that is, the output from each of the adders 13A to 13D is the l-word delay circuits 41A to 41D and the maximum absolute value hold circuit 4, respectively.
2A to 42D, the 1-word delay circuit 41A
Each output from ~ 41D is a mode changeover switch circuit 43.
Of the selected terminals a to d. That is, since one block has 1 word, the 1-word delay circuit 4
1A to 41D delays one block, and during this delay, the maximum absolute value (peak) hold circuits 42A to 42D detect the maximum absolute value within the block. is there. The maximum absolute values in these blocks are sent to the prediction / range adaptation circuit 21 and compared with each other, and the one having the minimum value is selected. At this time, the maximum absolute value in each block may be multiplied by a predetermined coefficient to perform so-called weighting and then compared. The prediction / range adaptation circuit 21 outputs mode selection information for selecting one block of data from which the minimum in-block maximum absolute value is obtained among the one block of data from each of the difference processing filters. Then, by transmitting the mode selection information to the changeover switch circuit 43, the changeover connection to the delay circuit for outputting the selected one block of data is performed. The output from the changeover switch circuit 43 is sent to the adder 14.
The mode selection information from the prediction / range adaptation circuit 21 is also sent to the predictor 20 and the output terminal 24. Here, the predictor 20 selects one having a characteristic equal to the selected one of the predictors 12A to 12D in order to make noise (see the above equation) appearing in the decoder output white, for example. That is, the predictor 20 also apparently has a quadratic predictor configuration, and the coefficients β ₁ and β ₂
The coefficient Ka, Kb corresponding to is equal to the coefficient of the predictor of the difference processing filter designated by the mode selection among the coefficient sets K ₁ , K _{2 to} K ₇ , K ₈ of the predictors 12A to 12D. Things are selected.

Further, in the specific example of the decoder 30 shown in FIG. 3, the predictor 34 has an apparent quadratic structure corresponding to the predictors 12A to 12D of FIG. 2, and each coefficient Kc, Kd
Is a set of coefficients K ₁ and K _{2 of the} predictors 12A to 12D.
Any one of K ₇ to K ₈ is selected according to the mode selection information from the input terminal 37.

The other configurations of FIGS. 2 and 3 are the same as those of FIG. 1 described above, and thus the description thereof will be omitted.

It should be noted that the encoder 10 having the specific configuration as described above
As a hardware configuration of the decoder 30 and the decoder 30, for example, it is not necessary to actually provide a plurality of predictors 12A to 12D and the like, and the coefficient of one predictor may be used by switching in a time division manner. It is needless to say that the entire 10 and the decoder 30 can be realized by software by a system including a DSP (digital signal processor) and a memory.

First Embodiment Here, as a first embodiment of the present invention, the sampling frequency s of the audio PCM signal supplied to the input terminal 11 is set to 18.9 kHz, and the word number l of one block is 28 bits per word 16 bits. The case of requantizing the data into words of 4 bits in the quantizer 16 will be described in detail. The secondary predictor 12 at this time
As the groups K ₁ , K _{2 to} K ₇ , K ₈ of the coefficients A to 12D, for example, 12A: K ₁ = 1.8426, K ₂ = −0.8649 12B: K ₃ = 0.875, K ₄ = 012C: K _{5 = 1.5155, K 6 = -0.81} 12D: preset as _{K 7 = 0, K 8 =} 0. The transfer function 1-P (z) of the differential processing filter in each mode at this time is: A: 1-1.8426z ^-1 + 0.8649z ^-2 B: 1-0.875z C: 1-1.5155z ^-1 +0.81 z ^-2 D: 1, and the frequency characteristics of each of these difference processing filters are
The curves A to D in FIG. 4 are obtained.

That is, the difference processing filter (comprising the predictor 12A and the adder 13A) corresponding to the characteristic curve A is the secondary difference PC.
It is a filter corresponding to the M mode, and the prediction gain of reduction, that is, the improvement amount of the instantaneous S / N is large. The difference processing filter (the predictor 12B and the adder 1 corresponding to the characteristic curve B
3B) is a filter corresponding to the first-order difference PCM mode, and the difference processing filter (the predictor 12C and the adder 13C) corresponding to the characteristic curve C has a large intermediate-range prediction gain. Since the coefficients K ₇ and K ₈ are both 0, the difference processing filter including the predictor 12D and the adder 13D does not have a frequency characteristic and has a so-called reference gain of 1 as shown by the characteristic curve D in FIG. It has a mere flat path characteristic and is equivalent to the straight PCM mode.

FIG. 5 shows a specific example of the word structure transmitted for each block. From re-quantized 1 word 4 bits, 28 words of audio data words W _{0 to} W ₂₇ and 1 word 16 bits are shown. mode selection information indicating 4 so-called range information word W _R that indicates the 4 bits of the take-out position when the re-quantization for each block to the bit, either has been selected out of four modes corresponding to the four filters a word W _M is transmitted block by block. Therefore,
The average number of bits per transmitted audio data word is (4 × 28 + 4 + 2) ÷ 28≈4.214 [bit].

In FIG. 4, when a single sine wave signal is input, the characteristic curve A is obtained when the input signal frequency is 0 to _1.
Of the characteristic curve C for frequencies ₁ to ₂
Of the characteristic curve D and the filter of the characteristic curve D are selected for frequencies from ₂ to _{s 2/2} . It is needless to say that _s / 2 or more of the frequency of the input signal is removed in advance by an LPF (low pass filter) before A / D conversion in order to prevent so-called aliasing.
The frequency response of each filter selected in this way becomes the predicted gain at that frequency, that is, the improvement amount of the instantaneous S / N, and the improvement amount of the instantaneous S / N with respect to the frequency is as shown by the hatched portion in FIG.

However, since the actual audio input signal is a composite spectrum signal, the selection based on the above-described clear boundary is not performed, and the filter of the characteristic curve B is also used relatively frequently.

Further, the above-mentioned filter selection is performed when the maximum absolute value (peak value) in the block from the filter of each mode is directly compared, but a predetermined coefficient is set for each peak value in the block of each mode. Low-order filter or straight PC by multiplying by and comparing
You may make it select more filters which output M data. In this case, as an example of the coefficient, the peak value of the second-order differential PCM data from the filter of the characteristic A is multiplied by the coefficient 2, and the peak value of the straight PCM data from the filter of the characteristic D is multiplied by the coefficient 0.7. Then, the peak value of the data from the other filter is multiplied by the coefficient 1 (or not multiplied by the coefficient), and the respective values are compared with each other.
Select a filter that minimizes its value. The multiplication of such a coefficient corresponds to translating the corresponding characteristic curve in the graph of FIG. 4 in the vertical axis direction (changing the response value). For example, the peak value from the filter of the characteristic curve A is calculated as a coefficient. Multiplying by 2 gives the characteristic curve A a fourth
This corresponds to a parallel translation of about 6 dB upward in the figure. Therefore, as a result, the filter switching frequencies ₁ and ₂
Will shift to the low frequency side, and the filter of the characteristic C will be selected more frequently than the filter of the characteristic A, and the filter of the characteristic D will be selected more frequently than the filter of the characteristic C.

By thus increasing the frequency of use of the low-order filter, it is possible to suppress the influence of code errors on the transmission line.

In the decoder side, the audio data words W ₀ to _W-27 of one block which is input to the contrary to the expansion processing or the encoder side in block units by the shifter 32 based on the data of the range information word W _R ranging process is performed, the filter next made of the adder 33 and predictor 34, inverse predictive processing in blocks on the basis of the data of the mode selection information word W _M is performed, the original straight PCM data is restored It

Further, the ranging position related to the gist of the present invention, that is, the position for extracting 4 bits from 16 bits in the requantization, which is determined according to the gain G of the shifter 15, is selected in the prediction / range adaptation circuit 21. It is determined based on the value obtained by multiplying the in-block peak value of the output from the difference processing filter of the mode by a coefficient γ of 1 or more.

According to the system of the first embodiment, it is possible to transmit a sound quality of low to medium fidelity, and a normal voice signal or the like is transmitted at an extremely low bit rate (4.214 bits per word, 1
Transmission bit rate per channel About 79.6 kbit /
Seconds) can be transmitted.

Second Embodiment Next, as an example of a system capable of transmitting a music signal with sound quality of middle to high fidelity (middle to high fidelity), predictions corresponding to the characteristic curves A, B and D of FIG. Using three types of filters having coefficients, the sampling frequency s is set to the first
A case will be described in which the frequency is set to 37.8 kHz, which is twice that in the embodiment of FIG.
Other specific numerical values and specific configurations are the same as those in the first embodiment.

In this case, the frequency characteristics of the above three types of filters become as shown by characteristic curves A, B and D in FIG. 6 as the sampling frequency s doubles. That is, the characteristic curve D corresponds to the straight PCM mode, the characteristic curve B corresponds to the first-order difference mode, and the characteristic curve A corresponds to the second-order difference mode.

By the way, when the quantizer 16 requantizes one word from 16 bits to 4 bits, 4 bits at a predetermined position are taken out. In the prior art of the present invention, the noise shaping is performed by the shifter 15. Since the range (bit shift amount) is determined based on the data before processing, that is, the peak value in the block of the selected mode, the feedback error from the noise shaper 17 is superposed by the adder 14. In particular, there is a risk that data may overflow when data is input that is close to the maximum value of the bits extracted during the ranging process.

In order to prevent this, in the present invention, the peak value (maximum absolute value) in the block of the selected mode is premultiplied by a predetermined coefficient γ of 1 or more, and the multiplication result is regarded as the peak value. The ranging position, that is, the extraction position of 4 bits out of 16 bits is determined. in this way,
Since the ranging position is determined by the predetermined number γ times the true peak value, even if an error from the noise shaper 17 is fed back, the overflow is less likely to occur. In this case, the coefficient γ is preferably set according to the predictor characteristic of the selected mode.

Here, the predictor 20 in the noise shaper 17 has a characteristic equal to that of the predictors 12A to 12D for difference processing and is selected according to the mode selection.
In the M mode, the coefficients Ka and Kb in FIG. 2 are both 0, the error feedback amount is 0, and there is no influence of noise shaping, and the coefficient γ = 1 may be set. In addition, in the first-order difference mode, γ ≈ 1. considering that the noise spectrum at the encoder output after the noise shaping processing is equal to the characteristic curve B in FIG.
In the second-order difference mode, γ≈1.33 may be set in consideration of the characteristic curve A of FIG.

That is, when the first-order difference mode is selected, the noise spectrum at the encoder output is as shown in FIG. This is the instantaneous S / when requantizing with 4 bits.
Since N is about 24 dB, when the full scale (maximum level that can be represented by 4 bits) is 0 dB of the reference,
The noise level before noise shaping becomes -24 dB, and the noise shaping process by the error feedback of the primary prediction is applied to this to obtain the noise level of the spectrum distribution as shown by the shaded area in FIG. .
Therefore, the noise level near the frequency s / 2 is raised by about 6 dB as compared with the level before noise shaping, which causes the above-mentioned overflow. This is a 4-bit LS obtained by requantizing the amplitude distribution of quantization noise before noise shaping.
B's Since it is random within, the maximum amplitude of the nozzle when taking the first-order difference is Becomes ± 1LSB, which is about 6 dB near s / 2.
Corresponding to the lifting of. Therefore, +6 even when there is no input
Since the nozzle of dB is transmitted, the peak value of noise may exist at a position of about -18 dB near s / 2 with respect to 0 dB of 4-bit full scale. here,
-18dB is about 0.125, so the peak value of the signal is 1-
If it is suppressed to 0.125, that is, 0.875 times, overflow due to noise shaping can be prevented beforehand.
Therefore, the multiplication coefficient γ for the peak value for determining the ranging position may be 1 / 0.875≈1.14.

Next, regarding the multiplication coefficient γ with respect to the peak value for determining the ranging position when the second-order differential mode is selected, since the lifting up of the characteristic curve A in FIG. Quantization full scale 0 dB
On the other hand, the peak value of the noise level can exist at the position of about -12 dB. Since −12 dB is about 0.25, overflow due to noise shaping can be prevented by suppressing the peak value of the signal by 1-0.25 = 0.75, and the coefficient γ becomes about 1.33 from 1 / 0.75.

By the way, when the ranging position at the time of requantization changes abruptly due to the rapid change of the signal, the noise of the previous block may be carried over to the next block, which may cause overflow. . This is especially likely to occur when the signal level sharply drops near the block boundary, and at this time, the ranging position, that is, the 4-bit extraction position for requantization, is as shown in FIGS. 8A and 8B, for example. , LS from MSB side in 16 bits
Although it moves abruptly to the B side, 4 bits extracted by requantization are obtained by directly returning the error amount of the data of the previous block (FIG. 8A) to the data of the next block (FIG. 8B). Data will overflow. This corresponds to the gain G of the shifter 15 which was small in the previous block but sharply increased in the next block.

Therefore, when overflow is allowed, and when overflow occurs, it is fixed to the maximum positive or negative value that can be expressed by 4 bits of requantization and so-called clipping is performed to prevent the inversion of the sign and to prevent abnormal noise. It is desirable to minimize the occurrence.

Prior to this clipping processing, the movement of the bit extraction position when the range becomes small (gain G becomes large) is limited, and, for example, as shown in FIG. The position moved to the side is defined as the bit extraction position or the ranging position.

Thus, by limiting the amount of movement when the ranging position moves from the MSB side of the original 16-bit data to the LSB side, it is possible to prevent a rapid range change,
The size of overflow due to noise shaping can be reduced.

Here, the limiting action of the ranging position movement when the above-mentioned secondary difference mode in which the overflow is most likely to occur, that is, the steepest noise shaping process is performed will be described.

The noise spectrum distribution at the encoder output when the second-order difference mode is selected appears in the same curve shape as the characteristic curve A in FIG. 6, and the peak value of the noise near s / 2 compared to before noise shaping. Is about 4 times or about +
It can be lifted by 12 dB. Therefore, when the full scale of 4 bits of requantized data is 0 dB,
The peak value of noise can exist at a position of -12 dB, and noise will occupy 0.25. Next, with respect to the movement of the ranging position, as described above, since there is a limitation that the LSB side is within 1 bit per block, the magnitude of the noise fed back in the next block is the full level at this time. When the scale is 0 dB, the maximum is -6 dB for one bit, that is, 0.5. In addition,
If the above limitation is not set, the maximum number of lens change steps is 12 bits, so -12 + 6 × 12 = 6.
0, that is, about 60 dB of noise will be propagated. On the other hand, when the above-mentioned limitation is set, noise occupies 0.5 at the maximum in the transient part, and even when the above multiplication coefficient γ is not set for the signal (γ = 1), the signal The sum of the maximum value 1 and the maximum noise value 0.5 is 1.5, that is, about +3.5 dB overflow is sufficient.

Next, after limiting the size of such overflow, the clipping processing as described above is performed.

Here, the clipping process means that when the above-mentioned overflow occurs, the data is fixed or clipped to the maximum positive or negative value of 4 bits extracted by requantization, thereby preventing the inversion of the sign and the error. Although it is suppressed to a small level, the error that occurs at this time, that is, the difference between the true value and the clipped positive or negative maximum value is fed back (error feedback), and is passed over to the next time, that is, the normal error. -It has been confirmed that if feedback is performed as it is, the adverse effects such as distortion due to clipping hardly occur.

On the other hand, if the error feedback is stopped at the time of the above clipping, noise shaping is temporarily lost, and the decoder output has a large level of low-frequency noise, that is, noise according to the characteristics of the decoding filter. Occurs, and the distortion propagates, which is very problematic in hearing.

Differences in operation due to the presence or absence of error feedback such as clipping processing will be described below.

Here, in order to simplify the explanation, as shown in FIG.
Consider the following case of noise shaping.

In FIG. 9, the output d ″ (n) from the shifter 15
Is requantized from 16 bits to 4 bits in the quantizer 16 and becomes the output d (n), which is output via the clipping circuit (clipper) 26 at the time of overflow. Shall be In addition, the input d ″ to the quantizer 16
(n) and output from clipper 26 Quantization noise or error e (n)
Via the shifter 19 in the noise shaper 17,
It is supplied as a subtraction signal to the adder 14 via the 1-word delay means 27.

It should be noted that, in contrast to the encoder having the configuration of FIG. 9 as a main portion, the main portion of the decoder is configured as shown in FIG. 10, and the predictor 34 is composed of the 1-word delay means 28.

Now, when moving from an arbitrary block (first block) to the next block (second block) with the passage of time, the peak value of the signal becomes small, and the gain G of the shifter 15 is reduced.
When G changes from G to G · g (where g> 1), the final encoder output data of the first block is Output from the decoder shifter 32 Is Becomes Next, the encoder output data at the beginning of the second block Is Also, the data in the decoder Is Becomes Where the decoder output When thinking about Then, from the formula, Also, from this formula and the formula, Becomes As shown in these equations, when there is no overflow, no interference occurs between blocks, and a large quantization error in the previous block does not cause a trail in the subsequent block.

Next, when overflow occurs in the first word of the second block, the term e (n) · g in the above equation causes overflow. E when the error amount when
Rewriting the formula as Therefore, the above equation becomes Becomes Also, the decoder output of the above equation is The difference between this equation and the above equation is E · G ⁻¹ · g ⁻¹ ···, which is the distortion that appears in the decoder output due to clipping.

Next, regarding the error propagation by the clip, when the error generated in the above formula is fed back as usual, the encoded output corresponding to n + 2 is Therefore, Is From this equation and the equation, the decoder output is And the effect of E disappears. On the other hand, if you do not feed back the error that occurred in the above formula, From this equation and the above equation, the decoder output is Therefore, as a result, the effect of not returning E + e (n + 1) propagates.

As is clear from the results of the above consideration, it is better to feed back the error due to clipping as well as the normal quantization error. In this case, only the word that caused the overflow is E · G ⁻¹ · g ^{−1 in the} above equation.
Error is not propagated to subsequent words.

Next, the effect of limiting the amount of movement when the ranging position, that is, the requantization bit extraction position moves to the LSB side will be described.

First, feeding back the error due to clipping to the next word is shown in the above equation. The error returned in this equation is {e (n + 1) + E} · G
It is a term of ⁻¹ · g ⁻¹ , which can cause overflow in the word of n + 2. That is, when the requantization bit extraction position (ranging position) is rapidly moved to the LSB side as shown in FIG. 8B, However, Fs may have a full-scale size that can be represented by the quantized bits of the current block, and the error E in the word n + 1 cannot be completely sucked in the word n + 2, causing continuous overflow. It will end up. In this way, errors due to overflow propagate one after another, resulting in a large distortion. This state is shown in FIG. In FIG. 11, the first word in the block when the ranging position is rapidly moved to the LSB side is W _0, and the previous word (the last word of the previous block) is added to the data D ₀ of this word W _0. The quantization error (error) is superposed to obtain the true value P _0. This true value P ₀ exceeds the range of the full scale FS that can be represented by the requantization bit of the current block. Therefore, the output value Q ₀ is pasted to the positive maximum value by performing the clipping process. The quantization error E _{0 at} this time is the output value Q _{0 and the} true value P ₀ . An inverted version of the sign of this error E ₀ , namely −
E ₀ is superposed on the data D ₁ of the next word W ₁ and becomes the true value P ₁ , but since this true value P ₁ also exceeds the full scale FS and becomes overflow, the output value Q ₁ becomes Clipped to maximum positive value. In this way, the error propagates. Note that, in FIG. 11, the true value P ₀ of the block head word W ₀ with error superposition is set within several times the full scale for the sake of illustration, but it is actually fed back to W _0. The error can be about 1000 times full scale (about 60 dB) and the error propagation can persist for long periods of time.

On the other hand, as described above, when the amount of movement when the ranging position (requantization bit extraction position) moves to the LSB side is limited to, for example, 1 bit, as shown in FIG. The error fed back at the first word W ₀ is suppressed to a small level, and error propagation is eliminated in a short time. Therefore, the distortion of the output signal from the decoder can be suppressed to a small level, and in combination with the temporal masking effect, it is possible to perform signal transmission without any trouble in hearing.

The present invention is not limited to the above embodiment, but the sampling frequency of the input digital signal, the number of bits of one word, the number of words in one block, the highest order N and the number of types of filters, or requantization. Of course, the number of bits and the like can be set arbitrarily.

〔The invention's effect〕

According to the signal transmission device of the present invention, the peak value (maximum absolute value) in the block of the output from the selected filter is multiplied by the coefficient γ (γ ≧ 1), and the ranging position is calculated based on the multiplied value. That is, since the extraction position of the requantized bit is determined, overflow is less likely to occur even if a feedback error due to noise shaping processing is superimposed.

In addition, high-order difference PCM, first-order difference PCM, straight P
Since a plurality of differential processing filters that output each CM data are used and these are adaptively switched and selected, efficient bit rate reduction can be performed, and at an extremely low bit rate without degrading signal quality. It becomes possible to transmit the signal. Further, since the output mode of the straight PCM data can be switched and selected, it is possible to solve the S / N deterioration at the time of inputting a high frequency signal and the occurrence of excessive error power when an error occurs.

Furthermore, when the input signal level sharply drops near the block boundary and the ranging position, that is, the requantized bit extraction position is about to suddenly move to the LSB side, by limiting the moving amount, It is possible to suppress the error due to the overflow of the quantization bit. At this time, when the requantization bit overflows, clipping is performed at the maximum positive or negative value, and the error at this time is fed back to perform noise shaping processing to suppress error propagation and generate abnormal noise. It is possible to effectively prevent the adverse effects of the above.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block circuit diagram showing a schematic configuration of an entire system to which a signal transmission device according to the present invention is applied, and FIGS. 2 and 3 are encoder and decoder of FIG. FIG. 4 is a block circuit diagram showing a more specific configuration example, FIG. 4 is a graph showing frequency characteristics of a plurality of difference processing filters, FIG. 5 is a diagram showing an example of a transmission word configuration in one block,
FIG. 6 is a graph showing the frequency characteristics of a plurality of difference processing filters used in another specific example, FIG. 7 is a graph showing the spectral distribution of noise subjected to noise shaping processing, and FIG. 8 is a graph of requantization. FIG. 9 is a block circuit diagram showing the main part of the encoder, FIG. 10 is a block circuit diagram showing the main part of the decoder, and FIGS. 11 and 12 are requantization diagrams. It is a figure for explaining error propagation by overflow at the time of conversion. 10 ... Encoder 12, 12A-12D, 20, 34 ... Predictor 15, 19, 32 ... Shifter 16 ... Quantizer 17 ... Noise shaper 21 ... Prediction / range adaptation circuit 30 ... Decoder

Claims (1)

[Claims] 1. A unit for dividing an input digital signal into blocks by a constant number of words along a time axis, performing a prediction process on the signal in each block, and requantizing the predicted signal and quantizing the signal. Means for feeding back the quantization error and performing noise shaping processing, and the maximum absolute value in the block of the signal subjected to the prediction processing is multiplied by a coefficient γ of 1 or more, and based on the multiplication result, the requantization is performed. And a means for determining the requantization bit extraction position of the signal transmission apparatus.