JPH06123772A - Encoded pulse doppler radar system - Google Patents

Abstract

PURPOSE:To realize an encoded pulse Doppler radar system which can measure the distance and speed of a target over a wide range without involving any ambiguity under various clutter environments. CONSTITUTION:A modulator 2 phase-modulates a transmission pulse train with a code sequence of a predetermined length to form repetitive encoded pulse groups which are then amplified and transmitted through a transmitter 3. Clutter of output signal from a receiver 6 is suppressed through an MTI filter 8 for each pulse train having same sign in a series of encoded pulse groups and then distance and speed information of a target is extracted over a wide range by means of a code correlator 9 and a Doppler processor 10. This system can obtain distance and speed information of multiple targets in a short time with high accuracy under various clutter environments and, thereby, it is employed effectively in radar system.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention is a reflected wave from a ground surface, a sea surface, etc., which is received together with a target signal in order to efficiently and quickly acquire information on a target moving at various speeds in a wide range of space. The present invention relates to a coded pulse Doppler radar system that directly suppresses clutter and improves target detection performance, and then simultaneously and directly measures the range and speed of a target over a wide range without ambiguity.

[0002]

2. Description of the Related Art The basic purpose of radar is to measure the distance and / or velocity of a target and, if necessary, add an angle, a signal strength and the like to provide necessary target information. FIG. 9 conceptually shows a signal transmission / reception time waveform in the pulse radar. The maximum detection distance R _max at which ambiguity does not occur in the distance measurement is conditioned on that the delay time T _d until the target signal is received after the pulse transmission is smaller than the transmission pulse repetition period T,

## EQU1 ## R _max = 0.5 cT (c: radio wave propagation velocity, T: pulse repetition period). Therefore, the pulse repetition period T is determined according to the maximum detection distance R _max required of the radar.

On the other hand, the transmission / reception time waveform (FIG. 9) of the pulse radar is shown in FIG. 10 in the frequency domain. That is,
The frequency spectrum interval of the transmission signal is determined by the pulse repetition frequency (Pulse Repetition Frequency).
NCY: _PRF) so given by f r = 1 / T, the target signal Doppler frequency

[Number 2] _{f d = 2 (v / c} ) f o (v: target relative speed, _{f o} = transmitted frequency) _| f d | a <range of 0.5f _r, each frequency spectrum of the target signal There is no ambiguity in the measurement of the Doppler frequency because it can be determined which transmission frequency spectrum it corresponds to. This results in unambiguous R _max and f _d
Between the upper limit value | f _{d · max} | of

## EQU00003 ## A bireciprocal relationship based on R _max .vertline.f _d .max .vertline. = 0.25c is established.

FIG. 11 shows the relationship between the maximum distance of the target without ambiguity obtained from the above equation and the maximum relative velocity determined from the maximum Doppler frequency. For example, the transmission frequency 10
In the case of [GHz], if the target with a relative speed of 100 [m / s] exceeds a distance of about 11 [Km], ambiguity occurs and the distance cannot be directly determined.

As a result, in an anti-aircraft search radar or the like, which places importance on distance information and is required to detect a long distance, pulse repetition corresponding to the required maximum detection distance is performed in order to prevent occurrence of ambiguity in distance measurement. A low PRF in which the period T is easily set is used. On the other hand, ambiguity occurs in the velocity measurement, and it becomes difficult to determine the relative velocity based on the Doppler frequency.Therefore, it is necessary to have a method for calculating the approximate movement velocity from the amount of change in the target position vector obtained for each antenna beam scan. Applied accordingly.

On the other hand, when high precision measurement of a wide range of distances and velocities is required, the maximum Doppler frequency f _{d · max} of the target signal is set so that the relative velocity of the target can be directly measured without causing ambiguity. Depending on the high or medium PRF used. On the other hand, ambiguity occurs in distance measurement, so
To eliminate this, multi-PRF ranging (Multi
i PRF Ranginxg) method (as a principle explanation, for example, MI Skolnik: “Rada
r Handbook ", pp. 19-3, McGra.
w-Hill, 1970).

Conventional high or medium PRF using this method
FIG. 12 shows an example of the basic system configuration of the radar. In FIG. 12, a fixed number of modulated pulses having a fixed PRF generated by the pulse generator 18 are input to the modulator 19 to pulse-modulate a carrier wave to form a transmission wave. The transmission wave is power-amplified by the transmitter 20, and then transmitted from the antenna 22 via the transmission / reception switch 21.

Next, the reflected wave from the moving target, the ground surface, the sea surface, etc., passes through the antenna 22 and the transmission / reception switching device 21, and then the receiver 2
3 and converted into an intermediate frequency signal. The receiver 2
The output of No. 3 is input to the distance gate switch 24, controlled by the distance gate pulse input from the distance gate pulse generator 25, and sequentially output to the processing system for each distance gate (# 1 to #M). FIG. 13 shows a situation in which the reception signal is sequentially time-divided and taken out by the distance gate within the pulse repetition period and the transmission / reception operation in the time domain described above. The reception target signal and clutter corresponding to a series of transmission pulses are extracted with time division for each distance gate between transmission pulses.

Here, the processing system for each distance gate is
The clutter removal filter 26 that operates in the same manner, I (In-phase) that decomposes the received signal into orthogonal components in the baseband
e) and Q (Quadrature-phase) both phase detector 27 and A / D (analog / digital) converter 28.

First, the clutter removing filter 26 receives an intermediate frequency (f _IF ) signal that has passed through a distance gate at every pulse repetition period T. This signal is shown in FIG. 14 in the frequency domain. That is, each frequency spectrum forming the transmission wave (transmission pulse repetition frequency _fr
To avoid occurrence of the target signal spectrum (ambiguity in clutter spectrum than this distant spread around in the interval presence), _| f d | designed _{<0.5f r)} is.

[0011] The clutter elimination filter 26 removes the clutter component in the vicinity of the intermediate frequency _{f IF,} the Doppler frequency _{f d} extracts the target signal in the range of -0.5f _{r <f} d <0.5f _r , Output with continuous wave. The output signal of the clutter removal filter 26 is converted into mutually orthogonal baseband signals by the phase detectors 27 for both I and Q channels, and further digitalized via an A / D converter 28 that operates at every transmission pulse repetition period T. Converted to a signal.
The output of the processing system provided for each of these distance gates is sent to the next input data switch 29.

FFT (Fast Fourier Tr)
The Doppler processing unit 30 composed of a filter or the like obtains data for each distance gate number from the input data switching unit 29, and performs Doppler processing (separation and identification processing of Doppler frequency) by a narrow band Doppler filter bank. Then, the result is sent to the target detector 31. The target detector 31 determines the filter number and the distance gate number in which the target signal exists and sends them to the distance calculator 32.

In order to perform the multi-PRF ranging here, the operation processing from the transmission / reception of the radar signal to the target detector 31 is performed by a few pulse repetition frequencies (PRF).
The distance gate number of the target signal output from the same Doppler filter for each PRF is input to the distance calculator 32.

The PRF used in this case is one in which the number of range gates within the pulse repetition period (the number M of range gates shown in FIG. 13) is substantially close to each other and has a prime relationship. The distance calculator 32 calculates the true distance from the combination of the distance gate numbers, and outputs the target distance and speed information.

[0015]

[Problems to be Solved by the Invention] As described above, low P
In the radar using RF, ambiguity is likely to occur in the target velocity measurement by the Doppler frequency, and the range that can be directly measured becomes narrow.

On the other hand, a radar using high or medium PRF uses a multi-PRF ranging method for distance measurement. In this case, if noise or residual clutter mixes into the distance calculator, incorrect calculation of the distance occurs, and the distance measuring function of the system deteriorates sharply. To prevent this, a high clutter suppression capability and high S / N (peak) Signal power to average noise power ratio). Further, extra time is required to perform transmission / reception processing of a pulse group by 2-3 types of PRF, and S / N is not directly improved.

In addition, in order to avoid an erroneous calculation that occurs when a target pulse straddles two adjacent distance gates, it is necessary to further transmit and receive pulse groups of different PRFs. In addition, when a plurality of targets having substantially the same relative speed are at different distances in substantially the same direction, distance calculation becomes difficult.

As described above, it is difficult to directly measure the velocity of the high speed target in the case of the low PRF radar in the measurement of the distance and velocity by the conventional pulse radar, while in the case of the high or medium PRF radar, The above-mentioned problems resulting from the principle of multi-PRF ranging impose restrictions on the burden on the radar system configuration and efficient acquisition of target information.

[0019]

The present invention has been made paying attention to the fact that the above-mentioned problems associated with the conventional pulse radar are caused by the characteristics of the ambiguity of the radar waveform used. Is. That is, a radar waveform including a pulse train that is periodically phase-modulated with a code sequence is transmitted / received, and the detected and demodulated code sequence signal is first subjected to MTI processing for suppressing clutter and extracting a target signal. By sufficiently removing the clutter in the MTI processing in advance, it is possible to suppress the time side lobe noise level generated by the clutter in the subsequent signal processing process and prevent the level saturation in the signal processing. After that, the target correlation information in a wide range is extracted by performing the code correlation processing according to the long period of the code sequence, and at the same time, the high or medium PR is performed.
Narrow band Doppler processing is performed on the target signal pulse train obtained in F to extract speed information in a wide range. The coded pulse Doppler radar system is characterized by having the above functions.

The purpose is to effectively reduce obstacles due to clutter even when the moving target is buried in strong clutter, and to cover a wide range of distances depending on the long-term cycle of the code sequence and high or medium transmission pulses. An object of the present invention is to provide a coded pulse Doppler radar system capable of directly measuring a wide range of velocities according to PRF simultaneously with high accuracy and without ambiguity. Hereinafter, the present invention will be described in detail with reference to the drawings.

[0021]

1 is a basic configuration diagram of a system according to the present invention. In FIG. 1, the periodic code sequence signal of code length N generated by the code generator 1 is input to the modulator 2 as a modulation signal, and the transmission carrier is subjected to phase modulation to form a transmission signal,
After the power of the transmission signal is amplified by the transmitter 3, the transmission / reception switch 4
And transmit via antenna 5.

FIG. 2A shows an example of the structure of the modulated signal. That is, using a binary code sequence of code length N, the pulse repetition period T, the pulse width ΔT, and the time width Tc of one code sequence
The transmission pause times T _s1 15 and T _s2 16 are alternately provided between the pulse groups 12 of each code sequence formed in _{step S1} , and pulse trains of the same code (for example, S _{k. 1} , S _{k + 1.1} , S
_{k + 2.1} , ..., Time interval T ₁ (=) of adjacent pulses
T _o −δT) 13 and T ₂ (= T _o + δT) 14 are alternately applied to perform staggering for two pulses. Usually, 2 to 4 types of pulse intervals are used. The code sequence used for the above-mentioned modulation is assumed to have good autocorrelation and is generally not limited to binary but multi-valued.

Next, the received wave composed of the moving target and the clutter is input to the receiver 6 via the antenna 5 and the transmission / reception switch 4. Here, the received code sequence signal is decomposed into two orthogonal baseband signals through the phase detectors for both I and Q channels in the receiver 6, and then A / D converted and output. , To the signal processor 11.

FIG. 2B shows the output of the phase detector of the I or Q channel corresponding to each modulation code pulse (equivalent to the transmission signal) of FIG. 2A on the time axis. That is, the target signal is output with respect to the transmission signal with the delay time T _d according to the distance and the amplitude modulation by the Doppler frequency f _d . On the other hand, the radar wave reflection area on the surface of the earth or the sea surface usually exists in a wide range in the distance direction. Therefore, when transmitting and receiving at high PRF, clutter overlaps and intensifies, and its amplitude level is averaged in positive and negative, and its fluctuation. Becomes loose.

FIG. 3 shows a functional system diagram of the signal processor 11, and includes a stagger trigger delay circuit 7 for receiving the outputs of the phase detectors for both the I and Q channels and for making the pulse intervals of the same sign constant. , I and Q channels have the same operating characteristics, an MTI filter 8, a code correlator 9 and a Doppler processor 10.

First, with respect to the output of the phase detector shown in FIG. 2 (B), the stagger trigger delay circuit 7 of FIG.
₁ 13 is given a delay time δT, and time interval T ₂ 14 is given a delay time of 0 to give the same code (for example, S _k . 1, S _k .
_{k + 1.1} , ...) The time period of the pulse train is a constant value T
After being set to _o , N kinds of pulse trains of the same code having the same code length are sequentially input to the MTI filter 8 at intervals of time T. The MTI filter 8 can be configured not only in a non-recursive type but also in a recursive type, and the number of delay tap stages Q is usually 2 to 4.

FIG. 4 (A) shows the same code input to the MTI filter 8 in the case of non-staggered transmission / reception in which the time period of the pulse train of the same code is T ₁ = T _o −δT (constant). The frequency spectrum distribution of the pulse train and the frequency characteristic of the MTI filter 8 for suppressing clutter are shown. Similarly, FIG. 4B shows a non-staggered case in which the time period of the pulse train of the same sign is T ₂ = T _o + δT (constant). In these cases, the target signal spectrum is M
Since it may happen that it enters the stop band (blind frequency region formed for each PRF) of the TI filter 8, the Doppler frequency f _d is 0 to f _{d. When} it occurs uniformly at _max , the target detection probability is approximately 15 to 30 numbers [%] on average.
To some extent.

On the other hand, in FIG. 4C, the above T ₁ and T
Shows the case of two pulses between staggered MTI process of switching ₂ alternately. By performing the stagger between pulses, the ratio of the stop band of the MTI filter 8 can be reduced equivalently to about several [%], so that the reduction of the target detection probability can be effectively suppressed. (For the explanation of the principle of the inter-pulse stagger, see, for example, MI Skolnik: “Radar Han.
"dbook", pp. 17-38, McGraw-Hi.
ll (1970). )

Thus, the MTI of both the I and Q channels
Since the filter 8 suppresses the clutter spectrum component that exists for each PRF of the pulse train of the same code to be transmitted and extracts the target signal, the target signal that is relatively stressed is the Doppler contrasted with FIG. 2B. The signal is amplitude-modulated by the frequency and output, and then input to the next code correlator 9 together with the minute remaining clutter.

The code correlator 9 has N taps at intervals of a delay time T equal to the transmission pulse repetition period as shown in FIG. Each code (S _k.1 , S _k.2 , ... S _k.N ) of the code sequence is given to the integrator 17 of each tap as an integration coefficient, and the target signal and the remaining clutter input to each tap are added. Code correlation with the code pulse train is performed.

As a result, the delay time T _d depending on the target distance
The timing for the input code to the multiplier 17 in all taps are matched simultaneously output is amplitude modulated at the Doppler frequency the target signal pulses as shown in FIG. 5 over the entire tap, which is time period T _o Occur in. On the other hand, at the timing of non-coincidence of the codes, the random pulse is time sidelobe noise and is the repetition period T of the transmission pulse.
, And becomes zero at other transmission suspension timings.

Each tap output of the code correlator 9 is subjected to the Doppler frequency separation and S through the Doppler processor 10 composed of a narrow band filter bank such as an FFT filter.
/ N is improved. Where the Doppler frequency is | f _d
| <Target signal in the 0.5 / T is improved S / N from #r _s filter Doppler frequency f _d of the plurality of Doppler filters as shown in FIG. 6 are aligned, the delay time due to the distance of the target output in the time period _{T o} with a T _d. Further, the remaining clutter disappearance component is separated from the target signal and output from the # 0 filter. In addition to receiver noise, remaining side clutter and time sidelobe noise generated from the target signal itself are output from all Doppler filters.

By the way, when the received clutter is gentle, the clutter suppression process by the MTI filter 8 constituting the present invention can be omitted. An invention corresponding to this is Japanese Patent Laid-Open No. 61-212781 (name: pulse Doppler radar system). However, radar is generally operated in various clutter environments. When strong clutter is received, if the MTI filter 8 does not perform clutter suppression processing, a high level of time side lobe noise due to clutter occurs in the processing process of the code correlator 9, and a weak target signal detection performance is obtained. Lower.

For example, the target signal power to clutter power ratio (S / C) ≦ −30 [dB] at the phase detection output, the average time sidelobe level (TSL) = − 30 [dB] at the output of the Doppler processor 10, S / N required for target detection
= 10 [dB], the average TSL noise power ratio (S / N) of the target signal power to the clutter ≤0 [dB] at the output of the Doppler processor 10, and the required S / N cannot be satisfied. In addition, large amplitude clutter having a Doppler frequency of almost zero may saturate the FFT filter and impair normal operation. Even before saturation, the clutter component itself mixes into the FFT filter that detects the target from the frequency side lobe region, and degrades the signal detection performance. From the above, in order to operate the radar system effectively in various clutter environments, MT
Clutter suppression by the I filter 8 is essential.

The present invention attaches importance to the above problems and takes measures against them. That is, as illustrated in FIG. 2A, in a transmission waveform periodically composed of pulse groups corresponding to a constant code sequence, a stagger between pulses or a non-staggered radar waveform is generated for each pulse train of the same code. The received demodulated signal is subjected to clutter suppression processing by the MTI filter 8 for each pulse train of the same code.
By this method, the S / C before code correlation processing can generally be improved by about 20 to 50 [dB], so that the obstacle due to clutter can be greatly reduced. At this time, the effect of combining the inter-pulse stagger MTI processing is to reduce the number of appearances of the periodic blind frequency in the MTI filter 8 as shown in FIG. .

In the case of the present invention, the distance measurement range R without ambiguity is the transmission code sequence S _k shown in FIG.
The pulse group (shown in FIG. 2 (B)) of the reception code sequence corresponding to the above does not overlap with the pulse group of the next transmission code sequence S _{k + 1} .

## EQU4 ## It is represented by 0 <R <0.5 (N-1) cT, and is expanded to (N-1) times that of the conventional pulse radar having the transmission pulse period T. On the other hand, the measurement range of the Doppler frequency f _d is | f _d depending on the repetition period T of the transmission pulse.
| <0.5 / T, and after all, the upper limit values of R and f _d are

## _EQU00005 ## R _{max .} | F _d. The relationship of _max | = 0.25c (N-1) is established.

Therefore, even if an unnecessary component such as noise is mixed in the output of the Doppler filter, by performing signal detection for each distance gate, a wide range of distance can be measured with the accuracy of the distance gate width and the bandwidth of the Doppler filter. A wide range of speeds can be measured simultaneously and directly with accuracy according to. this is,
Conventional high or medium PRF with multi-PRF ranging
This is in contrast to the radar method.

[0038] By way of example designed by specific numerical values in its application to long distance radar, a design condition: radar frequency f _o = 3 [GHz] Maximum detection distance R _max = 500 [km] maximum relative velocity V _max = 700 [M / s] (f
_{d. max} = 14 [KHz]) b Design value: Code length N = 95 Transmission pulse repetition period T = 0.25c (1 / f _o V _max ) = 35.7 [μ
s] code sequence pulse group duration Tc = NT = 3.36 [ms] lower limit of the time interval in the pulse train of the same sign _{T 1 = 2 (N-1} ) T = 6.71 [ms] of the MTI filter Number of delay taps Q = 3 Stagger ratio between pulses: In the case of stagger between 4 pulses T ₁ : T ₂ : T ₃ : T ₄ = 12: 16: 13: 18 (n ₁ = 12, n ₂ = 16, n ₃ = 13, n ₄ = 18:
(See FIG _. 4C _. ) Doppler frequency is 0 to _{f d. max} (= 14 [KH
z]), the target undetectable band ratio reduction effect due to staggering between pulses when non-staggered: R _BO = 36 [%]
In case of stagger, target undetectable band ratio at the time of stagger: R _BS ≈ R _BO / n ₀
= 2.4 [%]. Here, n ₀ = (n ₁ + n ₂ + n ₃ + n ₄ ) = 14.7
5

The above-mentioned design conditions cannot be directly coped with by the conventional pulse radar according to the principle shown in FIG. 11, but according to the present invention, a design value capable of directly measuring the required wide range of distance and speed at the same time can be obtained. You can

[0040]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS As a first embodiment, FIG. 7 shows the configuration of a signal processor 35 in which a function is added to the signal processor 11 in the basic system configuration diagram shown in FIG. With respect to the configuration of the signal processor 11, a constant false alarm rate (CFAR: Constant False) for stabilizing the probability of erroneously recognizing noise as a target signal and ensuring the credibility of a detection signal.
An alarm rate processor 33 and a target signal detector 34 based on a threshold value are combined.

In FIG. 7, when the Doppler processor 10 is composed of a plurality of filter banks, the CFAR processor 33 processes each output of each Doppler filter with a sample value on the time axis, or the same quantization time. For each, multiple Doppler filter outputs are processed as sample values. The latter method is CF
There is an advantage that the AR processor 33 can be configured in a small scale. In addition,
As the Doppler processor 10, it is possible to use a Doppler tracker that automatically tracks the Doppler frequency.

FIG. 8 shows a signal processor 3 as a second embodiment.
8 shows the configuration of No. 8. In order to improve the resolution of the Doppler frequency and the S / N in the first embodiment shown in FIG. 7, the Doppler processor 10 is made a two-stage process, and each filter output of the first-stage Doppler processor 10-1 is In addition, the second stage Doppler processor 10-2 further performs narrow band Doppler processing. The processing method of the CFAR processor 36 and the target signal detector 37 is the same as that of the signal processor 35.

[0043]

As described above, according to the present invention, a radar waveform in which a pulse having a high or medium repetition frequency is periodically encoded with a code sequence is transmitted and received, and a clutter suppression process utilizing the periodicity of a pulse train of the same code, By performing code correlation processing with a long period and Doppler processing over a wide range of frequencies, simultaneous and direct measurement of a wide range of distances and velocities is possible not only for a single target but also for multiple targets under various clutter environments. Therefore, it is possible to overcome the problems in principle associated with the conventional pulse radar system. As a result, an efficient configuration of the radar system is possible, and the system can be operated more effectively by acquiring the target information with high accuracy and in a short time.

[0044]

[Brief description of drawings]

FIG. 1 is a system basic configuration diagram of a coded pulse Doppler radar for implementing the present invention.

[Figure 2 (A)]

FIG. 1 is a modulation signal for performing inter-pulse stagger in the system of FIG.

FIG. 2B is a time domain waveform of the output from the phase detector in the receiver 6.

FIG. 3 is a functional system diagram of a signal processor 11.

FIG. 4 (A) and

FIG. 4B is a diagram showing a frequency spectrum of an input signal to the MTI filter 8 and a characteristic of the MTI filter 8 when the stagger is not performed.

FIG. 4C is a diagram showing a frequency spectrum of an input signal to the MTI filter 8 and a characteristic of the MTI filter 8 when staggering between two pulses.

FIG. 5 is a diagram showing each tap output when the code correlator 9 has the same code correlation.

6 is a diagram showing each filter output of the Doppler processor 10. FIG.

FIG. 7 is a functional system diagram of a signal processor 35 in the first embodiment.

FIG. 8 is a functional system diagram of a signal processor 38 in the second embodiment.

FIG. 9 is a diagram showing a principle of occurrence of ambiguity in distance measurement by a conventional pulse radar.

FIG. 10 is a diagram showing a principle of ambiguity in speed measurement by a conventional pulse radar.

FIG. 11 is a diagram showing a distance and velocity measurement range without ambiguity in a conventional pulse radar.

FIG. 12 is a diagram showing an example of a basic system configuration of a pulse Doppler radar using a conventional high or medium PRF.

13 is a diagram showing the passage of a distance gate in the distance gate switch 22 of FIG.

FIG. 14 is a diagram showing the operation principle of the clutter removal filter 23 in FIG.

[0045]

[Explanation of symbols]

1 code generator 2 modulator 3 transmitter 4 transmission / reception switcher 5 antenna 6 receiver 7 stagger trigger delay circuit 8 MTI filter 9 code correlator 10 Doppler processor 11 signal processor 12 each code sequence having a time width T to occupy ( S _k , S
_{k + 1} , ...) Pulse group 13 Time interval of same-code pulses between adjacent code sequences T ₁ = T _o −δT 14 Time interval of same-code pulses between adjacent code sequences T ₂ = T _o + δT 15 Transmission pause Time T _s1 16 Transmission pause time T _s2 17 Accumulator 33 C in the signal processor 35 according to the first embodiment
FAR processor 34 Target signal detector in signal processor 35 36 C in signal processor 38 according to the second embodiment
FAR processor 37 Target signal detector in signal processor 38

[Equation 1]

[Equation 2]

[Equation 3]

[Equation 4]

[Equation 5]

Claims (1)

[Claims] 1. A modulator that forms a transmission waveform of repetitive pulses, a transmitter that performs power amplification of the transmission waveform, and an antenna that transmits an output signal of the transmitter and receives a reflected wave from an object. In a pulse radar provided with a receiver for detecting a received signal output from the antenna and a signal processor for processing the output signal of the receiver, (a) the modulator (2) uses a code sequence having a code length N. Is periodically used to perform phase modulation on each code corresponding to a series of repetitive pulses, to form a transmission pulse train of encoded time period T. In this case, each encoding corresponding to the code sequence is performed. Two or more different transmission pause times T _si (≧ (N−1) T, i ≧) between the pulse groups
2) is sequentially repeated to perform stagger between pulses by sequentially changing the pulse intervals of the same code between the adjacent coded pulse groups, or by setting the transmission pause time T _si = constant to make non-stagger. ) When the pulse train of the same code in the periodic code sequence signal demodulated by the receiver (6) forms the inter-pulse stagger, the stagger trigger delay circuit 7 and the MTI (Moving Targ) of the signal processor (11).
et Indication: moving target detection) through the filter (8) to perform stagger MTI processing, while in the case of the non-stagger, the non-stagger M through the MTI filter (8).
Performing TI processing, (c) the code correlator (9) of the signal processor (11) has N taps at each delay time equal to the transmission pulse repetition period T, and each tap has an integrator (17). And the N integrators (17) have MT
The output of the I filter (8) is sequentially received, the code correlation (integration) processing with the code sequence signal used for the phase modulation is simultaneously performed, and the respective results are output in parallel, and the (d) signal processor (11) The Doppler processor (10) receives the tap output of the code correlator (9) and performs narrow band Doppler processing. A coded pulse Doppler radar system having the above functions.