JPH0584159B2 - - Google Patents

Description

[Detailed description of the invention]

[Industrial Application Field] The present invention relates to a control device for an induction motor using the circular approximation method, and in particular to an improvement in AC voltage detection means when controlling the voltage and magnetic flux of a three-phase induction motor using a high-speed digitally controlled PWM inverter. It is related to. [Prior Art] In a system that supplies power to a three-phase induction motor using a three-phase bridge PWM inverter composed of high-speed switching elements, the primary magnetic flux of the motor is calculated by integrating the motor terminal voltage, that is, the inverter output voltage, and the magnetic flux is Techniques for performing follow-up control to command values are disclosed in the following papers and the like. 1 Institute of Electrical Engineers of Japan study group material RM-84-76 (Sho 59-9)
"High-speed torque control method for induction motors based on new theory" 2. Transactions of the Institute of Electrical Engineers of Japan 60-B61 (1986-6) "Magnetic flux control type real-time processing PWM control using a one-chip microcomputer" These papers detect motor input voltage. The motor magnetic flux is obtained by integrating this within the control circuit. That is, it is a so-called magnetic flux calculation type control system, and the greatest goal is to match the calculated magnetic flux and the actual magnetic flux as much as possible. The conventional technology related to the magnetic flux calculation method will be explained below. Figure 2 is a main circuit diagram of an example of a conventional PWM inverter.
a, 3-phase PWM inverter 2 via negative bus 1b
Power is supplied to the 3-phase induction motor 3 via the 3-phase induction motor 3. The control circuit 4 processes the commands and detected information 42;
An energization signal 41 for the switching element of the PWM inverter 2 is generated. PWM inverter 2 is transistor Q1~Q
6. It is composed of six arms each having diodes D1 to D6 connected in antiparallel, and these transistors can be replaced with other high-speed switching elements such as GTO or SI thyristor. Also, 2a, 2b, 2c are PWM inverter 2
These are output terminals for AC phases a, b, and c, and power is supplied from these output terminals to the three-phase induction motor 3 via current detectors 5a, 5b, and 5c, and from each output terminal to voltage detectors 6a, 6b, and 6c is Y-connected,
Each phase current and each phase voltage can be detected. In such a system, the most direct method to calculate the motor flux is to integrate the voltage across the terminals of the three-phase induction motor 3.
That is, as an example, the main points of the arithmetic expression detailed in the paper 1) are as follows. If the primary terminal voltage and current of a three-phase squirrel cage induction motor are v ₁ and v ₁ , respectively, and the secondary current is i ₂ , then the voltage equation is v ₁ → 0 → = R ₁ + pL ₁₁ pM (p−jθ〓 m) M R ₂ + (p-jθ〓m) L ₂₂ i ₁ → i ₂ → ... However, the symbols v ₁ , i ₁ , i ₂ are the direct axis and the horizontal axis, that is, the d and q2 axes, are the transformed quantities. It is a vector display,
For example, v ₁ is expressed as v ₁ →=v _1d +jv ₁ q, where v _1d is the d-axis component and v ₁ q is the q-axis component, and i ₁ →, i ₂ → are similarly defined. In addition,
0 on the left side of the equation represents the case where both d and q axis components are 0→, and in the case of a squirrel cage rotor, the secondary voltage is 0→
becomes. The constants in the equation are R ₁ ; Primary winding resistance L ₁₁ ; Primary inductance R ₂ ; Secondary winding resistance L ₂₂ ; Secondary inductance M ; Mutual inductance θ〓m is each rotation speed, p is a differential operator, j represents a vector product. On the other hand, as a definition of magnetic flux, the primary magnetic flux ₁ → is ₁ →=L ₁₁ i ₁ →+Mi ₂ →... Expanding the first line of the equation, v ₁ →=(R ₁ +pL ₁₁ )i ₁ →+pMi ₂ → Substituting and rearranging the equation, v ₁ →−R ₁ i ₁ →=p ₁ → ... Integrating both sides, we get ₁ →=∫(v ₁ →−R _i i ₁ →) dt ... In other words, the motor first order The magnetic flux is obtained by integrating the equation. FIG. 3 is a diagram showing the high-speed switching elements of a PWM inverter replaced with contacts, and the same reference numerals as in FIG. 2 indicate the same parts. The switching elements of a voltage source inverter are often expressed using two-position switching contacts Sa, Sb, and Sc. Each of the switching contacts Sa, Sb, and Sc may fall toward the positive bus line 1a side or fall toward the negative bus line 1b side, and never take an intermediate position. Assuming that the former is in state 1 and the latter is in state 0, the output states of the inverter can all be expressed in the switch state variable table shown below.

[Table] Here, k is a number indicating each switching contact state, and there are only eight of these. Also, , is d,
A switch state variable expressed as a q2-axis component, and the actual d- and q-axis voltages v _1d and v _1q are added to the voltage Vdc of DC voltage source 1.

Multiply by [formula]

It can be expressed as [C]. FIG. 4 is a diagram illustrating the above-mentioned switch state variable table, and represents a voltage vector that steps in steps of 60° in the time direction as k increases. Note that k=0 and k=7 are called zero vectors, and correspond to the origin in the figure. k=0 and k=7 are the outputs of the inverter, respectively.
There is a difference in whether the switching contacts Sa, Sb, and Sc in the figure all fall to the positive bus 1a side or to the negative bus 1b side, but the line voltage of the induction motor 3 is all 0, and the three-phase short circuit mode It is. Also, a,
The reference axes of the b and c phases correspond to the directions of k=1, k=3, and k=5, respectively, according to equations described later. FIG. 5 is a block diagram showing an example of the internal configuration of the control circuit 4, where 4a is a microprocessor and 4b is a microprocessor.
ROM memory, 4c is an input/output port, 4d is an address bus, and 4e is a data bus. The signal 41 output to the input/output port 4c is a signal that drives the bases of the transistors Q1 to Q6, and normally the signal 41 and the transistors Q1 to Q
An amplification circuit is installed between 6 and 6 for both insulation and current amplification, but it is omitted in this drawing. A group of instructions for controlling the system is stored in the ROM memory 4b, and the microprocessor 4
A sequentially executes instructions and performs necessary calculations and communication with the outside via the input/output port 4c. In addition,
The internal RAM of the microprocessor 4a is used to temporarily store the calculation results. An example of a command group necessary for control is shown in the flowchart of FIG. The blocks will be explained one by one below. Starting from 401, initial values necessary for the calculation are set in 402, and the calculation is executed from the program point P1. 403 is a block that inputs information necessary for control from the outside through the input/output port 4c, and includes a frequency command f ^* for the inverter, a motor magnetic flux command ₁ , and an inverter output current ia,
The detected values of ib and ic are sent to the current detectors 5a, 5b, and
Read from 5c. 404 is a waiting time Td corresponding to a energization prohibition time to prevent a short circuit between the upper and lower arms due to the accumulation time of the transistor. An actual base drive signal to the transistor is not generated until a waiting time Td elapses after an energization signal corresponding to an inverter output voltage k based on a calculation result to be described later is output to the input/output port 4c. A time delay generation circuit that delays the base drive signal by a waiting time Td is provided in a transistor base amplifier circuit (not shown). The waiting time Td is the time from the program point P3 at which the inverter output voltages va, vb, and vc are read from the input/output port 4c again to the program point P2 at which the inverter output voltages va, vb, and vc are read from the input/output port 4c,
Other processing times such as block 403 may be included. 405 is a block for reading inverter output voltages va, vb, and vc by voltage detectors 6a, 6b, and 6c. 406 represents the voltages va, vb, vc as d,
This is a block that converts to the q2 axis component, using a well-known conversion formula. The same conversion is performed for voltages ia, ib, and ic to obtain id and iq.

407 is a block that corrects the voltage drop in the resistor due to the primary current in order to perform the integration of the equation, and calculates the d and q components of the voltage effective for generating the magnetic flux of the motor using the equation. This is called an AC output voltage equivalent signal. V _1d ′=V _1d −R ₁ i _1d V _1q ′=V _1q −R ₁ i _1q ... Block 408 calculates the magnetic flux vector ₁ →, and the value of the magnetic flux vector ₁ → in the previous calculation cycle The product of the voltage value according to the formula and the calculation cycle time Δt is added to . Δt is a time equivalent to the calculation cycle time, and does not need to be equal to the calculation time, but it is necessary to have a size proportional to the calculation time. ₁ is the length of the magnetic flux vector, and is obtained by calculating the square root of the sum of squares of the d and q2 axis components ₁ d and ₁ q of the magnetic flux vector ₁ →. Block 409 calculates the phase command θ ^* from the frequency command f ^* according to θ ^* =2πf ^* t (t is time). Block 410 has d and q2 axis components _1d and _1q of magnetic flux.
This is a block that detects the phase θ from θ=cos ^-1 _1d /√ _1d ² + _1q ² .... The calculation of cos ^-1 is based on a function table, but the details are not directly related to the essence of the present invention, so they will be explained below. is omitted. The magnetic flux vector ₁ is d, q as shown in Figure 6.
It is controlled to draw a circular locus on a plane. In Figure 6, the magnetic flux command vector is expressed as ₁ → ^* ,
Its length is the magnetic flux command read in block 403.
₁ ^* , the angle with the d axis is θ ^* calculated in block 409. The controlled magnetic flux vector ₁ → follows the above-mentioned magnetic flux command vector ₁ → ^* , and is controlled so that its tip is always held within an annular ring having a width Δ. Here Δ
is the permissible error range of magnetic flux ₁ . In block 411, it is determined whether the flag set in the previous calculation cycle is fg=1 or fg=-1, and the process proceeds to either 412 or 412a. 41
2,412a has a magnetic flux vector length ₁ of 403
This block determines whether or not the magnetic flux command ₁ ^* read in exceeds the upper and lower limits, respectively, so as to draw a limit cycle as shown in FIG. When the upper limit ₁ ^* is exceeded, block 413a sets the voltage control flag to fg=-1 and commands magnetization. When it becomes smaller than the lower limit ₁ ^* -Δ, block 413 sets the voltage control flag to fg=1 and commands demagnetization. The flag remains set in the previous calculation cycle until the length ₁ of the magnetic flux vector exceeds the upper or lower limit. Blocks 414 and 414a are for determining whether or not the phase θ has overtaken the command phase θ ^* . If θ>θ ^* , the block 415a
The zero vector of k=0 or k=7 explained in the figure is output and the process proceeds to block 416. When the inverter outputs a zero vector, both the phase θ and the length ₁ of the magnetic flux vector remain unchanged. If θ≦θ ^* , block 415 searches the PWM signal table shown below, and block 41
Proceed to step 6.

[Problem that the invention seeks to solve]

When the inverter output voltage changes as a result of the voltage control described above, for example, the a-phase and b-phase remain in their previous states, but when the switch state variable of the c-phase changes from 1 to 0, that is, the transistor Q5 Consider when Q2 switches from conductive to non-conductive and transistor Q2 switches from non-conductive to conductive. The base signal of transistor Q5 can be immediately disappeared, but it is transistor Q5 that sends the base signal of transistor Q2 instead.
You have to wait at least the time it takes for it to turn off completely. In consideration of variations in transistor characteristics, a predetermined margin is added to this, and the energization prohibition time becomes a fixed waiting time Td. Therefore, a normal voltage will not be generated at the inverter output until the waiting time Td elapses from program point P3 in FIG. This is the waiting time Td
resulting in a time delay and a decrease in the inverter output voltage during that period. In fact, as described above, in order to simplify programming, a time delay generation circuit for delaying the energization signal by the waiting time Td is provided in the base amplifier circuit (not shown) of the transistor. Figure 9 is a diagram showing the influence of waiting time Td on the waveform, where a is the input/output port 4c for c-phase control.
It is a waveform that occurs at the output point 2c of the c phase if possible.
This is the waveform that we want to occur. b is the waveform actually generated at the output point 2c of the c phase due to the delay of the transistor, and c is the waveform generated at the voltage detector 6c. Although the DCPT used for the voltage detector 6c is expensive, it has low frequency characteristics. There are many cases where the waveform is poor, and the waveform generally has a sluggish rise as shown in the figure. Moreover, with digital control using a microprocessor program as shown in FIG. 8, even if a voltage with a waveform as shown in FIG. 9c is detected,
It can only be observed for a moment shown in block 405 of the instruction cycle, and cannot provide accurate information. [Means for Solving the Problems] Fig. 1 is a main circuit diagram of a PWM inverter to which the present invention is applied, and Fig. 10 is a flowchart of an embodiment of a command group of a control circuit corresponding to this. The same reference numerals as in FIGS. 2 and 8 indicate the same parts. Comparing Fig. 1 and Fig. 2 and mentioning only the differences, the voltage detector 6 on the AC side of the PWM inverter 2
a, 6b, and 6c are removed, and instead a voltage detector 7 is inserted between the positive bus 1a and the negative bus 1b on the DC side. Also, in the flowchart of the instruction group, the first
In Figure 0, the inverter output voltage vector v ₁ → is calculated using the formula, and the voltage Vdc of the DC voltage source 1 detected by the voltage detector 7 and the switch state variable,
Calculated from vQ and program points P
A block 421 is provided in the process of returning from program point 3 to program point P1, and after the inverter voltage vector v ₁ → stored this time is stored as the previous one, a new program cycle begins. [Operation] The operating principle of one embodiment of the present invention will be explained with reference to the flowchart of FIG. Block 4 the switch state variable of the output voltage output by the inverter this time.
Remember it at 21. Based on this switch state, block 404
At step a, the switch state variables are read with reference to the switch state variable table, and block 405
Using the voltage Vdc of the DC voltage source 1 detected by the voltage detector 7 at step a and the formula, v ₁ d, v ₁ q, which are the d and q axis components of the inverter output voltage to be actually detected, are calculated. Obtainable. Since there is no sudden change in the voltage Vdc of the DC voltage source 1, even a DCPT whose frequency characteristics are not very good can be used satisfactorily, and only one DCPT is required. In this way, there is no need to read the inverter's phase output voltages va, vb, and vc from the input/output port 4c.
Moreover, since the waiting time Td is known, there is no need to set the transistor energization prohibition time as the waiting time Td. Therefore, accurate information will be obtained faster. The voltage drop caused by the primary voltage i ₁ → that occurs across the primary resistor R ₁ in block 407 allows processing of the AC output voltage equivalent signal in the same manner as in FIG. The magnetic flux calculation in block 408a is performed by replacing Δt, which is the time component of the increment, with (Δt−Td).
As shown in FIG. 9b, the reduction in the inverter output voltage due to the influence of the waiting time Td is taken into consideration. This is to set the AC output voltage equivalent signal to 0 during the waiting time Td. Blocks 406a and 406b in FIG. 10 are for further perfecting the effect of the present invention, and when there is no change in the energization signal to each transistor output to the input/output port 4c during the instruction cycle, Since no transistor energization prohibition time occurs, the waiting time Td is set to 0 to prevent the voltage waveform from decreasing. That is, block 406a compares the inverter voltage vector v ₁ → in the previous instruction cycle with the current inverter voltage vector v ₁ →, and if there is no change, block 406b sets the waiting time Td to 0, and
The AC output voltage equivalent signal only during the actual transistor energization prohibition time is set to 0. [Effect of the invention] In a system that digitally controls the voltage or magnetic flux of an induction motor driven by a PWM inverter, the voltage detection means is simplified,
Utilizing the memory function of digital control, it is possible to accurately detect inverter output voltages that often pulsate, and it also reduces calculation time, making it especially suitable for high-speed sampling control where instantaneous values are a problem. It is possible to provide the following equipment. Further, although the present invention has been described using primary magnetic flux control of a three-phase squirrel cage induction motor as an example, it can be applied to many control devices that accurately control the AC output voltage of a PWM inverter.

[Brief explanation of the drawing]

Figure 1 is a main circuit diagram of a PWM inverter to which the present invention is applied, Figure 2 is a main circuit diagram of an example of a conventional PWM inverter, and Figure 3 is a main circuit diagram of a PWM inverter in which the high-speed switching elements are replaced with contacts. The circuit diagram, Fig. 4 is a vector diagram illustrating the switch state variable table, Fig. 5 is a block diagram showing an example of the internal configuration of the control circuit, Fig. 6 is a magnetic flux vector diagram, Fig. 7 is a limit cycle diagram, Fig. Figure 8 is a flowchart of an example of a group of commands necessary for conventional control, Figure 9 is a graph showing the effect of waiting time on the output voltage according to the flowchart of Figure 8, and Figure 10 is the output voltage according to the present invention. 7 is a flowchart of an example of a group of instructions for a control circuit when a detection means is used. 1...DC voltage source, 2...PWM inverter,
3...Induction motor, 4...Control circuit, 5a to 5c
...Current detector, 6a-6c, 7...Voltage detector, Q1-Q6...Transistor, D1-D6...
...Diode, Sa~Sc...Switching contact.

Claims (1)

[Scope of Claims] 1. It comprises at least a DC voltage source, a three-phase voltage type PWM inverter, a three-phase induction motor, and a DC voltage source voltage detector, and the control circuit includes a microprocessor and a DC voltage source voltage detector.
It comprises at least a ROM memory, an input port, and an address bus and a data bus that connect these chips, and supplies a three-phase induction motor from a DC voltage source via the PWM inverter according to an instruction program stored in the ROM memory in advance. In a device that supplies a phase AC voltage and controls a PWM inverter output voltage so that a motor magnetic flux signal calculated from a signal equivalent to the three-phase AC voltage draws an approximate circular locus, one sample outputted by the microprocessor is used. The multiplicative product of the switch status signal representing the previous output voltage of each phase, the time obtained by subtracting the energization prohibition time due to the switch status change of the PWM inverter from the switch status signal output time, and the output of the DC voltage source voltage detector. An output voltage detection device for a PWM inverter, characterized in that the motor magnetic flux is calculated as a PWM inverter AC output voltage equivalent signal. 2. Before calculating the magnetic flux, compare the inverter voltage vector in the previous command cycle with the current inverter voltage vector, and if there is no change in the comparison, set the energization prohibition time of the switching elements of the PWM inverter to 0, and then change the AC current of the PWM inverter. An output voltage detection device for a PWM inverter according to claim 1, wherein the output voltage detection device is configured to calculate an output voltage equivalent signal.