JPH0568379A - Power supply circuit - Google Patents

Abstract

PURPOSE:To stop oscillation in a short time by preventing hunting of self- exciting switching inverter type power supply circuit when a starting circuit is additionally provided. CONSTITUTION:Voltage of a capacitor C2, being charged with a DC power supply 1, is applied onto main switching transistors 25, 26. The main switching transistors 25, 26 are self oscillated through positive feedback operation of the combination of windings 20, 21, 22, time constant circuits 27, 28, and auxiliary switching transistors 29, 30. Upon turn ON of a power switch SW1, self oscillation takes place through starting circuits 15, 15'. When the power switch SW1 is turned OFF to stop oscillation, capacitors 41, 41' are charged reversely but the voltages thereof are prevented from increasing due to voltage division through resistors 50, 50' and the main switching transistors 25, 26 are held in OFF state thus preventing resumption of oscillation.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a self-excited switching inverter type power supply circuit, and when a starter circuit is added, it prevents the phenomenon of repeated start-up after turning off the power supply switch and stops oscillation in a short time. It was made possible.

[0002]

2. Description of the Related Art As a self-excited inverter type power supply circuit, a power supply circuit disclosed in Japanese Patent Publication No. 3-1914 filed by the present applicant has been proposed. An example of the circuit configuration will be described with reference to FIG. This circuit is the circuit shown by U1 and U
The circuit indicated by 2 is alternately turned on and off to generate an alternating voltage in the winding 20. The windings 21 (feedback winding), 22 (feedback winding) and 20 (load winding) are wound on the same core of the transformer, and the windings 20 and 21 and the windings 20 and 22 are positive feedback, respectively. It is connected in the direction. The voltage + B, -B of the DC power supply 1 is the power switch SW1.
Is applied to the power supply capacitors C2 and C2 via.

In the circuit U1, the transistor 25 is a main switching transistor, the transistor 29 is an auxiliary switching transistor, and the capacitor 27C is a capacitor forming a time constant circuit 27 together with a resistor 27R. The diodes 11 and 12 are for operating the transistor 25 in a non-saturated state. That is, the winding 21
When a voltage is induced on the transistor, it is applied to the base of the transistor 25 via the diode 11 and turns it on.
Assuming that the voltage drops of the diodes 11 and 12 are equal, if the collector-emitter voltage V _{CE of} the transistor 25 becomes equal to the base-emitter voltage V _BE , the diode 12
Turns on, and excess current flows between the collector and emitter of the transistor 25, leaving the current necessary to maintain V _CE = V _BE . This V _CE is always kept in the V _CE ≧ V _BE. Normally, the saturation voltage V _CE of the transistor is smaller than the base-emitter voltage V _BE , so that the transistor 25 is prevented from being saturated. The circuit U2 is also configured similarly to the circuit U1.

The circuit of FIG. 2 operates as follows. Circuit U
When 1 is on, the positive feedback action of the windings 21 and 20 keeps the transistor 25 on. Capacitor 2
The voltage of 7C rises with time with a time constant of 27R × 27C, and after a predetermined time, the transistor 29 is turned on and the transistor 25 is turned off. When the transistor 25 is turned off, the voltage applied to both ends of the winding 20 is inverted by the self-induction of the winding 20, and the circuit U2 side is turned on this time.

It takes a certain period of time from when the transistor 25 is turned off until when the transistor 26 is turned on. During this time, the transistor 25 must be kept in the off state. Fortunately, this is ensured for the following two reasons. One is that the transistor 29 remains on for a while due to overcharge accumulated in the capacitor 27C during the time from when the transistor 29 is turned on to when the transistor 25 is turned on (off time of the transistor 25). One is due to the off time of the transistor 29 itself (the transistor 29 is completely saturated and switched, and it takes some time to turn off). By turning on the transistor 26 within the time period during which the off state of the transistor 25 is held, stable operation becomes possible. Once the transistor 26 is turned on, the transistor 25 has 0 bias and continues to be turned off.

After a fixed time with a time constant of 28R × 28C,
From now on, the transistor 30 will be turned on and the transistor 26 will be turned off. In this way, the oscillation mode is formed and operates as a switching inverter.

The circuit of FIG. 2 cannot be self-started as it is. An example in which a starting circuit is added is shown in FIG. This is a drawing figure of Japanese Patent Application No. 3-166383 related to the applicant's application.
It has been proposed to 0.

In FIG. 3, a starting circuit 15 including a resistor 40, a capacitor 41 and a diode 42 is added to the base circuit of the main switching transistor 25. The resistor 40 has a high resistance because it causes a power loss, and the capacitance of the capacitor 41 is increased for strong starting.

When the power switch SW1 is turned on, the winding voltage of the transformer is zero and the charge charged in the capacitor 41 is also zero. The current flowing through the resistor 40 is small, and while passing through the resistor 35 while charging the capacitor 41, the winding 21
Flowing to the point B, but the current is not so large as to cause a large potential difference in the resistor 35. Therefore, at the beginning, point B and C
Since the points and the points D have almost the same potential, the main switching transistor 25 is also off. The capacitor 41 is charged with time, and when the potential difference between the points B and D reaches the base-emitter forward voltage of the main switching transistor 25, the transistor 25 enters the active region. At this time, a forward / return loop is formed from the return winding 21 through the resistor 35 and the capacitor 41 to the base of the transistor 25. The positive switching turns on the main switching transistor 25 at an accelerated rate, and the winding 21 causes the resistor 35 to pass. Continues to turn on due to the large base current flowing through it. At this time, since the current flowing from the resistor 35 to the capacitor 41 is much larger than the current reversely charged from the resistor 40, the potential of the capacitor 41 becomes the forward potential of the diode 42, and after the startup, the forward potential of the diode 42 is maintained. Fixed. Further, after the start-up, the diode 42 forms and maintains the forward and return loop.
Therefore, after the start-up, the oscillation is maintained and the charging current from the winding 21 always becomes dominant, and the minute reverse charging current from the resistor 40 becomes equal to nothing, and the resistor 40 turns off the main switching transistor 25. There is no accidentally turning on at the right time.

[0010]

The starting circuit 1 shown in FIG.
According to 5 (15 '), when the power switch SW1 is turned off, it operates as follows. Even if the power switch SW1 is turned off, since the electric charge is stored in the capacitor C2, the oscillation continues as it is.

Since the replenishment of electric charge from the DC power supply 1 to the capacitor C2 is stopped, the voltage of the capacitor C2 (potential at point A) gradually decreases, and the voltage induced in the winding 21 also decreases.

When the voltage applied between the base and emitter of the main switching transistor 25 becomes lower than its forward voltage (about 0.6V) due to the voltage induced in the winding 21, the main switching transistor 25 can no longer be turned on. Therefore, the oscillation stops.

Since the electric charge remains in the capacitor C2 even when the oscillation is stopped, the discharge current of the capacitor C2 is the resistance 40 → the capacitor 41 → the resistor 35 → the winding 21 → the winding 2
After that, the capacitor 41 gradually reversely charges. Since the oscillation is stopped, the reverse charge canceling current from the winding 21 cannot occur.

When the capacitor 41 is reversely charged with time and the potential difference between the points B and D reaches the base-emitter forward voltage of the main switching transistor 25, the transistor 25 enters the active region again. At this time, the voltage induced in the return winding 21 itself is low, but since it is in series with the voltage of the capacitor 41, a forward return loop is formed and the main switching transistor 25 is turned on at an accelerated speed to restart oscillation.

When the oscillation restarts, the current from the winding 21 charges the capacitor 41 in the forward direction with respect to the diode 42. Since the voltage induced in the winding 21 is low, the base-emitter voltage of the main switching transistor 25 drops below its forward voltage, which transistor 25 can no longer turn on and stops oscillating.

However, when the oscillation is stopped, the capacitor 40 is reversely charged again through the route of the resistor C → the capacitor 40 → ... From the capacitor C2, and then the oscillation is restarted again.

As described above, the starting circuit 15 (1
According to 5 '), even after the power switch SW1 is turned off, A
Repeat the oscillation and stop until the potential at the point becomes sufficiently small,
There was a drawback that the oscillation could not be stopped immediately. When viewed as a device, the power lamp or the like blinks even after the power is turned off, and further, the frequency of oscillation decreases, so that the transformer or the like makes an abnormal noise at the end. This invention is
It is an object of the present invention to solve the above-mentioned drawbacks of the conventional technology and provide an amplifier circuit that can stop oscillation in a short time after turning off a power switch.

[0018]

According to the present invention, a series connection circuit of a second resistor and a second capacitor is inserted between a power supply capacitor and a feedback winding, and the main switching transistor is turned off when the main switching transistor is off. A self-excited inverter type with a starting circuit consisting of a second time constant circuit that forms a current path from the capacitor to the load winding through the feedback winding and applies the voltage of the second capacitor to the base of the switching transistor. In the power circuit, the voltage dividing means divides the voltage of the power source capacitor between the second resistor and the second resistor and applies the voltage to the second capacitor.

[0019]

According to the present invention, the voltage of the power supply capacitor is divided by the voltage dividing means and applied to the second capacitor of the starting circuit. Therefore, the power switch is turned off to lower the voltage of the power supply capacitor. Then, when the oscillation is stopped, even if the second capacitor is charged, it is possible to prevent the voltage for turning on the main switching transistor from being reached, so that the restart of the oscillation can be surely prevented, or the restart of the oscillation can be completed completely. The number of restarts can be reduced, if not prevented, and oscillation can be stopped in a short time.

[0020]

FIG. 1 shows an embodiment of the present invention. FIG. 3
The same parts as those of the above are denoted by the same reference numerals. The power source in Figure 1 is U
The circuit indicated by 1 and the circuit indicated by U2 are alternately turned on and off to generate an AC voltage in the winding 20. Winding 21 (feedback winding), 22 (feedback winding), 20
The (load winding) is wound on the same core of the transformer, and the windings 20 and 21 and the windings 20 and 22 are connected in the positive feedback direction. DC power supply 1 voltage + B,
-B is a power supply capacitor C through a power switch SW1
2 and C2.

In the circuit U1, the transistor 25 is a main switching transistor, the transistor 29 is an auxiliary switching transistor, and the capacitor 27C is a capacitor that constitutes the time constant circuit 27 together with the resistor 27R. A starting circuit 15 including resistors 40 and 50, a capacitor 41, and a diode 42 is added to the base circuit of the main switching transistor 25. The resistor 40 has a high resistance because it causes a power loss, and the capacitance of the capacitor 41 is increased for strong starting. The resistor 50 is a resistor for voltage division. The circuit U2 is also configured similarly to the circuit U1.

The power supply circuit of FIG. 1 operates as follows. When the power switch SW1 is turned on, the winding voltage of the transformer is zero and the charging voltage of the capacitor 41 is also zero. The current flowing through the resistor 40 is small, and flows through the resistor 35 while charging the capacitor 41, flows through the winding 21, and reaches the point B, but it is not a current that causes a large potential difference in the resistor 35. Therefore, since the points B, C, and D are initially at substantially the same potential, the main switching transistor 25 is also off.

The capacitor 41 is charged with time,
The potential difference between the points B and D is the main switching transistor 25.
When the forward voltage between the base and emitter of the transistor is reached, the transistor 25 enters the active region. At this time, a positive feedback loop is formed from the return winding 21 through the resistor 35 and the capacitor 41 to the base of the transistor 25. The positive feedback causes the main switching transistor 25 to turn on at an accelerating speed.
It continues to turn on due to the large base current flowing from 1 through the resistor 35. At this time, the current flowing from the resistor 35 to the capacitor 41 is much larger than the current reversely charged from the resistor 40.
It becomes a forward potential of 2 and is fixed at the forward potential of the diode 42 after starting. Therefore, the main switching transistor 25 is not accidentally turned on by the resistors 40 and 50 at the off timing of the transistor 25 after startup.

When the circuit U1 is on, the winding 21,
The positive feedback action of 20 keeps the transistor 25 on. The voltage of the capacitor 27C rises with time with a time constant of 27R × 27C, and after a predetermined time, the transistor 2
9 is turned on and the transistor 25 is turned off. When the transistor 25 turns off, induction of the winding 20 causes the winding 20 to
The voltage applied to both ends of is inverted, and the circuit U2 side is turned on this time.

It takes a certain time from when the transistor 25 is turned off until when the transistor 26 is turned on. During this time, the transistor 25 is kept in the off state due to the overcharge accumulated in the capacitor 27C and the off time of the transistor 29 itself. There is. By turning on the transistor 26 within the time period during which the off state of the transistor 25 is held, stable operation becomes possible. Once the transistor 26 is turned on, the transistor 25 has 0 bias and continues to be turned off.

After a fixed time with a time constant of 28R × 28C,
From now on, the transistor 30 will be turned on and the transistor 26 will be turned off. In this way, the oscillation mode is formed and operates as a switching inverter.

When the power switch SW1 is turned off, it operates as follows. Even if the power switch SW1 is turned off, since the electric charge is stored in the capacitor C2, the oscillation continues as it is.

Since the supply of electric charge from the DC power supply 1 to the capacitor C2 is stopped, the voltage of the capacitor C2 (potential at point A) gradually decreases, and the voltage induced in the winding 21 also decreases.

When the voltage applied between the base and the emitter of the main switching transistor 25 becomes lower than its forward voltage (about 0.6V) due to the voltage induced in the winding 21, the main switching transistor 25 can no longer be turned on. Therefore, the oscillation stops.

Since the electric charge remains in the capacitor C2 even when the oscillation is stopped, the discharge current of the capacitor C2 is the resistance 40 → the capacitor 41 → the resistor 35 → the winding 21 → the winding 2
After that, the capacitor 41 gradually reversely charges. Since the oscillation is stopped, the reverse charge canceling current from the winding 21 cannot occur.

However, since the capacitor 40 is charged only to the voltage divided by the resistor 50 by this current path, this divided voltage (correctly, the voltage divided by the resistor 50 + the resistor 35) is the main switching. If the value is lower than the base-emitter forward voltage (about 0.6 V) of the transistor 25, the main switching transistor 25
Cannot be turned on and the oscillation will not resume. Therefore, the electric charge of the capacitor C2 is directly applied to the resistors 40, 50, 35,
It is gradually discharged through the windings 21 and 20.

As is apparent from the above, the values of the voltage dividing resistor 50 are set so that the resistors 40 and 3 satisfy the following (a) and (b).
If set in relation to 5 or the like, it is possible to prevent the restart of oscillation when the power switch 2 is turned off. (A) When the power switch SW1 is turned on from the off state, the capacitor 41 can be charged to the voltage + B, -B of the capacitor C2 to a voltage at which the main switching transistor 25 can be turned on. (B) When the power switch SW1 is turned off from the on state, the capacitor 41 is not charged to a voltage at which the main switching transistor can be turned on with respect to the lowered voltage of the capacitor C2 when oscillation is stopped for the first time.

Note that (b) is a condition for preventing the oscillation from being restarted even once. If the voltage divided by the resistor 50 is set to a voltage higher than this, oscillation will restart several times, but the effect of reducing the number of restarts can be obtained, and the voltage dividing resistor in FIG. Oscillation can be stopped in a short time as compared with the case of the absence. The voltage dividing resistor may be replaced with the resistor 50, and the resistor 51 may be inserted at the position indicated by the dotted line.

Next, the present invention will be described in Japanese Patent Application No. 3-166383.
An embodiment applied to the power supply circuit of No. 1 will be described. First, this power supply circuit will be described. This power circuit
By using both voltage resonance and current resonance to reduce switching loss to the limit and improving conversion efficiency, the operating waveform of each voltage and each current in the circuit was made closer to a sine wave to reduce noise. It is a thing.

As shown in FIG. 4, this power supply circuit includes a DC power supply 1 and a switching element that can be turned on and off at arbitrary timings, and switching means 2 for switching the input DC power supply to convert it into an alternating current and output it. When,
DC output means 3 for full-wave rectifying the supplied AC input and smoothing with a capacitor to obtain a DC output, series resonance means 4 formed in series with the current flowing through the output terminal of the switching means, and the switching Parallel resonance means 5 formed in parallel to the voltage produced at the output terminal of the means
And a timing control means 6 for controlling the switching element of the switching means to be turned on intermittently.

FIG. 5 is a basic principle block diagram showing the block of FIG. 4 a little more structurally. The operation of the basic principle configuration shown in FIG. 5 will be described with reference to FIG. 6 showing the operation timing of each unit.

In FIG. 5, switching elements S1 and S2
However, when the power supply voltage + VI and -VI are repeatedly turned on and off at the timings of (f) and (ki) in FIG. 6, the power supply voltages + VI and -VI become AC with a peak value VI at a point A, and the inductance L2,
Diode D1, diode D through capacitor C2
It is rectified by 2 and smoothed by the capacitors C3 and C4 to become a direct current of ZV1, and a current flows through the load RL.

When S1 is on, D1 is in the forward direction and the charge current flows into C3. However, assuming that the impedance of S1 and D1 is sufficiently small, C3 >> C2 is set. A sinusoidal series resonance current due to C2 is obtained (see (a) in FIG. 6). This resonance current becomes a reverse voltage and turns off when a half wave passes and the direction of the current is reversed. Therefore, series resonance cannot be performed and resonance stops. That is, the resonance automatically stops when the resonance current ends in half wave and the current returns to zero.

At this time, in C2, charges corresponding to the flowing resonance current are accumulated, and a voltage remains at both ends (see (e) in FIG. 6). This charge QC2 = C2 · VC2 is released to the load in the cycle in which the next S2 is turned off, so that it does not cause energy loss. Since the energy stored in the inductance is proportional to the current, the energy of L2 is zero when the resonance stops at zero current. This means that the generation of harmful noise here is extremely small, and is a major point in which the voltage resonance mode is established in the final circuit.

In order to completely reduce the magnetic energy of L2 to zero, it is necessary to turn on S1 until the resonance current returns to zero. After the resonance current becomes zero, nothing happens even if S1 is kept turned on, but it is inefficient because the time during which energy is not transmitted becomes long, so it may be turned off with some margin. Since the resonance time due to L2 and C2 is constant, the ON time of S1 may be a constant value.

When S1 is turned off, the current resonance ends and the current becomes zero. Therefore, the current flowing through S1 is only the current flowing through the inductance L1. The value of L1 is L2,
It can be set independently of C2, and by setting L1 >> L2,
Since the current flowing through L1 can be set to a value sufficiently smaller than the resonance currents of L2 and C2, S1 is almost zero current off, and the loss at the time of off is extremely small. When S1 is turned off (since S2 has not been turned on yet, both S1 and S2 are turned off). Since D1 and D2 are also turned off, the operation here is only L1 and the capacitor C1.

The magnetic energy (current) stored in L1 while S1 is on becomes energy for operating parallel resonance with C1, and the voltage at point A is reduced in a sine wave form.
It goes beyond zero and approaches -VI. The operation during this period is the voltage resonance mode. Note that, in principle, the voltage resonance waveform is vertically symmetrical with respect to a point intersecting with the voltage reference potential (potential expressed as zero in FIG. 6A), and is formed as shown in FIG. 6A. However, depending on the circuit configuration (specifically, it may be considered that the timing control circuit or the like partially consumes the energy of this voltage resonance through its coupling winding), the waveform deformation may occur. Can happen.

When the potential at the point A becomes close to -VI (lower than the potential at one end of C4), D2 is turned on and the remaining energy (current) of L1 is released to C4 through L2, C2, D2, but L1. Since the current of is originally set to be small, it does not change greatly in terms of current and the potential at point A is-
It will be stopped near VI. As it is, S1, S2
If you keep turning off, the magnetic energy (current) of L1 becomes zero in about half the time that S1 was on and L1
The voltage across (C1) drops from a potential near -VI toward zero. Conversely, for about half the on-time of S1, the magnetic energy of L1 causes point A to be -VI.
Since the potential can be maintained at a near potential, if S2 is turned on during that time, S2 becomes a zero voltage ON operation in which the voltage across the S2 is very small, and the loss at the time of ON is extremely small.

Voltage at both ends when S2 is turned on (-V above)
Strictly speaking, the difference between the value expressed as near I and −VI is not zero, and VC2 that remains after the current resonance when S1 is on is mainly
And so on. However, VC2 has different values depending on the value of C2. L2 at the same resonance frequency
In consideration of this, there is a degree of freedom in the setting of C2, and generally, in the range where series resonance normally occurs, the larger C2 is and the smaller L2 is, the smaller the loss is. Compared to, the voltage is almost negligible.

When S2 is turned on, negative side current resonance occurs and C
A charge current flows through 4. Thereafter, as shown in FIG. 6, the above-described operation is repeated while switching the positions of S1 and S2.

From the time when S1 is turned off to the time when S2 is turned on, point A is -VI due to voltage resonance caused by L1 and C1.
It only takes a little longer than the time it takes to get closer, and this too is just inefficient if taken too long. This time does not require a strict setting and may be a fixed value.

As a reminder, the ON period of S1 and S2 and the time from when S1 or S2 is turned off to when S2 or S1 is turned on will be examined a little now. It can be said that the ON period of the switch element is longer than the resonance half cycle of the series resonance means, and the OFF period of both switch elements is shorter than 1/2 of the resonance cycle of the parallel resonance means. Also, L
It should be noted that the amount of energy given to the voltage resonance circuit by C1 and C1 in advance is examined, and additionally, how to set each value of L1 and C1 even when the same parallel resonance frequency is set. That is, the ON period of each switch element determines the applied energy, and the OFF period is naturally restricted by the given energy (that is, the ON period equivalent value). According to the analysis, in practice, if the on period and the off period are determined, the switching frequency is determined at that time, and the parallel resonance (voltage resonance) frequency that satisfies the operation of the present invention and the portion of use of the parallel resonance waveform are determined. Has been determined to be unique. For example, when the ON period is set to a finite small value (nearly zero), the voltage resonance waveform in that case appears to change substantially on a sine wave at the same frequency as the switching frequency. It should be noted that in some cases, even when the voltage peak value of the voltage resonance is reached, the desired change in the output terminal potential of 2VI may not be realized yet.

Further, as is clear from the above description, L1 >> L2, C2 >> are set as the condition for setting the value of each resonance circuit.
> C1 is desirable, the rectification method needs to be a full-wave rectification method, the smoothing method is a capacitor input method for current resonance, and the smoothing capacitor has a capacitance considerably larger than that of the series resonance means. Make it big,
It is necessary to prevent the Q of current resonance from decreasing.

Next, the practical value of the above-described principle configuration will now be examined in a little more detail. First, the setting of the output voltage of the principle configuration shown in FIG. 5 will be described. At first glance, the voltage conversion means such as a transformer cannot be seen in this principle configuration, so it seems that the output voltage cannot be adjusted. However, if the output rectifier circuit configuration is doubled or doubled, the input voltage and the output voltage change in an integral multiple relationship. Further, if the inductance of the parallel resonance means is configured as a self-transformer, an arbitrary voltage output can be obtained. That is, if an intermediate tap is provided at an arbitrary winding position of a transformer or the like forming the inductance L1 and one end of the switching circuit output end and one end of the series resonance circuit on the switching circuit output end side are connected between required terminals, an arbitrary voltage output can be obtained. can get. As described above, the configuration shown in FIG. 4 or 5 has a corresponding degree of freedom regarding the output voltage. Next, the isolation function between input and output will be described. 1 function for this kind of power supply
Although there is insulation on the secondary side and the secondary side (isolate), if it is expressed as shown in FIG. 4 or FIG. 5, it seems that the function does not exist, but an insulating function is provided by using an intermediate transformer or the like. be able to. Next, output regulation with respect to load fluctuation will be described. 4 or 5
The configuration shown in (1) does not have a special constant voltage function. As one of the uses of this power supply circuit, a power supply circuit such as a power amplifier for audio is assumed. In such a use, a load abnormally smaller than the rated load is connected to the amplifier output. This is because it is usually better not to make the output stage a constant voltage in order to prevent the circuit from becoming overpowered and burned out.However, if this power supply circuit is used for another purpose or the power supply of the amplifier is fixed for some reason, When voltage conversion is required, an additional configuration for constant voltage conversion can be added, or at the subsequent stage of this power supply circuit,
For example, a series regulator is enough. Regarding the basic load power followability (the function in which the output voltage is almost constant and only the output current changes even if the load current changes), the configuration of FIG. I have it. That is, the voltage drop of the output extraction capacitor is determined according to the load current, and the voltage difference V of this voltage drop is applied to the impedance Z at both ends of the series resonance means (in principle, zero, but actually a finite small value). , A series resonance current that is almost determined by V / Z flows. That is, the resonance current is large when a large amount of power is consumed and small when a small amount of power is consumed.

When the above-described principle configuration is to be embodied as an actual circuit, as is clear from the above-described description of the principle configuration, the actual setting condition of the value of each resonance circuit is as follows.
Since L1 >> L2 and C2 >> C1 are desirable, L1 is the primary self-inductance of the transformer, L2
Can be effectively used by using independent inductance or by using leakage inductance between the primary and secondary sides of the transformer. Since the rectifier circuit is located on the secondary side of the transformer, either a center tap method or a bridge method may be used, but it is necessary to use a full-wave rectification method because it is necessary to perform current resonance with positive and negative currents. The smoothing method is
Capacitor input method for current resonance, C
3 >> C2 is set so that the Q of current resonance is not lowered.

When the transformer is viewed from the primary side, it looks like FIG. Since the transformer originally has self-inductance and leakage inductance, it can be used in place of L1 and L2 in FIG. 5 by setting these values to appropriate values at the time of design. Also, in general transformers originally L1
> L2.

FIG. 8 is a modification of the principle configuration circuit shown in FIG.
become that way. In FIG. 8, current resonance is performed by L2 and C2 which is divided into two, and voltage resonance is performed by C1 which is divided into two.
And L1. In the loop of voltage resonance, L2, C
It seems to be different from that of FIG. 5 in that it also includes 2, but since L2 << L1, C2 >> C1, L2, C
The presence of 2 does not affect the voltage resonance, and substantial voltage resonance is performed by C1 and L1 as in the configuration of FIG.

FIG. 9 is a more specific implementation circuit using a transformer T1 having a self-inductance L1 and a leakage inductance L2. The output circuit is of center tap type. The reason for adopting the center tap method is to reduce the number of diodes on the rectification path in each rectification cycle and minimize the loss due to these diodes to contribute to the improvement of the efficiency of the entire circuit. Then, the bases of S1 and S2 are driven at a fixed timing by the drive circuit having the timing as shown in FIGS. With such an extremely simple circuit, a low noise and highly efficient power supply circuit can be realized.

The effects of the configuration of the power supply circuit described above can be summarized as follows. The effect of current resonance is to reduce current noise. A large amount of current noise is generated when a sudden change in current is generated especially in a large amount of current, but noise is extremely generated because the current that has changed sinusoidally due to current resonance automatically stops when it reaches zero. Few. Next is efficiency improvement, but S1 and S
2 is turned off at zero current, and the voltages of D1 and D2 are also inverted after the current is zero. Therefore, the influence of recovery time is small, and the deterioration of efficiency due to this is eliminated. The effect of voltage resonance is to reduce noise and improve efficiency.
Components such as semiconductors used in the power supply circuit are attached to a chassis or the like via an insulator for heat dissipation, and thus the component electrodes and the chassis have an electric capacity. Therefore, current flows through this capacity where the component electrode has an AC signal,
It is the main cause of common mode noise. In addition, the semiconductor itself has a junction capacitance, and the inductance and the transformer also have a line capacitance. Although these capacitances are not shown in the circuit diagram, they actually exist in each component or circuit board, so that current flows through all these capacitances when the circuit is operating. Since this current is a current flowing through the capacitor, the larger the voltage change (dV / dT), the larger the current becomes, and when switching with a square wave, it becomes a pulsed current, causing current noise, or the current flowing in the chassis. Causes pulsed common mode noise. Further, since this pulsed current is supplied from the switching transistor, it naturally causes a loss and reduces efficiency. Since a large voltage of dV / dT contains a high frequency component, a radio wave (unnecessary secondary radiation) directly emitted from the circuit naturally becomes large.

These improvements can be realized by making the waveform part of a sine wave by utilizing voltage resonance and reducing dV / dT. In the present invention, however, this voltage resonance is S1, S2.
Since both L1 and C1 are made when both are on, no loss of switching element occurs, and the current flowing through L1 and C1 is only energy transfer between them, only reactive power, and voltage resonance The loss due to is extremely small (in principle, it is zero).

What is important here is that in order to reduce voltage noise, dV / of voltage waveforms of all terminals in the circuit is reduced.
It is necessary that dT is small. If there is a square waveform even at one place, it will become a noise source. In a general voltage resonance type power supply circuit, although a certain point in the circuit (for example, a transformer output) has a sine wave shape, a square waveform exists (in another circuit portion) in many cases. This power supply circuit has the most important goal of practical noise reduction, and is characterized in that all voltage waveforms are similar to the voltage resonance waveforms of L1 and C1. The reason for satisfying this point is that the voltage resonance is used separately from the current resonance. By current resonance, the currents of S1, S2 and D2 are set to zero,
After the magnetic energy of L2 is also zeroed, it is brought into the voltage resonance mode and S1, S2, D1, and D2 are kept in the OFF state, so that the current movement of L2 and C2 in the voltage resonance mode is zeroed. The waveforms at points A and A'are the same.
As a result, the terminal voltage waveforms of L1 and C1 and S1, S2,
The waveforms of all terminals of L2, C2, D1, and D2 become the same (similar), and the square waveform disappears from the circuit.

FIG. 10 shows an embodiment in which the present invention is applied to the power supply circuit described above. Voltage of DC power supply 1 + B,-
B is capacitors C2 and C2 via the power switch SW1.
Is being applied to. In addition to the configuration shown in FIG. 1, the circuits U1 and U2 have a time constant for delaying the on-timing of the main switching transistors 25 and 26 to form the off period of both transistors as shown in FIGS. A circuit is provided. That is, capacitors 35C and 36C connected between the bases and collectors of the main switching transistors 25 and 26, which cooperate with resistors 35R and 36R corresponding to the resistors 35 and 36 in the configuration of FIG. It acts to delay the on-timing by a predetermined time.
The operation of the starting circuit is similar to that of FIG.
It self-starts at 5'and self-oscillates. The voltage resonance is performed by the C1 divided into two and the self-inductance L2 of the winding 20. The current resonance is performed by the two-divided C2 and the leakage inductance L1 of the winding 20. The secondary side output of the transformer T1 is rectified and smoothed by the diodes D1 and D2 and the capacitor C ₀ and supplied to the load RL. Also in this circuit, as in the case of FIG. 1, the voltage dividing resistors 50, 50 'or 5 are used.
1, 51 'prevent re-oscillation when the power switch SW1 is turned off.

A specific circuit configuration of the power supply circuit of FIG. 10 is shown in FIG.
Shown in 1. In FIG. 11, the constants attached to each component are examples of specific design constants. Further, the reference numerals of the respective constituent parts are the same as those of the corresponding constituent elements in the above description. In this specific circuit configuration, a 100V system / 200V system can be switched and used as a commercial AC power source. When a 100 V AC power source is used, a voltage doubler rectification configuration is used. Since the intermediate connection point of the input smoothing capacitor can be used as a reference potential, the series resonance capacitor (C2) is not a two-divided structure to the power supply line, but is a single C coil in the transformer winding.
Connected as 2 '. According to such a capacitor insertion method, the withstand voltage of the capacitor itself can be small, and a film capacitor with a small loss can be easily used.
This power supply circuit is designed with a rated output of 500 W (maximum 1 kW), a switching frequency of 35 kHz, a current resonance frequency of 50 kHz, and a voltage resonance frequency of 60 kHz.
Therefore, the actual measured value of the self-inductance of the transformer is 2.
The measured leakage inductance was 3 mH and 2.3 μH. In this concrete circuit configuration, the actual measured value of the efficiency, which is represented by the ratio of the output power extracted from the smoothing capacitor to the input power of the commercial AC power source, is about 97% (normal transformer power supply is about 80%, the conventional switching power supply is about 80%). It was about 80-85% with the power supply. In addition, this power supply circuit can reduce the noise generation amount by nearly 30 dB as compared with the conventional switching power supply. Specifically, this power supply circuit is operated without any shield or the like, and the AM radio reception is performed nearby. Is practically no problem at all.

[0059]

As described above, according to the present invention, since the voltage of the power source capacitor is divided by the voltage dividing means and applied to the second capacitor of the starting circuit, the power source switch is turned off. If the voltage of the power supply capacitor drops and the oscillation stops, the voltage that turns on the main switching transistor cannot be reached even when the second capacitor is charged, so it is possible to reliably prevent the oscillation from restarting. Alternatively, the number of restarts can be reduced even if the restart of the oscillation cannot be completely prevented, and the oscillation can be stopped in a short time.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a circuit diagram showing a conventional self-excited switching inverter type power supply circuit.

FIG. 3 is a circuit diagram in which a starting circuit is added to the circuit of FIG.

FIG. 4 is a schematic block diagram showing the basic principle of a power supply circuit described in Japanese Patent Application No. 3-166383.

5 is a configuration circuit diagram showing a basic principle configuration of the power supply circuit of FIG.

FIG. 6 is a timing diagram illustrating the operation of the principle configuration shown in FIG.

FIG. 7 is an explanatory diagram illustrating an equivalent circuit of a transformer.

FIG. 8 is a circuit diagram illustrating a modification of the principle configuration shown in FIG.

9 is a circuit diagram showing a modified configuration of the principle configuration shown in FIG.

FIG. 10 is a circuit diagram showing an embodiment in which the present invention is applied to the power supply circuit described in FIGS.

11 is a specific circuit configuration diagram of the circuit of FIG.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 DC power supply 15,15 'Starting circuit (second time constant circuit) 20 Load windings 21,22 Feedback winding 25 First main switching transistor 26 Second main switching transistor 27 First time constant circuit 28th 1 time constant circuit 27R, 28R First resistor 27C, 28C First capacitor 29 First auxiliary switching transistor 30 Second auxiliary switching transistor 40, 40 'Second resistor 41, 41' Second capacitor 50 , 50 ', 51, 51' Voltage dividing resistance (voltage dividing means)

Claims (1)

[Claims] 1. A DC power supply, a power switch for turning the DC power supply on and off, a power supply capacitor charged by the DC power supply when the power supply switch is on, and a main current path for the power supply capacitor. First connected in series
A second main switching transistor, a load winding connected between a connection point between the first and second main switching transistors and an intermediate potential of the power supply capacitor, the first and second main switching transistors, It is connected between the base and the emitter of the switching transistor, and when a current flows from the first or second main switching transistor to the load winding, the mutual induction electromotive force generated by the load winding is changed to the first or the second. A feedback winding generated so as to be positively fed back to the second main switching transistor, and a first resistor and a first resistor connected in series at both ends of the feedback winding and driven by an induced electromotive force of the feedback winding. A first time constant circuit composed of one capacitor and a main current path connected between the base and emitter of the first and second main switching transistors, respectively. When the output of the first time constant circuit reaches a predetermined level while the first or second main switching transistor is on, the output is turned on to change the base current of the first or second main switching transistor. Short circuit
A first and second auxiliary switching transistor for turning off the first or second main switching transistor, and a series connection circuit of a second resistor and a second capacitor between the power supply capacitor and the feedback winding. To form a current path from the power supply capacitor to the load winding via the feedback winding when the main switching transistor is off, and the voltage of the second capacitor is used as the base of the main switching transistor. And a voltage dividing means for dividing the voltage of the power source capacitor between the second resistor and the second resistor to apply the voltage to the second capacitor. Power supply circuit.