JP3030974B2 - Power supply circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply circuit of a self-excited switching inverter type, and in a case where a start-up circuit is added, prevents a phenomenon in which start-up is repeated after turning off a power switch and stops oscillation in a short time. It is made possible.

[0002]

2. Description of the Related Art As a power supply circuit of a self-excited inverter type, a circuit disclosed in Japanese Patent Publication No. Hei 3-1914 filed by the present applicant has been proposed. An example of the circuit configuration will be described with reference to FIG. This circuit includes a circuit indicated by U1 and a circuit indicated by U1.
2 alternately turns on and off so that an AC voltage is generated in the winding 20. The windings 21 (feedback winding), 22 (feedback winding) and 20 (load winding) are wound on the same core of the transformer, and windings 20 and 21 and windings 20 and 22 are respectively positive feedback. It is connected in the direction as follows. The voltages + B and -B of the DC power supply 1 are connected to the power switch SW1.
To the power supply capacitors C2 and C2.

In the circuit U1, a transistor 25 is a main switching transistor, a transistor 29 is an auxiliary switching transistor, and a capacitor 27C is a capacitor which forms a time constant circuit 27 together with a resistor 27R. The diodes 11 and 12 operate the transistor 25 in an unsaturated manner. That is, the winding 21
Is applied to the base of the transistor 25 via the diode 11 to turn it on.
Assuming that the drop voltages of the diodes 11 and 12 are equal, when the collector-emitter voltage V _{CE of} the transistor 25 becomes equal to the base-emitter voltage V _BE , the diode 12
Turns on, and an extra current flows between the collector and the emitter of the transistor 25 except for a current necessary to maintain V _CE = V _BE . As a result, V _CE is always kept at V _CE ≧ V _BE . Normally, the saturation voltage V _CE of the transistor is smaller than the base-emitter voltage V _BE , so that the transistor 25 is prevented from being saturated. The circuit U2 is configured similarly to the circuit U1.

The circuit shown in FIG. 2 operates as follows. Circuit U
When 1 is in the ON state, the transistor 25 keeps on due to the positive feedback action of the windings 21 and 20. Capacitor 2
The voltage of 7C rises with time at a time constant of 27R × 27C, and after a predetermined time, the transistor 29 is turned on and the transistor 25 is turned off. When the transistor 25 is turned off, the voltage applied to both ends of the winding 20 is inverted by the self-induction of the winding 20, and the circuit U2 is turned on.

A certain time is required from the time when the transistor 25 is turned off to the time when the transistor 26 is turned on. During this time, the transistor 25 must be kept off. Fortunately, this is secured for two reasons: One is that the transistor 29 is in the ON state for a while due to the overcharge stored in the capacitor 27C during the time from when the transistor 29 is turned on to when the transistor 25 is turned on (the off time of the transistor 25). One is due to the off-time of the transistor 29 itself (the transistor 29 has completely saturated switching, and the turn-off takes a certain time). When the transistor 26 is turned on within a time period in which the off state of the transistor 25 is held, stable operation can be performed. When the transistor 26 is turned on, the transistor 25 becomes 0 bias and keeps off.

[0006] After a certain time by a time constant 28R × 28C,
From now on, the transistor 30 will be on and the transistor 26 will be off. The oscillation mode is formed in this way, and operates as a switching inverter.

The circuit shown in FIG. 2 cannot start itself in this state. FIG. 3 shows an example in which a starting circuit is added. This is shown in FIG. 1 of Japanese Patent Application No. 3-166383 filed by the present applicant.
0 has been proposed.

In FIG. 3, a starting circuit 15 including a resistor 40, a capacitor 41, and a diode 42 is added to the base circuit of the main switching transistor 25. Since the resistor 40 involves a power loss, the resistor 40 has a high resistance, and the capacity of the capacitor 41 is increased for a strong start.

When the power switch SW1 is turned on, the winding voltage of the transformer is zero, and the charge of the capacitor 41 is also zero. The current flowing through the resistor 40 is small and passes through the resistor 35 while charging the capacitor 41.
To the point B, but the current is not large enough to cause a large potential difference in the resistor 35. Therefore, first, point B, C
The points D and D have substantially the same potential, so the main switching transistor 25 is also off. The capacitor 41 is charged with time, and when the potential difference between the points B and D reaches the forward voltage between the base and the emitter of the main switching transistor 25, the transistor 25 enters the active region. At this time, a positive return loop is formed from the return winding 21 to the base of the transistor 25 through the resistor 35 and the capacitor 41, and the main switching transistor 25 is rapidly turned on by positive feedback. It continues to turn on due to the large base current flowing through it. At this time, since the current flowing from the resistor 35 to the capacitor 41 is much larger than the current reversely charged from the resistor 40, the potential of the capacitor 41 becomes the forward potential of the diode 42. Fixed. After starting, the diode 42 forms and maintains a positive feedback loop.
Therefore, after startup, oscillation is maintained and the charging current from the winding 21 is always dominant, and the slight reverse charging current from the resistor 40 is no longer equal to nothing, and the main switching transistor 25 is turned off by the resistor 40. There is no accidental turning on at the right time.

[0010]

The starting circuit 1 shown in FIG.
According to 5 (15 '), when the power switch SW1 is turned off, the following operation is performed. Even if the power switch SW1 is turned off, since the electric charge is stored in the capacitor C2, the oscillation continues as it is.

Since the supply of electric charge from DC power supply 1 to capacitor C 2 is stopped, the voltage of capacitor C 2 (potential at point A) gradually decreases, and the voltage induced in winding 21 also decreases.

When the voltage applied between the base and the emitter of the main switching transistor 25 by the voltage induced in the winding 21 falls below its forward voltage (about 0.6 V), the main switching transistor 25 can no longer be turned on. Therefore, oscillation stops.

[0013] Even if the oscillation stops, the electric charge remains in the capacitor C2, so that the discharge current of the capacitor C2 is changed from the resistance 40 → the capacitor 41 → the resistance 35 → the winding 21 → the winding 2
It flows to 0, and the capacitor 41 is gradually reverse-charged. Since the oscillation is stopped, the reverse charge canceling current from the winding 21 cannot occur.

When the capacitor 41 is reversely charged with time, and the potential difference between the points B and D reaches the forward voltage between the base and the emitter of the main switching transistor 25, the transistor 25 enters the active region again. At this time, although the voltage induced in the return ring winding 21 itself is low, it becomes in series with the voltage of the capacitor 41, so that a positive return loop is formed, and the main switching transistor 25 is turned on at an accelerated speed to resume oscillation.

When the oscillation restarts, the capacitor 41 is charged in the forward direction with respect to the diode 42 by the current from the winding 21. Since the voltage induced in the winding 21 is low, the base-emitter voltage of the main switching transistor 25 drops below its forward voltage, and this transistor 25 can no longer be turned on, and stops oscillating.

However, when the oscillation stops, the capacitor 40 is again charged from the capacitor C2 through the route of the resistor 40 → the capacitor 40 →.

As described above, the starting circuit 15 (1) shown in FIG.
According to 5 '), even after the power switch SW1 is turned off, A
Oscillation and stop are repeated until the potential of the point becomes sufficiently small.
There was a disadvantage that oscillation could not be stopped immediately. When viewed as a device, the power lamp or the like flashes even after the power is turned off, and furthermore, the frequency of oscillation decreases, so that a transformer or the like finally emits an unusual sound. The present invention
An object of the present invention is to provide an amplifier circuit which can stop oscillation in a short time after turning off a power switch, by solving the drawbacks of the conventional technique.

[0018]

According to the present invention, a series connection circuit of a second resistor and a second capacitor is interposed between a power supply capacitor and a feedback winding. A self-excited inverter type including a start circuit including a second time constant circuit for forming a current path from a capacitor to a load winding via a feedback winding and applying a voltage of the second capacitor to a base of a switching transistor. And a voltage dividing means for dividing the voltage of the power supply capacitor between the power supply capacitor and the second resistor and applying the voltage to the second capacitor.

[0019]

According to the present invention, since the voltage of the power supply capacitor is divided by the voltage dividing means and applied to the second capacitor of the starting circuit, the power supply switch is turned off to lower the voltage of the power supply capacitor. When the oscillation stops, the voltage that turns on the main switching transistor can be prevented from being reached even when the second capacitor is charged, so that the oscillation can be reliably prevented from restarting, or the oscillation can be completely restarted. Even if it cannot be prevented, the number of restarts can be reduced, and oscillation can be stopped in a short time.

[0020]

FIG. 1 shows an embodiment of the present invention. FIG. 3
The same reference numerals are given to portions common to. The power supply in FIG.
The circuit indicated by 1 and the circuit indicated by U2 are alternately turned on and off, so that an AC voltage is generated in the winding 20. Windings 21 (feedback winding), 22 (feedback winding), 20
The (load winding) is wound on the same core of the transformer, and the windings 20 and 21 and the windings 20 and 22 are respectively connected in a positive feedback direction. The voltage of DC power supply 1 + B,
-B is a power supply capacitor C via a power switch SW1.
2, C2.

In the circuit U1, the transistor 25 is a main switching transistor, the transistor 29 is an auxiliary switching transistor, and the capacitor 27C is a capacitor constituting the time constant circuit 27 together with the resistor 27R. A starting circuit 15 including resistors 40 and 50, a capacitor 41, and a diode 42 is added to the base circuit of the main switching transistor 25. Since the resistor 40 involves a power loss, the resistor 40 has a high resistance, and the capacity of the capacitor 41 is increased for a strong start. The resistor 50 is a resistor for voltage division. The circuit U2 is configured similarly to the circuit U1.

The power supply circuit of FIG. 1 operates as follows. When the power switch SW1 is turned on, the winding voltage of the transformer is zero, and the charging voltage of the capacitor 41 is also zero. The current flowing through the resistor 40 is small, passes through the resistor 35 while charging the capacitor 41, flows through the winding 21, and reaches point B, but is not enough current to cause a large potential difference in the resistor 35. Therefore, initially, the points B, C, and D have substantially the same potential, so that the main switching transistor 25 is also off.

With time, the capacitor 41 is charged,
The potential difference between the points B and D is the main switching transistor 25
Transistor 25 enters the active region. At this time, a positive return loop is formed from the return winding 21 to the base of the transistor 25 through the resistor 35 and the capacitor 41, and the main switching transistor 25 is turned on at an accelerated rate by positive feedback.
It remains on due to the large base current flowing from 1 through resistor 35. At this time, since the current flowing from the resistor 35 to the capacitor 41 is much larger than the current reversely charged from the resistor 40, the potential of the capacitor 41 is
2 and is fixed at the forward potential of the diode 42 after startup. Therefore, the main switching transistor 25 is not accidentally turned on by the resistors 40 and 50 at the off timing of the transistor 25 after startup.

When the circuit U1 is on, the windings 21 and
Due to the positive feedback action of 20, the transistor 25 keeps on. The voltage of the capacitor 27C increases with time with a time constant of 27R × 27C, and after a predetermined time, the transistor 2
9 is turned on and the transistor 25 is turned off. When the transistor 25 is turned off, the winding 20
Is inverted, and the circuit U2 side is turned on this time.

A certain period of time is required from the time when the transistor 25 is turned off to the time when the transistor 26 is turned on. During this time, the off state of the transistor 25 is maintained due to the overcharge stored in the capacitor 27C and the off time of the transistor 29 itself. I have. When the transistor 26 is turned on within a time period in which the off state of the transistor 25 is held, stable operation can be performed. When the transistor 26 is turned on, the transistor 25 becomes 0 bias and keeps off.

After a certain time by a time constant 28R × 28C,
From now on, the transistor 30 will be on and the transistor 26 will be off. The oscillation mode is formed in this way, and operates as a switching inverter.

When the power switch SW1 is turned off, the following operation is performed. Even if the power switch SW1 is turned off, since the electric charge is stored in the capacitor C2, the oscillation continues as it is.

Since the supply of charge from the DC power supply 1 to the capacitor C 2 is stopped, the voltage of the capacitor C 2 (potential at point A) gradually decreases, and the voltage induced in the winding 21 also decreases.

When the voltage applied between the base and the emitter of the main switching transistor 25 becomes lower than its forward voltage (about 0.6 V) due to the voltage induced in the winding 21, the main switching transistor 25 can no longer be turned on. Therefore, oscillation stops.

Even if the oscillation is stopped, since the electric charge remains in the capacitor C2, the discharge current of the capacitor C2 is changed from the resistance 40 → the capacitor 41 → the resistance 35 → the winding 21 → the winding 2
It flows to 0, and the capacitor 41 is gradually reverse-charged. Since the oscillation is stopped, the reverse charge canceling current from the winding 21 cannot occur.

However, since the capacitor 40 is charged only by the current path up to the voltage divided by the resistor 50, the divided voltage (more precisely, the voltage divided by the resistor 50 + the resistor 35) is used for the main switching. If the value is lower than the forward voltage between the base and the emitter of the transistor 25 (about 0.6 V), the main switching transistor 25
Cannot be turned on and oscillation is not resumed. Therefore, the electric charge of the capacitor C2 is directly transferred to the resistors 40, 50, 35,
It is gradually discharged through the windings 21 and 20.

As is clear from the above, the value of the voltage dividing resistor 50 is adjusted so that the following (a) and (b) are satisfied.
If the setting is made in relation to 5 or the like, the restart of oscillation can be prevented when the power switch 2 is turned off. (A) When the power switch SW1 is turned on from the off state, the capacitor 41 can be charged to a voltage at which the main switching transistor 25 can be turned on with respect to the voltages + B and -B of the capacitor C2. (B) When the power switch SW1 is turned off from the on state, the capacitor 41 is not charged to a voltage at which the main switching transistor can be turned on with respect to the reduced voltage of the capacitor C2 when the oscillation is stopped for the first time.

(B) is a condition for preventing oscillation from being restarted at all. When the voltage divided by the resistor 50 is set to a higher voltage, the oscillation is restarted several times, but the effect of reducing the number of restarts is obtained, and the voltage dividing resistor in FIG. Oscillation can be stopped in a short time as compared with the case where there is no oscillation. Note that the voltage dividing resistor may be such that the resistor 51 is inserted at a position indicated by a dotted line instead of the resistor 50.

Next, the present invention is disclosed in Japanese Patent Application No. 3-166383.
An embodiment applied to the power supply circuit of FIG. First, this power supply circuit will be described. This power supply circuit
By using both voltage resonance and current resonance to reduce switching loss to the utmost and improve conversion efficiency, the operating waveform of each part voltage and each part current in the circuit is made closer to a sine wave to reduce noise. Things.

As shown in FIG. 4, the power supply circuit includes a DC power supply 1 and a switching element which can be turned on / off at an arbitrary timing, and a switching means 2 for switching the input DC power supply to convert the input DC power supply into AC and output it. When,
DC output means 3 for full-wave rectifying the supplied AC input and smoothing the output with a capacitor to obtain a DC output; series resonance means 4 formed in series with a current flowing through an output terminal of the switching means; Parallel resonance means 5 formed in parallel with the voltage developed at the output terminal of the means
And timing control means 6 for controlling the switching element of the switching means to be turned on intermittently.

FIG. 5 is a block diagram showing the basic principle of the block shown in FIG. The operation of the basic principle configuration shown in FIG. 5 will be described with reference to FIG. 6 showing the operation timing of each unit.

In FIG. 5, switching elements S1, S2
However, when on and off are repeated at the timings shown in FIGS. 6F and 6G, the power supply voltages + VI and -VI become substantially alternating currents of the peak value VI at the point A, and the inductance L2,
Diode D1, diode D through capacitor C2
2, is smoothed by the capacitors C3 and C4, becomes DC of ZV1, and a current flows to the load RL.

When S1 is on, D1 becomes forward and the charge current flows into C3. However, since the impedances of S1 and D1 are set to be sufficiently small, C3 >> C2. A sinusoidal series resonance current is generated by C2 (see (a) in FIG. 6). This resonance current is turned off when the direction of the current is reversed after a lapse of a half-wave, because D1 becomes a reverse voltage, so that the series resonance cannot be performed and the resonance stops. In other words, the resonance automatically stops when the half-wave of the resonance current ends and the current returns to zero.

At this time, at C2, a charge corresponding to the flowing resonance current is accumulated, and a voltage remains at both ends (see (e) in FIG. 6). Since this charge QC2 = C2.VC2 is released to the load in the next cycle in which S2 is turned off, no energy is lost. Since the energy stored in the inductance is proportional to the current, the energy of L2 is zero when the resonance stops at zero current. This means that the generation of harmful noise here is extremely small, and is a major point where the voltage resonance mode is established in the final circuit.

In order to completely reduce the magnetic energy of L2 to zero, it is necessary to keep S1 on until the resonance current returns to zero. After the resonance current becomes zero, nothing happens even if S1 is kept on, but it is inefficient because only the time during which energy is not transmitted is lengthened. Therefore, it is sufficient to turn off S1 with some margin. Since the time of resonance by L2 and C2 is constant, the ON time of S1 may be a constant value.

When S1 is turned off, the current resonance ends and the current becomes zero, so that the current flowing through S1 is only the current flowing through the inductance L1. The value of L1 is L2,
It can be set independently of C2, and by setting L1 >> L2,
Since the current flowing through L1 can be a sufficiently small value compared to the resonance currents of L2 and C2, S1 is almost zero current off, and the loss during off is extremely small. When S1 is turned off (S2 is not turned on yet, both S1 and S2 are turned off), D1 and D2 are also turned off, so that the operation here is only L1 and capacitor C1.

The magnetic energy (current) stored in L1 while S1 is on becomes the energy for operating parallel resonance with C1, and lowers the voltage at point A in a sinusoidal manner.
Beyond zero, approaching -VI. The operation during this is the voltage resonance mode. The voltage resonance waveform is in principle symmetrical with respect to a point intersecting with a voltage reference potential (potential denoted as zero in FIG. 6A) and formed as shown in FIG. 6A. However, depending on the circuit configuration (specifically, it is conceivable that the timing control circuit or the like partially consumes the energy of the voltage resonance via the coupling winding), the waveform deformation may occur. Can occur.

When the potential at point A becomes close to -VI (below the potential at one end of C4), D2 turns on and the remaining energy (current) of L1 is released to C4 through L2, C2 and D2, but L1 is released. Is originally set to be small, the current does not significantly change, and the potential at the point A becomes −
It is in a state of stopping near VI. S1, S2 as is
Is kept off, the magnetic energy (current) of L1 becomes zero in about half the time when S1 is on, and L1
The voltage across (C1) falls from a potential near -VI toward zero. Conversely, about half of the on-time of S1, the point A is -VI with the magnetic energy of L1.
Since S2 can be turned on during that time, if S2 is turned on during that time, S2 is turned on with zero voltage, and the loss at the time of turning on becomes extremely small.

The voltage at both ends when S2 is turned on (-V
Strictly speaking, the difference between the value expressed as close to I and -VI) is not zero, and VC2 mainly remaining after the current resonance when the S1 is turned on.
And so on. However, VC2 has a different value depending on the value of C2. L2 even at the same resonance frequency
In consideration of the above, there is a degree of freedom in setting C2. Generally, when C2 is increased and L2 is decreased in a range where the series resonance normally occurs, the loss is reduced. The voltage is almost negligible compared to.

When S2 is turned on, negative side current resonance occurs and C
4, a charging current flows. Thereafter, as shown in FIG. 6, the above operation is repeated while exchanging the positions of S1 and S2.

The time from turning off S1 to turning on S2 is such that the point A is -VI due to the voltage resonance caused by L1 and C1.
It only needs to be a little longer than the time to get close, and taking too long is just inefficient. This time does not need to be set so strictly, and may be a fixed value.

It should be noted that, just in case, the ON period of S1 and S2 and the time from when S1 or S2 is turned off to when S2 or S1 is turned on are slightly examined. It can be said that the on-period of the switch element is longer than the resonance half cycle of the series resonance means, and the off-period of both switch elements is shorter than 共振 of the resonance cycle of the parallel resonance means. Also, L
Attention should be paid to the consideration of the amount of energy given to the voltage resonance circuit by C1 and C1, and to the setting of each value of L1 and C1 even when the same parallel resonance frequency is used. That is, the ON period of each switch element determines its applied energy, and the OFF period is naturally restricted by the applied energy (ie, the ON period equivalent value). According to the analysis, when the ON period and the OFF period are actually determined, the switching frequency is determined at that time, and the parallel resonance (voltage resonance) frequency and the use of the parallel resonance waveform satisfying the operation of the present invention. Has been found to be uniquely determined. For example, when the ON period is set to a finite small value (substantially zero), the voltage resonance waveform in that case appears to change substantially on a sine wave at approximately the same frequency as the switching frequency. It should be noted that depending on the case, even if the voltage peak value of the voltage resonance is reached, the desired output terminal potential change of 2VI may not yet be realized.

Further, as is clear from the above description, L1 >> L2, C2 >>
> C1, it is necessary that the rectification method be a full-wave rectification method, and the smoothing method is a capacitor input method for current resonance, and the capacity of the smoothing capacitor is considerably larger than that of the series resonance means capacitor. Bigger,
It is necessary to prevent the Q of the current resonance from lowering.

Next, the practical value of the above-mentioned principle configuration will be examined in more detail. First, the setting of the output voltage of the principle configuration shown in FIG. 5 will be described. In this principle configuration, it seems that the output voltage cannot be adjusted because the voltage conversion means such as a transformer cannot be seen at first glance. However, if the output rectifier circuit configuration is a voltage doubler configuration that is twice or more, the input voltage and the output voltage change in an integer multiple relationship. If the inductance of the parallel resonance means is configured as a self-transformer, an arbitrary voltage output can be obtained. That is, if an intermediate tap is provided at an arbitrary winding position of a transformer or the like constituting the inductance L1, and an output terminal of the switching circuit and one end of the series resonance circuit on the side of the switching circuit output terminal are connected between necessary terminals, an arbitrary voltage output can be obtained. can get. Thus, the configuration shown in FIG. 4 or FIG. 5 has a corresponding degree of freedom for the output voltage. Next, the isolation function between input and output will be described. One function of this kind of power supply is
Although there is insulation (isolate) on the secondary side and secondary side, if it is expressed as shown in FIG. 4 or FIG. 5, it seems that there is no such function, but it is provided with an insulation function by using an intermediate transformer or the like. be able to. Next, output regulation with respect to load fluctuation will be described. FIG. 4 or FIG.
Does not have a special constant voltage function. This is because a power supply circuit such as an audio power amplifier is assumed as one of uses of this power supply circuit. In such a case, a load abnormally smaller than the rated load is connected to the amplifier output. In order to prevent the circuit from being overpowered and burned out, it is generally better not to set the output stage to a constant voltage. When voltage conversion is required, add an additional configuration for constant voltage conversion, or add
For example, it is only necessary to attach a series regulator. Regarding the basic load power followability (a function in which the output voltage is almost constant and the output current changes even if the load current fluctuates), the configuration of FIG. 4 or FIG. Have. That is, the voltage drop of the output take-out capacitor is determined in accordance with the load current, and the voltage difference V of this voltage drop is applied to the impedance Z (both is zero in principle, but actually a finite small value) across the series resonance means. , A series resonance current determined substantially by V / Z flows. That is, the resonance current is large when the power consumption is large, and small when the power consumption is small.

When the above-described principle configuration is embodied as an actual circuit, as is clear from the description of the above-described principle configuration, the actual setting conditions of the values of the respective resonance circuits are as follows.
Since it is desirable that L1 >> L2 and C2 >> C1, L1 is the primary self-inductance of the transformer, L2
In this case, a method using an independent inductance or a method utilizing a leakage inductance between the primary and secondary of the transformer can be effectively used. The rectifier circuit may be of a center tap type or a bridge type because it comes to the secondary side of the transformer. However, since it is necessary to perform current resonance with positive and negative currents, it is necessary to use a full-wave rectification type. The smoothing method is
Capacitor input method for current resonance, C
By setting 3 >> C2, the Q of the current resonance is prevented from lowering.

When the transformer is viewed from the primary side, it looks as shown in FIG. Transformers originally have self-inductance and leakage inductance, and can be used in place of L1 and L2 in FIG. 5 by setting them to appropriate values at the time of design. Also, with a general transformer, L1
> L2.

When the principle configuration circuit shown in FIG. 5 is modified, FIG.
become that way. In FIG. 8, current resonance is performed in L2 and C2 divided into two parts, and voltage resonance is performed in C1 divided into two parts.
And L1. L2, C in the voltage resonance loop
5 is different from that of FIG. 5 in that L2 << L1, C2 >> C1.
The presence of 2 does not affect the voltage resonance, and substantial voltage resonance occurs at C1 and L1 as in the configuration of FIG.

FIG. 9 shows a more specific implementation circuit using a transformer T1 having a self inductance L1 and a leakage inductance L2. The output circuit is of a center tap type. The reason for employing the center tap method is to reduce the number of diodes on the rectification path in each rectification cycle, minimize the loss due to these diodes, and contribute to improving the efficiency of the entire circuit. The bases of S1 and S2 are driven at a fixed timing by a drive circuit having timings as shown in FIGS. As described above, a very simple power supply circuit with low noise and high efficiency can be realized.

The effects of the configuration of the power supply circuit described above are summarized as follows. First, as an effect of current resonance, there is a reduction in current noise. A large amount of current noise is generated especially when a sudden change of current is caused in a large amount of current.However, the noise automatically stops when a sinusoidally changed current becomes zero due to current resonance. Few. Next is the improvement of efficiency.
2 turns off the zero current, and the voltages of D1 and D2 also become inverted after the currents become zero. Therefore, the influence of the recovery time is small, and the deterioration of efficiency due to this is eliminated. The effects of voltage resonance also reduce noise and improve efficiency.
Components such as semiconductors used for the power supply circuit are attached to a chassis or the like via an insulator for heat dissipation, and thus the component electrodes and the chassis have electric capacity. Therefore, a current flows through this capacitance where the component electrodes have an AC signal,
It is the main cause of common mode noise. Further, the semiconductor itself has a junction capacitance, and the inductance and the transformer also have a line capacitance. Although these capacitances do not appear on the circuit diagram, they actually exist in the respective components and circuit boards, so that when the circuit is operating, all the currents flow through these capacitances. Since this current is a current flowing through the capacitor, the larger the change in the voltage is (dV / dT), the larger the current becomes. If the switching is performed by a square wave, the current becomes a pulse-like current, resulting in a current noise or a current flowing through the chassis. Causes pulsed common mode noise. Also, since this pulsed current is supplied from the switching transistor, it naturally causes a loss and lowers the efficiency. Since a large voltage of dV / dT contains high frequency components, radio waves (unnecessary secondary radiation) directly radiated from the circuit naturally increase.

These improvements can be realized by making the waveform a part of a sine wave using voltage resonance and reducing dV / dT, but in the present invention, this voltage resonance is caused by S1 and S2.
When both are turned on, they are made only of L1 and C1. Therefore, no loss of the switching element occurs, and the current flowing through L1 and C1 is only mutual energy transfer, only reactive power, and voltage resonance. Is very small (in principle zero).

What is important here is that in order to reduce voltage noise, dV / dV of the voltage waveforms at all terminals in the circuit is reduced.
It is necessary that dT is small. If there is a square waveform even at one place, it becomes a noise source. In a general voltage resonance type power supply circuit, although a certain point in the circuit (for example, a transformer output) has a sine wave shape, a square waveform (in another circuit portion) exists in many cases. This power supply circuit has a primary goal of practical noise reduction, and is characterized in that all voltage waveforms are similar to the voltage resonance waveforms of L1 and C1. The reason for satisfying this point is that voltage resonance is used separately from current resonance. The current of S1, S2 and D2 is reduced to zero by current resonance,
After the magnetic energy of L2 is also reduced to zero, the voltage is transferred to the voltage resonance mode, and S1, S2, D1, and D2 are turned off so that the current movement of L2 and C2 during the voltage resonance mode is reduced to zero. The waveforms at points A and A 'are the same.
Thereby, the terminal voltage waveforms of L1, C1 and S1, S2,
The waveforms of all terminals of L2, C2, D1, and D2 become the same (similar), and the square waveform disappears from the circuit.

FIG. 10 shows an embodiment in which the present invention is applied to the power supply circuit described above. Voltage of DC power supply 1 + B,-
B is connected to the capacitors C2 and C2 via the power switch SW1.
Has been applied. The circuits U1 and U2 have, in addition to the configuration of FIG. 1, a time constant for delaying the ON timing of the main switching transistors 25 and 26 to form an OFF period of both transistors as shown in FIGS. A circuit is provided. That is, capacitors 35C and 36C connected between the bases and collectors of the main switching transistors 25 and 26, which cooperate with resistors 35R and 36R corresponding to the resistors 35 and 36 in the configuration of FIG. It acts to delay the ON timing for a predetermined time.
The operation of the starting circuit is the same as that of FIG.
Self-excited oscillation occurs at 5 '. Voltage resonance is performed by the divided C1 and the self-inductance L2 of the winding 20. Current resonance is performed by the divided C2 and the leakage inductance L1 of the winding 20. Secondary output of the transformer T1 is rectified by a diode D1, D2 and capacitor C _0, it is supplied to the smoothed by load RL. In this circuit, the voltage dividing resistors 50, 50 'or 5
1, 51 'prevents re-oscillation when the power switch SW1 is turned off.

FIG. 1 shows a specific circuit configuration of the power supply circuit shown in FIG.
It is shown in FIG. In FIG. 11, the constants added to each component are examples of specific design constants. In addition, the same reference numerals as those of the corresponding components in the above description are given to the reference numerals of each component. This specific circuit configuration is configured so that a commercial AC power supply can be used by switching between a 100 V system and a 200 V system. When a 100 V AC power supply is used, a double voltage rectification configuration is used. Since the intermediate connection point of the input smoothing capacitor can be used as a reference potential, here the series resonance capacitor (C2) is not connected to the power supply line in a two-part configuration, but is connected to the transformer winding alone.
2 '. According to such a capacitor insertion method, the withstand voltage of the capacitor itself can be reduced, and a film capacitor with a small loss in characteristics can be easily used.
This power supply circuit is designed with a rated output of 500 W (maximum 1 kW), a switching frequency of 35 kHz, a current resonance frequency of 50 kHz, and a voltage resonance frequency of 60 kHz.
And the measured self-inductance of the transformer is 2.
The measured leakage inductance was 3 μH, and the measured leakage inductance was 2.3 μH. In this specific circuit configuration, the measured efficiency of the commercial AC power supply, which is represented by the ratio of the output power extracted from the smoothing capacitor to the input power of the smoothing capacitor, is about 97% (about 80% for a normal transformer power supply, (Approx. 80-85% with power supply). Also, this power supply circuit can reduce the amount of noise generation by about 30 dB as compared with the conventional switching power supply. Specifically, this power supply circuit operates naked without a shield or the like, and performs AM radio reception nearby. Is at a practically acceptable level.

[0059]

As described above, according to the present invention, the voltage of the power supply capacitor is divided by the voltage dividing means and applied to the second capacitor of the starting circuit, so that the power switch is turned off. When the voltage of the power supply capacitor drops and oscillation stops, the voltage can not reach the voltage that turns on the main switching transistor even if the second capacitor is charged, so that restart of oscillation can be reliably prevented. Alternatively, even if the restart of the oscillation cannot be completely prevented, the number of restarts can be reduced, and the oscillation can be stopped in a short time.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing one embodiment of the present invention.

FIG. 2 is a circuit diagram showing a conventional self-excited switching inverter type power supply circuit.

FIG. 3 is a circuit diagram in which a starting circuit is added to the circuit of FIG. 2;

FIG. 4 is a schematic block diagram showing a basic principle of a power supply circuit described in Japanese Patent Application No. 3-166383.

FIG. 5 is a configuration circuit diagram showing a basic principle configuration of the power supply circuit of FIG. 4;

FIG. 6 is a timing chart for explaining the operation of the principle configuration shown in FIG. 5;

FIG. 7 is an explanatory diagram illustrating an equivalent circuit of a transformer.

FIG. 8 is a circuit diagram illustrating a modification of the principle configuration shown in FIG. 5;

FIG. 9 is a circuit diagram showing a configuration obtained by modifying the principle configuration shown in FIG. 5;

FIG. 10 is a circuit diagram showing an embodiment in which the present invention is applied to the power supply circuits described with reference to FIGS.

FIG. 11 is a specific circuit configuration diagram of the circuit of FIG. 10;

[Explanation of symbols]

 REFERENCE SIGNS LIST 1 DC power supply 15, 15 ′ starting circuit (second time constant circuit) 20 load winding 21, 22 feedback winding 25 first main switching transistor 26 second main switching transistor 27 first time constant circuit 28 1 time constant circuit 27R, 28R first resistor 27C, 28C first capacitor 29 first auxiliary switching transistor 30 second auxiliary switching transistor 40, 40 'second resistor 41, 41' second capacitor 50 , 50 ', 51, 51' Voltage dividing resistor (voltage dividing means)

Claims (1)

(57) [Claims] A DC power supply; a power switch for turning on and off the DC power supply; a power supply capacitor charged by the DC power supply when the power supply switch is on; and a main current path connected to the power supply capacitor. Are connected in series
And a second main switching transistor; a load winding connected between a connection point of the first and second main switching transistors and an intermediate potential of the power supply capacitor; and a first and second main switching transistor. The first and second main switching transistors are connected between the base and the emitter of the switching transistor, and when a current flows through the load winding from the first or second main switching transistor, the mutual induced electromotive force generated by the load winding is reduced to the first or the second. A feedback winding generated so as to provide positive feedback to the two main switching transistors; a first resistor connected in series to both ends of the feedback winding and driven by an induced electromotive force of the feedback winding; A first time constant circuit comprising a first capacitor, and a main current path connected between the base and the emitter of the first and second main switching transistors, respectively. When the first or second main switching transistor is on, the output of the first time constant circuit reaches a predetermined level and is turned on to reduce the base current of the first or second main switching transistor. Short-circuit,
A first and a second auxiliary switching transistor for turning off the first or the second main switching transistor; a series connection circuit of a second resistor and a second capacitor between the power supply capacitor and the feedback winding; To form a current path from the power supply capacitor to the load winding via the feedback winding when the main switching transistor is off, and to apply the voltage of the second capacitor to the base of the main switching transistor. An activation circuit comprising a second time constant circuit to be applied; and voltage dividing means for dividing the voltage of the power supply capacitor between the second resistor and the voltage and applying the voltage to the second capacitor. Power circuit.