JPH05191176A - High frequency power amplifier - Google Patents

Abstract

PURPOSE:To obtain a small sized and highly efficient high frequency power amplifier. CONSTITUTION:Loss due to a secondary higher harmonic can be eliminated by inserting a parallel resonance circuit 15 resonating with the secondary higher harmonic of a fundamental wave between the drain of a field effect transistor 101 and a fundamental wave matching circuit 134. Since the parallel resonance circuit 15 displays no loss for the fundamental wave and high impedance for the secondary higher harmonic, only the fundamental wave can be supplied to a load impedance 150 via the fundamental wave matching circuit 134. The parallel resonance circuit 15 can be comprised in extremely small shape, thereby, the small sized and highly efficient power amplifier can be obtained.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a transmission power amplifier for a wireless communication device, and more particularly to a small-sized high frequency power amplifier with high power efficiency.

[0002]

2. Description of the Related Art A high-frequency power amplifier of about 1 W to several W is used in a transmitter of a UHF band mobile radio to output a high power signal. In general, a mobile wireless device needs to be smaller and consume less power than a fixed wireless device. Therefore, the power amplifier that consumes most of the DC power of the wireless device is required to have a high DC-RF power conversion efficiency and a small size.

Particularly, in a portable radio telephone or the like, a power amplifier is used because the radio is small and the length of a talk time per charge of a battery is an important selling point of the product. The miniaturization and high conversion efficiency have been attempted.

The power amplifier is used in the form of an amplifier module having a gain of, for example, about 30 dB, and the module is composed of an amplifier array of 2-3 stages. Since the preamplifier on the input side is a driving stage with a low power level, the efficiency of the power conversion as an amplifier module is dominated by the efficiency of the final stage power amplifier.

The power amplifier has class A (theoretical efficiency 50%),
It is well known that there are B-class (theoretical efficiency 78.5%), C-class, and other operating modes, but in addition to this, the theoretical efficiency is 1
A theoretical study of high-efficiency operation modes called D-class, E-class, F-class, etc., which are close to 00%, is introduced in the following document by Raab.

Reference 1. Frederick H. Rabe (Fre
derick H. Raab) "Idealized Operation of the Class E Tu"
nedPower Amplifier) IEEE Circuits Syst., vol. CAS
-24, No.12, pp. 725-735 December 1977

Reference 2. Frederick H. Raab “Classification of high-efficiency amplifier circuits” Nikkei Electronics
121-146 August 23, 1976

In order to perform high efficiency amplification in the high frequency band, active elements having excellent high frequency characteristics are required. In recent years, the advent of high-speed semiconductor devices such as gallium arsenide field effect transistors (GaAs-FETs) and the market needs for higher efficiency of high-frequency amplifiers have been combined with experimental research on high-frequency amplifiers. A drain efficiency of 70 to 80% can be obtained even in the band.

The concept of high conversion efficiency operation of the high frequency amplifier will be described with reference to FIGS. 6 and 7. FIG. 6 shows the basic configuration of the power amplifier. The high frequency signal from the input terminal 121 is applied to the gate G of the field effect transistor 101 whose source S is grounded, and the amplified high frequency signal is obtained from its drain D. Then, high frequency power is supplied to the load impedance 150, which is an antenna, through the fundamental wave matching circuit 134 for impedance matching in the fundamental wave. Here, L6 and L7 are inductances, and the DC power supplies V _GS and V _DS are respectively connected to the gate G and the drain D.
Is being supplied to. Here, I _DC is a direct current to the drain D, V _D (t) is a high frequency drain voltage, I _D (t) is a high frequency current to the drain D, and I _M (t) is a high frequency current to the fundamental wave matching circuit 134. , I _OUT (t) is the high frequency current to the load impedance 150.

FIG. 7A shows a current waveform 171 of the high frequency current I _D (t) to the drain D of FIG. 6, and FIG. 7B shows a voltage waveform of the high frequency drain voltage V _D (t) of FIG. Waveform 172
It is shown. The high frequency current I _D (t) is I _D (t) = ΣI _Ck exp (jkωt) = I ₀ + I ₁ + I ₂ + I ₃ + I ₄ + ... (1) where I ₁ = Re {2I _C1 exp (jωt )} I ₂ = Re {2I _C2 exp (j2ωt)} I ₃ = Re {2I _C3 exp (j3ωt)} I ₄ = Re {2I _C4 exp (j4ωt)} ... Here, Σ represents that k is changed from −∞ to + ∞ and addition is performed, I _C represents a complex current including a real part and an imaginary part, and Re represents the real part.

Similarly, the high frequency drain voltage V _D (t) is V _D (t) = ΣV _Ck exp (jkωt) = V ₀ + V ₁ + V ₂ + V ₃ + V ₄ + ... (2) where V ₁ = Re {2V _C1 exp (jωt)} V ₂ = Re {2V _C2 exp (j2ωt)} V ₃ = Re {2V _C3 exp (j3ωt)} V ₄ = Re {2V _C4 exp (j4ωt)} ... . Here, Σ represents that k is changed from −∞ to + ∞ and addition is performed, V _C represents a complex current including a real part and an imaginary part, and Re represents the real part.
If the impedance seen from the point 181 in FIG. 6 toward the fundamental wave matching circuit 134 side is Z _M (ω) and ω = 2πf, then each current I _C and voltage V _{C in} equations (1) and (2)
_{Between, V C1 = -Z M (ω} ) I C1 V C2 = -Z M (2ω) I C2 V C3 = -Z M (3ω) I C3 V C4 = -Z M (4ω) I C4 ... Since there is a relation of (3), the high frequency current I _D (t) and the high frequency drain voltage V _D of the formulas (1) and (2) are calculated by the impedance Z _M (kω) for each harmonic seen from the point 181 to the right. It can be seen that the waveform of (t) changes.

Power loss P of field effect transistor 101
_F is P _F = T ⁻¹ ∫ V _D (t) I _D (t) dt (4) where T = 2π / ω, ω = 2πf, and ∫ is integrated from 0 to T Is represented.

Output P _OUT (1) from the drain D of the field effect transistor 101 at the frequency f ₀ of the fundamental wave
Is When ω ₀ = 2πf ₀ , P _OUT (1) = 2Re {Z _M (ω ₀ )} | I _C1 | ² (5) From the drain D of the field-effect transistor 101 at the frequency kf ₀ of the kth harmonic. Output power P _OUT (k)
Is P _OUT (k) = 2Re {Z _M (kω ₀ )} | I _Ck | ² (6)

The power consumption P _DC P _DC = V _DS I _DC = P _F + ΣP _OUT (k) (7) supplied from the DC power supply V _DS of the circuit of FIG. Here, V _DS is the voltage of the DC power supply V _DS , and Σ represents the total when changing from k = 1 to ∞. The drain efficiency η for obtaining the output power at the frequency f ₀ of the fundamental wave at the drain D of the field effect transistor 101 is η = P _OUT (1) / P _DC (8).

In order to increase the drain efficiency η shown in the equation (8) to achieve high efficiency, the power consumption P _DC is reduced and
It is necessary to increase the output power P _OUT (1) at the fundamental wave frequency f ₀ .

Therefore, in order to reduce the power consumption P _DC , it is necessary to reduce the power loss P _F of the field effect transistor 101 of the equation (7), which is represented by the equation (4).
The product of the high-frequency drain voltage V _D (t) and the high-frequency current I _D (t) of the drain D, which causes the power loss P _F shown in Fig. 7, that is, I _D (t) and V _D (t) in Fig. 7. The time overlapping part (falling part of current waveform 171a and rising part of voltage waveform 172a, falling part of 172a and rising part of 171b, falling part of 171b and rising part of 172b) to be small. So point 1 in Figure 6
It is necessary to set the impedance as viewed from 81 to the right (load side). In FIG. 7, current waveforms 171a and 171b representing I _D (t) and voltage waveform 1 representing V _D (t).
If there is no temporal overlap with 72a and 172b, the power loss P _F in equation (4) can be removed. In order to the fundamental wave (k = 1) of the k-th order harmonics except the output power P _{OU T} (k) (where k is 2 or more) 0 of the formula (7), k of formula (6) Next Of the harmonic component of Re {Z _M
(Kω ₀ )} may be set to 0, or the complex current I _Ck which is a high frequency current in the kth harmonic may be set to 0.

As a method for realizing such a condition, a high-frequency current I _D (t) of the drain D is set to a half-wave rectified waveform with a flow angle of 180 ° and an even-order impedance is set to 0, for example, a class B amplifier. Is used. The high frequency current I _D (t) of the drain D at the time of half-wave rectification is I _D (t) = 1 / π + (1/2) cos ω ₀ t + (2 / 3π) cos (2ω ₀ t) − ( 2 / 7π) cos (4ω ₀ t) + (2 / 35π) cos (6ω ₀ t) −... (9). Looking at equation (9), this high frequency current I _D (t)
Is the fundamental wave indicated by ω ₀ and, for example, 2ω ₀ , 4ω ₀ ,
Since even-order harmonic components represented by ... are included, if the impedance for even-order harmonics is set to 0, the voltage at even-order harmonics becomes 0, so the high-frequency drain voltage V
It is known that _D (t) becomes only the fundamental wave component, and the drain efficiency η shown in Expression (8) becomes high.

In practice, since the harmonic power in higher harmonics of k = 3 or more is relatively smaller than that in the case of k = 2, for the second harmonic (k = 2), the expression ( The impedance Re {Z _M (2ω ₀ )} of 6) is set to 0 to obtain a high drain efficiency η. In reality, the impedance for a frequency 2f ₀ (second harmonic) that is twice the fundamental wave as seen from the point 181 in FIG.
I am trying to realize it by bringing it closer to.

As described above, the operation of the field effect transistor 101 which is an active element involves not only the fundamental frequency but also its harmonic components. In an actual amplifier, the output of such a harmonic component must be reduced from the viewpoint of unnecessary radiation, and preventing conversion of DC power into such unnecessary wave power improves efficiency. Are also connected. For this reason, the fundamental wave matching circuit 134 must have a function of blocking harmonics.

This means that the impedance Z with respect to the harmonic seen from the field effect transistor 101 which is an active element.
(Ω) (ω = 2ω ₀ , 3ω ₀ , ..., Here, ω ₀ is the angular frequency of the fundamental wave) means that it must be almost a reactance component. That is, when expressed by the voltage reflection coefficient Γ = (Z (ω ₀ ) −Z ₀ ) / (Z (ω) + Z ₀ ), its absolute value | Γ | needs to be almost 1. Here, Z ₀ = 50Ω.

On the other hand, impedance Z related to the fundamental wave
(Ω) needs to be set to an appropriate value to output the required power. For example, when the power supply voltage is 6 V and the output power is about 2 W, Z (ω) ≈10Ω. Therefore, the fundamental wave matching circuit 134 is required to operate as a low-loss impedance that converts the load impedance 150 (impedance Z ₀ is usually 50Ω) into a predetermined impedance with respect to the fundamental wave, and also regarding the second harmonic. Also needs to satisfy the condition of | Γ | = 1.

A so-called class F operation is known as an impedance condition for realizing a highly efficient operation. This tries to make the current a half-wave rectified wave and the voltage a rectangular wave, and for even harmonics, it is short-circuited (Γ = 1 exp
(J180 °)), and the impedance Z (ω) is set so as to be open (Γ = 1 exp (j0 °)) for odd harmonics.

As a method for realizing the impedance condition for such harmonics, there is, for example, the conventional circuit shown in FIG. 8 (Japanese Patent Laid-Open No. 62-111).

In FIG. 8, C21, L21 to C23,
L23 is a series resonator C3 that resonates for even harmonics.
1, L31 to C33, L33) are series resonators that resonate with respect to odd harmonics, and each series resonator is a point 181 that is an output terminal of the field effect transistor 101 that is an active element and a load impedance 150. It is shunt-connected to the main transmission line 110 that connects between and.

The series resonator has an impedance of 0 at the resonance frequency and has a relatively high impedance at frequencies far from the resonance frequency. Therefore, the series resonator shunt-connected to the main transmission line 110 grounds the main transmission line 110 in a frequency selective manner at the connection point, and the reflection coefficient Γ seen from the point 181 is | The condition of Γ | = 1 is realized.

Field effect transistor 10 which is an active element
The impedance Z (ω) seen from 1 is shorted (Γ = 1
exp (j180 °) or open (Γ = 1 exp (j0
°), the main transmission line 11 connecting each series resonator
The position on 0 should be selected appropriately.

That is, the voltage reflection coefficient Γ (= │Γ│ exp (jθ)) seen from the point 181 which is the output terminal of the field effect transistor 101 which is an active element to the right side is the resonator connection from the point 181 which is the output terminal. Since the phase angle θ changes depending on the length of the transmission line up to the point, by properly selecting the connecting position of the resonator, θ = 180 ° for even harmonics and θ = 0 ° for odd harmonics. The condition of can be realized.

The main fundamental wave matching circuit 134, which performs the function of impedance matching with respect to the fundamental wave, is arranged behind the series resonator group. Therefore, the fundamental wave matching circuit 1 is provided so as to have a predetermined fundamental wave impedance Z (ω) after realizing the impedance condition for harmonics.
By determining 34, the impedance conditions of the harmonic wave and the fundamental wave can be set almost independently.

The operation of an actual high frequency semiconductor is complicated, and a floating circuit element such as an inductance or a capacitance is provided inside the element or in a package or the like. Therefore, the class F operation is an ideological concept, and the optimum impedance condition with high efficiency is different from the so-called class F condition. Also, it is not realistic to consider the effect of higher-order components on higher harmonics. The main frequency components of the half-wave rectified wave are the fundamental wave and the even-order harmonics, and the intensity of the even-order harmonics also decreases as the higher-order component. For this reason, the impedance for the second harmonic is dominant in improving efficiency, and it is important to select the impedance optimally. Another conventional circuit example (Japanese Patent Laid-Open No. 60-5615) implemented from this viewpoint is shown in FIG.
Shown in.

In FIG. 9, reference numeral 212 is a sub line provided in parallel so as to be coupled to the main transmission line 211 with a minute line spacing s, and has a length of about ¼ of the wavelength of the second harmonic. The sub line 212 has a central portion grounded. Capacitors C11 and C12 for termination are connected to both ends of the sub line 212 to form a resonance circuit that resonates with the second harmonic. Since it is electromagnetically coupled to the main transmission line 211, a characteristic of resonating with the second harmonic can be obtained from the right side of the point 181 which is the output terminal of the field effect transistor 101 which is an active element. By choosing the values of capacitors C11 and C12 appropriately,
The phase angle of the voltage reflection coefficient Γ (2ω ₀ ) with respect to the second harmonic can be widely changed. Therefore, the impedance condition for the second harmonic can be set so that the efficiency is maximized by appropriately selecting the values of the termination capacitors C11 and C12.

A Smith chart representing the reflection coefficient plane of the amplifier is shown in FIG. Here, the point 300 represents open and the point 301 represents short.

An output reflection coefficient Γ (2) related to the second harmonic required for high efficiency operation of an amplifier used in mobile pair communication and having an output of about 1 W to several W in the UHF band to several gigahertz band.
When the phase angle of ω ₀ ) is examined, it is in the region shown by the bold line in the reflection coefficient plane of FIG. This is a result analyzed by experiments and numerical simulation by the inventor of the present application, and the reflection coefficient indicated by the point 305 is the reflection coefficient indicated by the output circuit with respect to the actual output terminal of the active element.

The phase angle θ of the proper reflection coefficient Γ is approximately −
It lies between 120 ° <θ <0 ° (thick line in the figure). If it is within this range, the efficiency of the amplifier does not decrease so much from its maximum value (drain efficiency of about 80%), but as it goes out of this range, the efficiency drops sharply. It goes down to about%.

Regarding the optimum impedance condition of the second harmonic, similar results have been reported in the following documents by other researchers.

Reference 3. Ikeda et al., “High-efficiency operating characteristics of FETs by the double-wave injection method” Spring Conference of IEICE
C-64 1990

Reference 4. LC Hall et al., “Maximum Efficiency T Tuning of Microwave Amplifiers”
uning of Microwave Amplifiers) IEEE MIT-S Inter.
Symp. Digest 123-126 pages 1991

With the conventional circuit configuration described with reference to FIG.
A case where the reflection coefficient phase angle of the second harmonic is set within an appropriate area will be described with reference to FIG. In FIG. 11A, the capacitance C19 and the inductance L19 are series resonators that are connected to the main transmission line 110 in a shunt and resonate with the second harmonic. Load impedance 15 from the connection point of the series resonator
Since the impedance for the second harmonic viewed from the 0 side is almost short-circuit, its reflection coefficient Γ _a has an absolute value of 1 and a phase angle of 180 °. Therefore, the point 181 which is the output terminal of the field effect transistor 101 which is an active element.
Therefore, in order to obtain the optimum reflection coefficient Γ (for example, Γ = 1 exp (−j30 °)) that maximizes the conversion efficiency, the point 181 that is the output terminal of the field effect transistor 101 that is an active element is connected in series. The line length between the resonator and the resonator is l = 0.
It is necessary to rotate the phase angle of the reflection coefficient as 3125λ (where λ is the wavelength of the second harmonic). Now, taking an amplifier with a fundamental frequency of 800 MHz as an example, the relative permittivity ε _r
= 10, assuming a microstrip line (wavelength shortening rate of about 1 / 2.5) configured on a dielectric substrate of 2
The wavelength of the second harmonic (1.6 GHz) is λ = 75 mm. Therefore, a line length of about 23 mm is required between the field effect transistor 101 which is an active element and the series resonator.

A case where an appropriate second harmonic impedance is realized by the conventional circuit of FIG. 9 will be described. The phase angle of the second harmonic can be set to an optimum angle by appropriately setting the capacitances C11 and C12 for terminating the sub line 212 coupled to the main transmission line 211 of FIG. The extra line length (length of the main transmission line 110) for such phase rotation is not necessary. But 2
Since the sub-line 212, which is a coupling line, needs to have a length of approximately λ / 4 for processing the second harmonic impedance, a line length of about 19 mm is required in the case of an 800 MHz amplifier. In addition, the sub line 212 which is a coupling line
Since it is not a single transmission line, it cannot be flexibly bent to form an area-efficient pattern layout.

In order for this circuit to function as an amplifier circuit for blocking the frequency band and second harmonics, it is necessary to set 2
A sufficiently large coupling is required between the coupled lines of the book. For example, a coupled line system ((Z _OO
Z _0e ) ^1/2 = 50Ω, where Z _OO is an odd mode impedance and Z _0e is an even mode impedance), the line coupling degree between the main transmission line 110 and the sub line 212 is at least −4 dB or more. Is necessary. However, in the microstrip line, the degree of coupling obtained even if the distance s between the two coupling lines is narrowed to the limit is about -8 dB.

[0040]

In the conventional example shown in FIG. 8, a long main transmission line 1 for suppressing the generation of second harmonics.
You need 10. Since the main transmission line 110 does not contribute to the impedance matching of the fundamental wave, it is an obstacle to the miniaturization of the amplifier, which is a problem to be solved.

In the conventional example shown in FIG. 9, a coupling line of about λ / 4 is required to realize an appropriate second harmonic impedance that suppresses the generation of the second harmonic, and the degree of coupling is sufficient. There is an unsolved problem that the suppression effect is small and a sufficient conversion effect cannot be obtained because it cannot be increased.

The components for mobile communication are desired to be small, lightweight and highly efficient, and the harmonic power amplifier also needs to be integrated in an area of, for example, 10 × 10 mm. Therefore, none of the conventional examples shown in FIGS. 8 and 9 can satisfy all such demands. This is the problem to be solved by the present invention.

[0043]

[MEANS FOR SOLVING THE PROBLEMS] By connecting a parallel resonant circuit that resonates with a second harmonic in series immediately after an output terminal of an active element, a small circuit satisfying an appropriate impedance condition for the second harmonic can be obtained. Configured to achieve.

[0044]

Since the parallel resonant circuit exhibits an extremely large impedance with respect to the second harmonic, it has a great effect in suppressing the second harmonic. Since this parallel resonant circuit is easy to miniaturize, it has become possible to obtain high conversion efficiency while being compact and lightweight.

[0045]

EXAMPLE 1 Example 1 is shown in FIG. In FIG. 1A, the capacitance C15 and
The inductance L15 is a parallel resonance circuit 15 made to resonate with the second harmonic, and is a point 181 which is an output terminal of the field effect transistor 101 which is an active element.
It is connected in series with the fundamental wave matching circuit 134.

Since the parallel resonant circuit 15 including the capacitance C15 and the inductance L15 resonates with the second harmonic, the second harmonic viewed from the point 181 which is the output terminal of the field effect transistor 101 which is an active element, to the load side. The impedance Z (2ω ₀ ) for the wave is the point 19
It will have a high impedance regardless of the impedance value viewed from 1 to the right. That is, as represented by a reflection coefficient, as shown by a point 311 on the Smith chart showing the reflection coefficient plane of FIG. 1B, the second harmonic wave satisfies the harmonic blocking condition of | Γ | = 1. Moreover, since the phase angle is near 0 °, it is within the region of the proper reflection coefficient for the second harmonic, which is indicated by the bold line.

The impedance Z (ω ₀ ) of the fundamental wave
Is basically the same as the conventional circuit that can be set by the fundamental wave matching circuit 134. Parallel resonance circuit 1 for the fundamental wave
Since 5 has a slight inductive reactance X, the impedance Z ′ (ω ₀ ) = Z (ω ₀ ) −jX of the fundamental wave after subtracting the contribution of the reactance inserted in series is set by the fundamental wave matching circuit 134. It should be realized.

Comparing the circuit of the present invention with the conventional example shown in FIG. 8, for example, about 20 m required to set the phase angle of the reflection coefficient of the second harmonic within an appropriate range.
Since m main transmission lines 110 are unnecessary, the size can be reduced. Further, the circuit size is smaller than that of the conventional example shown in FIG. 9 and, of course, the conversion efficiency is lowered without causing the secondary line 212, which is a resonant circuit of the second harmonic, to be tightly coupled to the main transmission line 211. And problems such as harmonic leakage can be solved.

(Embodiment 2) FIG. 2 shows another embodiment of the present invention. In Embodiment 1 shown in FIG. 1A, the parallel resonant circuit 15 and the field effect transistor 10 as an active element are used.
A transmission line 215 (line length 1) shorter than ¼ of the wavelength λ of the second harmonic is connected between the output terminal 1 and the point 181. In the case of this circuit configuration, as shown in FIG. 2A, the field effect transistor 101 which is an active element is used.
2B as seen from FIG. 2B, the reflection coefficient Γ (2ω ₀ ) of the second harmonic of the point 313 on the Smith chart representing the reflection coefficient plane.
2 is a view seen from a point 192 immediately before the parallel resonance circuit 15.
The phase angle φ (= βl) is shifted clockwise with respect to the reflection coefficient Γ _a (2ω ₀ ) of the point 312 in (b). However, since the point 312 representing the reflection coefficient Γ _a itself is at one end of the appropriate range of the reflection coefficient for the second harmonic shown by the thick line in FIG. 2B, the reflection coefficient Γ is It is in the optimum region, and it can be seen that there is no problem in operation even if some line length is added. For example, assuming that the range of the optimum phase angle is approximately −120 ° <θ <0 °, the transmission line 215
If the line length is within the range of 0 <l <0.2λ, the reflection coefficient Γ stays within the proper range.

In actual circuit mounting, even when the parallel resonant circuit 15 and the point 181 which is the output terminal of the field effect transistor 101 which is an active element are directly connected as in the first embodiment (FIG. 1), 1-2 for connection such as soldering
Since a circuit pattern of about mm is required, the short transmission line 215 is actually connected as in the second embodiment (FIG. 2). In a high-frequency circuit, even such a connecting line may cause operational problems due to phase rotation of the reflection coefficient, but in the present invention, there is a floating line length that is inevitably generated due to mounting or the like. Also has the advantage that it does not deviate from the proper operating conditions.

On the other hand, there may be a case where the reflection coefficient Γ (2ω ₀ ) for the second harmonic is more optimally set within the proper phase angle range for the active reason of circuit design. In this case, the transmission line length 1 may be properly selected by the method of the second embodiment shown in FIG. Even in this case, the length of the transmission line 215 is not longer than ¼ of the wavelength λ of the second harmonic, so that it is smaller than the conventional circuit.

(Embodiment 3) FIG. 3 shows still another embodiment of the present invention. In a high frequency circuit, a transmission line having a relatively high characteristic impedance is generally used as a method of equivalently realizing an inductance. The third embodiment of FIG. 3 is the parallel resonant circuit 15 of the first embodiment (FIG. 1).
The inductance L15 configuring the above is replaced by the transmission line 216, and the basic operation is the same.

(Embodiment 4) As described above, the fundamental impedance seen from the field effect transistor 101 which is an active element is a relatively low impedance of, for example, about 10Ω. Therefore, the fundamental wave matching circuit 134
Is, for example, 2 from the viewpoint of wide band property and low circuit loss.
A transmission line having a low characteristic impedance of about 5Ω is suitable.

Since the transmission line having such a characteristic impedance has a wide line width (for example, the dielectric substrate thickness is 0.8 m).
m, and the relative permittivity is 10, the line width of the microstrip line is about 2.3 mm. If the parallel resonance circuit 15 is configured on this transmission line, the area of the output circuit can be further reduced and downsized. Becomes

4 and 5 show the structure and the operating principle. FIG. 4 is an equivalent circuit, and FIG. 5 is a partial structural diagram.

4 and 5, reference numeral 231 is a fundamental wave matching circuit 134 formed on the surface of the dielectric substrate 220.
Is a conductor thin film line of a micro strip line connected to the left side of. On the conductor film line 231, another dielectric substrate piece 230 having conductor thin film line pieces 221a and 222a formed on the surface thereof is arranged in close contact with the conductor film line 231. A microstrip plane circuit is formed using the conductor thin film line 231 as a ground conductor.

In this microstrip planar circuit,
Two line pieces 221a and 222 connected to a point 181 which is an output terminal of the field effect transistor 101 which is an active element and a connection line 228 and branched at the connection point.
a is arranged, one end of the line piece 221a is grounded to the line 231 which is a ground conductor by a connection line 229 to form a short stub, and one end of the line piece 222a is opened to form an open stub. ing.

The microstrip circuit on the dielectric substrate piece 230 has the field effect transistor 10 which is an active element.
It functions as a parallel resonance circuit 15 that is connected in series between the point 181 that is the output terminal of 1 and the fundamental wave matching circuit 134 and that resonates with the second harmonic. Connection point 24 in FIG.
0 and 241 correspond to the connection points 240 and 241 in FIG. 5, and the connection point 241 in FIG. 5 is in the conductor thin film portion of the line 231 immediately below the connection point 240.

The line piece 221a on the dielectric substrate piece 230,
222a is equivalent to forming line pieces 221b and 222b electrically floating from the ground conductor of the fundamental wave matching circuit 134 on the dielectric substrate 220 in the conductor thin film portion of the line 231. Since the line pieces 221b and 222b are short stubs and open stubs, the former constitutes the parallel resonance circuit 15 which acts as an inductance and the latter acts as a capacitance. Therefore, if the characteristic impedance and the length of the line are appropriately selected. It is possible to resonate with the second harmonic. Therefore, the circuit on the dielectric substrate 220 should function as the second harmonic parallel resonant circuit 15 that connects the point 181 that is the output terminal of the field effect transistor 101 that is an active element and the fundamental wave matching circuit 134 in series. I understand.

To give concrete dimensions, the thickness is 0.8 m.
1.6 GH on a dielectric substrate 220 having a dielectric constant of 10 m
When the parallel resonant circuit 15 that resonates at z (second harmonic of 800 MHz) is configured, the size of the dielectric substrate 220 is 2
mm × 7 mm. Dielectric substrate 22 of this size
0 is a size that can be sufficiently accommodated on the line of the fundamental wave matching circuit 134.

According to the present invention, not only the area occupied by the second harmonic resonance circuit itself is reduced, but also a margin for separating the second harmonic resonance circuit from the other circuit portions so as not to interfere with other circuit parts. The space for is also unnecessary, and the output circuit can be made even smaller.

(Modification of Fourth Embodiment) In the fourth embodiment shown in FIGS. 4 and 5, the line 2 of the fundamental wave matching circuit 134 is used.
In the parallel resonance circuit 15 of the microstrip formed on 31, the short stubs 221a and 221b and the open stubs 222a and 222b are used for the description. However, since the open stub is a capacitive element, it is natural that the same effect can be obtained even if the open stub is replaced with a lumped constant capacitance element such as a chip capacitor.

[0063]

As described above, according to the present invention, the circuit configuration in which the circuit that resonates in parallel with the second harmonic is connected in series between the active element and the fundamental wave matching circuit provides high efficiency. A small power amplifier can be realized. Therefore, the effect of the present invention is extremely large.

[Brief description of drawings]

FIG. 1 is a Smith chart (b) showing a reflection coefficient plane and a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a circuit diagram showing another embodiment of the present invention and a Smith chart (b) showing a reflection coefficient plane.

FIG. 3 is a circuit diagram showing still another embodiment of the present invention.

FIG. 4 is a circuit diagram showing another embodiment of the present invention.

5 is a structural diagram showing a structure of a main part of the circuit shown in FIG.

FIG. 6 is a circuit diagram for explaining the operation principle of a conventional example.

FIG. 7 is a waveform diagram showing the drain current and voltage waveforms of FIG.

FIG. 8 is a circuit diagram showing a conventional example.

FIG. 9 is a circuit diagram showing another conventional example.

FIG. 10 is a Smith chart showing a reflection coefficient plane showing the operation of the conventional example.

FIG. 11 is a circuit diagram (a) and a Smith chart (b) showing a reflection coefficient plane in the case of optimizing the reflection coefficient for the second harmonic in the conventional example.

[Explanation of symbols]

15 parallel resonance circuit 101 field effect transistor 110 main transmission line 121 input terminal 134 fundamental wave matching circuit 150 load impedance 171 current waveform 172 voltage waveform 181, 191, 192 points C11, C12, C16, C19, C21 to C23, C31 to C33 Capacitance I _DC DC current I _D (t), I _M (t), I _OUT (t) High frequency current L6, L7, L15, L19, L21 to L23, L31 to L33 Inductance 211 Main transmission line 212 Sub line 215, 216 Transmission line 220 Substrate 221a, b, 222a, b Line piece 228,229 Connection line 230 Dielectric substrate piece 231 Line 240,241 Connection point 300,301,305 points s Line spacing V _DS , V _GS DC power supply V _D (t ) High frequency drain voltage

Claims (3)

[Claims] 1. A power amplifying means (101) for power-amplifying a high-frequency signal to obtain high-frequency power at an output terminal, and an output impedance of the power amplifying means is converted into a load impedance (150) at a fundamental frequency. And a parallel resonance means that is connected in series between the output terminal of the power amplification means and the fundamental wave matching means and that resonates in parallel with a frequency twice the fundamental wave frequency. A high frequency power amplifier including (15). 2. The high frequency power amplifier according to claim 1, wherein the parallel resonance means includes a transmission line (216, 221a, 221b, 229) and a capacitance connected in parallel to the transmission line. 3. The fundamental wave matching means includes a line (231) having a microstrip structure for transmitting the fundamental wave and a conversion unit for impedance conversion, and the parallel resonance means includes: The microstrip structure is formed by using the line of the microstrip structure as a ground conductor.
High frequency power amplifier.