JPH0518292U - Resonant switching power supply - Google Patents

Abstract

(57) [Abstract] [Purpose] In a resonant switching power supply, while ensuring zero voltage switching and suppressing frequency fluctuations,
Enables PWM output voltage control. [Structure] Apart from a first switch S1 for output control provided on the primary side, a secondary winding 22 of a transformer 2 is provided on the secondary side.
A second switch S2 for connecting and disconnecting the output circuit 5 and the output circuit 5 is connected, and a control means 8 for synchronously turning on and off the first and second switches S1 and S2 is provided. This prevents the resonance on the primary side from being adversely affected by the secondary side.

Description

[Detailed description of the device]

[0001]

[Industrial applications]

 The present invention relates to a forward type resonant switching power supply.

[0002]

[Prior Art]

 This type of resonance type switching power supply has an advantage that switching loss is small and therefore noise accompanying switching is also small. A conventional example thereof is shown in FIG. 4 (see Japanese Patent Laid-Open No. 3-135367). In the figure, a primary winding 21 of a transformer 2 and a capacitor (capacitor) C1 are connected in series to a DC power supply 1, and a resonance circuit 3 is formed by the primary winding 21 and the capacitor C1. .. A diode D1 and an output control switch S1 are connected in parallel to the capacitor C1. The output circuit 5 is connected to the secondary winding 22 of the transformer 2 via the saturable reactor L1.

The output circuit 5 includes two diodes D2 and D3, a choke L2 and a smoothing capacitor C2. When the switch S1 is on, the diode D2 conducts to output the energy of the secondary winding 22. When the switch S1 is off, the diode D3 conducts and the energy stored in the choke L2 is released to obtain a DC output at the output terminals 6 and 6.

The signal waveform in the above circuit is as shown in FIG. In the figure, when the switch S1 is turned on, a current flows through the primary winding 21 of the transformer 2 to store energy. The polarities of the two windings 21 and 22 at this time are as shown in FIG. 4, and the diode D2 on the secondary side is made conductive to discharge the energy of the secondary winding 22 to the output circuit 5. In other words, it is the forward method.

When the switch S1 is turned off from this state (FIG. 5 (d)), the energy stored in the primary winding 21 charges the capacitor C1 and the voltage V1 across the capacitor C1 rises (FIG. 5 (a)). , Then discharge and V1 drops. At this time, the diode D2 on the secondary side is non-conductive. When the capacitor voltage V1 falls below the voltage Vin of the DC power supply 1, the secondary winding 22 returns to the polarity shown in FIG. 4, and the diode D2 tries to conduct again. However, when the diode D2 becomes conductive, the secondary side becomes low impedance, and the discharge of the capacitor C1 becomes insufficient. As a result, the capacitor voltage V1 does not fall to the zero level and the switch S1 cannot be turned on at zero voltage. become. Therefore, a saturable reactor L1 is provided on the secondary side to keep the secondary side at a high impedance and sufficiently discharge the capacitor C1 to lower the capacitor voltage V1 to the zero level.

On the other hand, the capacitor voltage V1 passes through the zero point and tries to become negative as shown by the broken line, and a negative diode current I2 is passed through the diode D1 (FIG. 5 (c)). The slope of the diode current I2 decreases under the influence of the saturable reactor L1 on the secondary side, and when I2 = 0, the capacitor voltage V1 rises again. Therefore, before that, that is, during a short period TS. Then, the switch S1 is turned on to realize zero voltage switching.

[0007]

[Problems to be solved by the device]

 By the way, the output voltage Vout in the switching power supply is generally controlled by PWM control. At this time, in order to control the output voltage Vout to be constant even when the input voltage Vin fluctuates greatly, the power feeding time to the primary winding 21, that is, the on-time A of the switch S 1 (FIG. 5 (d)) is increased. Although it needs to be changed, as described above, the period TS is short, so that the turn-on time t1 of the switch S1 cannot move much back and forth. Therefore, the turn-off time point t2 of the switch S1 must be greatly moved. On the other hand, the off period B of the switch S1 is determined by the charging / discharging time of the resonance circuit 3. As a result, when the input voltage Vin becomes low or the load becomes large, the switching cycle T = A + B becomes long and the frequency becomes low, which is disadvantageous for downsizing the transformer 2 and the smoothing capacitor C2. In other words, with a small switching power supply, it is difficult to control the output voltage by PWM while suppressing frequency fluctuations and ensuring zero voltage switching, even for large fluctuations in input voltage and load. ..

Further, when the switch S1 is turned off, the switching current I1 falls not slightly vertically but with a slight inclination because of the circuit time constant. On the other hand, immediately after the switch S1 is turned off, the impedance of the secondary side is low because the diode D2 is still in the conductive state due to the inertia of the secondary winding 22. Therefore, the inductance of the primary winding 21 is apparently low, and the resonance frequency is high. As a result, as shown in FIG. 6, the capacitor voltage V1 rises steeply with respect to the sloping switching current I1, so that a switching loss represented by a shaded portion F where the two intersect is generated. That is, switching loss exists even in the resonance type.

Further, since the saturable reactor L1 is required, the circuit is large and expensive.

The present invention has been made in view of the above-mentioned conventional problems. The output voltage control by PWM is performed while ensuring zero-voltage switching and suppressing frequency fluctuations even with large fluctuations in the input voltage. In addition, the objective is to provide a resonant switching power supply with less switching loss.

[0011]

[Means for Solving the Problems]

 In order to achieve the above object, the present invention connects and disconnects the secondary winding and the output circuit on the secondary side, in addition to the first switch for output control provided on the primary side. A control means is provided for connecting the second switch and for turning on and off the first and second switches in synchronization.

[0012]

[Action]

 According to this invention, since the first switch on the primary side and the second switch on the secondary side are turned on and off at the same time, the resonance that occurs during the off period of the first switch affects the secondary side. Therefore, the turn-on possible period at zero voltage can be set longer. Therefore, the turn-off time of the first switch is set to be constant, and the off-time is set to be longer than the charge / discharge time of the resonant circuit, while the turn-on time is set within the period in which zero voltage switching is possible. By making a large movement, the output voltage can be controlled to be constant with respect to large fluctuations in the input voltage and load. As a result, it is possible to control the output voltage by PWM at a constant frequency while ensuring zero voltage switching.

Further, since the resonance is not affected by the secondary side as well, the rise of the capacitor voltage at the time of turn-off can be made gradual, so that the crossing between the switching current and the capacitor voltage is reduced and the switching loss is reduced. descend. Furthermore, the saturable reactor is no longer needed.

[0014]

【Example】

 An embodiment of the present invention will be described below with reference to the drawings. In FIG. 1, a resonance circuit 3 formed by a primary winding 21 of a transformer 2 and a capacitor C1 is provided on the primary side connected to a DC power supply 1, and the capacitor C1 is provided for output control. The first switch S1 and the diode D1 are connected in parallel. An output circuit 5 for extracting an output from the secondary winding 22 of the transformer 2 is provided on the secondary side, and a second switch S2 for connecting and disconnecting the secondary winding 22 and the output circuit 5 is further connected. Has been done.

The control means 8 has a PWM IC that outputs a PWM signal to control both switches S1 and S2, and receives a detection signal from the output voltage detection circuit 9 to switch both switches S1 and S1. The output voltage Vout is controlled to be constant with respect to changes in the input voltage Vin and the load 10 by synchronously turning on and off S2 and performing PWM control. FETs, for example, are used as the switches S1 and S2.

The operation of the above circuit will be described with reference to the signal waveforms shown in FIG. (1) Time point t1-t2: First, when the switch S1 is turned on, the switch S2 is also turned on, a current I5 flows to the secondary side, and the smoothing capacitor C2 is charged through the choke L2. When the number of turns of both windings 21 and 22 is Np, Ns, the current I1 on the primary side (equal to the switching current) is proportional to the current I5 on the secondary side, Ii = I5 × Ns / Np, Excitation current Ie = Vin / Lp × (t−t0) + Ieo, and I1 = Ii + Ieo. Here, Lp is the inductance of the primary winding 21, and Ieo is the initial value of the diode current I2 described later.

(2) Time point t2-t3: At time point t2, both switches S1 and S2 are turned off, and the inductance L p of the primary winding 21 and the capacitance Cr start resonance. At this time, since the second switch S2 is off and the secondary winding 22 of the transformer 2 and the output circuit 5 are cut off, the second switch S2 resonates without being affected by the secondary side. Therefore, since the secondary side is kept at high impedance, the capacitor voltage V1 can drop to zero level without the conventional saturable reactor, and the resonance inductance at this time is Lp. . Therefore, the capacitor voltage V1 becomes a resonance of V1 = Vin + (Iep ² Lp / Cr + Vin ² ) ^1/2 · sin (ωt−α).

(3) Time point t3-t4 (t0-t1): When the capacitor voltage V1 becomes zero level, the diode D1 becomes conductive and the diode current I2 flows. The initial value Ieo of the diode current I2 is −Ieo = Iep when the primary winding current is Iep, and thus I2 = Vin / Lp × (t−t3) −Iep. If both switches S1 and S2 are turned on before I2 = 0, zero voltage switching is realized.

At the time ta when the diode current I2 = 0, the initial value Ieo of the diode current I2 and the primary winding current Iep have a relationship of −Ieo = Iep, as described above. As shown in (c), ta is the midpoint between t0 and t2. In other words, the zero voltage turn-on possible period TS is (t2-t0) / 2, which can be longer than in the conventional case. Therefore, by fixing the turn-off time point t2 and moving the turn-on time point t1 back and forth, the on-period A, that is, the power supply time to the primary winding 21 is adjusted, and large fluctuations in the input voltage Vin and the load 10 are caused. Also, the output voltage Vout can be controlled to be constant. In this way, if the turn-off time point t2 is fixed and the off period B is set to be longer than the charging / discharging time of the resonance circuit 3, the off period B is charged. Since it can be freely adjusted within the range of not shorter than the discharge time, the time between the off times t2 and t2, that is, the cycle T can be made constant regardless of the length of the on period A 1. That is, the frequency can be kept constant.

Further, as shown in FIG. 3, when the switch S1 is turned off, the switching current I1 drops with a slight inclination as in the conventional example. However, since the secondary winding 22 is opened immediately after the switch S1 is turned off, the impedance on the secondary side becomes high. Therefore, since the inductance of the primary winding 21 is kept large, it does not affect the resonance frequency to increase it, and the resonance frequency becomes an original value determined only by the circuit constant of the resonance circuit 3. As a result, the capacitor voltage V1 rises gently, and the switching loss represented by the shaded portion F1 where the switching current I1 and the capacitor voltage V1 intersect becomes small.

In the above embodiment, both the switches S1 and S2 are turned off at the same time, but unlike this, when the first switch S1 is turned off after a slight delay after the second switch S2 is turned off, switching is performed. Loss can be zero. That is, when the second switch S2 is turned off, the primary-side current I1 falls, and when the first switch S1 is turned off, the capacitor voltage V1 rises. As can be seen from FIG. 3, the primary-side current I1 starts to fall. By making the timing ti of (1) earlier than the timing tv of the rising start of the capacitor voltage V1, the switching loss represented by the shaded portion F1 can be made zero.

[0022]

As described above, according to the present invention, since the resonance occurring in the off period B of the first switch S1 is not influenced by the secondary side, zero voltage switching is ensured and the frequency is changed. It is possible to control the output voltage with PWM even when the input voltage Vin and the load largely change while controlling the output voltage to be constant. In addition, since the resonance is not affected by the secondary side, the intersection between the switching current I1 and the capacitor voltage V1 at the time of turn-off t2 is reduced and the switching loss is reduced. Further, since the saturable reactor is unnecessary, the circuit is small and inexpensive.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a signal waveform diagram of the circuit of the embodiment.

FIG. 3 is a waveform diagram showing enlarged waveforms of a switching current and a capacitor voltage in the circuit of the embodiment.

FIG. 4 is a circuit diagram showing a conventional example.

FIG. 5 is a signal waveform diagram of the circuit of the conventional example.

FIG. 6 is an enlarged waveform chart showing waveforms of a switching current and a capacitor voltage in the circuit of the conventional example.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... DC power supply, 2 ... Transformer, 3 ... Resonance circuit, 5 ... Output circuit, 8 ... Control means, 10 ... Load, 21 ... Primary winding, 2
2 ... Secondary winding, C1 ... Capacitor, S1 ... First switch, S2 ... Second switch.

Claims (1)

[Scope of utility model registration request] 1. A resonance circuit formed by a primary winding of a transformer and a capacitor on a primary side connected to a DC power supply,
In a resonance type switching power supply having a first switch for output control and an output circuit for extracting an output from a secondary winding of a transformer on a secondary side, a connection between the secondary winding and the output circuit And a second switch for shutting off and a control means for synchronously turning on and off the first and second switches.