JPH05175749A - Power amplifier - Google Patents

Abstract

PURPOSE:To make the linearity and high efficiency compatible for a final stage power amplifier of a radio equipment by eliminating generated distortion through a negative feedback loop while operating the power amplifier in the nonlinear region offering a high efficiency. CONSTITUTION:A frequency conversion means 4 frequency-converts a transmission frequency with a 1st reception local signal with a frequency being a difference between that of a transmission local signal and a 1/n frequency division signal of a reference signal. Then an intermediate frequency signal with a frequency being 1/n of that of a reference signal is generated. A 2nd reception local signal generating means 5 obtains a phase difference between a phase of the intermediate frequency signal and a phase of an input base band signal by means of the reference signal. Then a 2nd reception local signal with a frequency being 1/n of the frequency of the reference signal having a prescribed phase difference from the input base band signal is generated and demodulated by an orthogonal demodulator 2. Thus, the phase of the local signal fed to an orthogonal demodulator 2 applying orthogonal detection to the modulation wave signal is always controlled to the optimum state, and an ideal distortion compensation characteristic is obtained.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power amplifier in a wireless device or the like, and more particularly to a linear amplifier and high efficiency in a final stage power amplifier of the wireless device using a digital linear modulation system such as π / 4 shift QPSK modulation. The present invention relates to a power amplifier designed to be compatible with each other.

In a radio device such as a car phone and a mobile phone, a digital linear modulation system such as π / 4 shift QPSK modulation having high frequency utilization efficiency is used from the viewpoint of effective utilization of radio frequency. In the final stage power amplifier of a wireless device using such a linear modulation system, it is necessary that the linearity is good because of its characteristics.

In mobile communication equipment, it is required that the device be small and lightweight, and that it be used for a long time. However, the battery used for the power source occupies a large ratio in terms of volume and weight of the device. Therefore, in order to reduce the size of the battery and prolong the usage time, it is required to improve the efficiency of the power amplifier of the transmitter.

Therefore, there is a demand for a power amplifier capable of achieving both of these characteristics by operating the power amplifier in a highly efficient non-linear region and removing the generated distortion by a negative feedback loop. There is.

[0005]

2. Description of the Related Art In a linear modulation system, the linearity of a power amplifier greatly affects the spectrum characteristic of a modulated wave.
In order to realize the narrow spectrum property of the linear modulation method,
A power amplifier with good linearity is required.

However, in general, efficiency and linearity of an amplifier are contradictory characteristics. Therefore, by operating the power amplifier in a non-linear region with good efficiency but poor linearity and removing the generated distortion by applying appropriate feedback, distortion compensation that balances the conflicting requirements of linearity and high efficiency. A method is needed. The Cartesian method has been conventionally known as one of such distortion compensation methods.

FIG. 10 shows a conventional Cartesian power amplifier. A quadrature modulator that quadrature modulates a local signal with the input baseband signals I and Q to generate a modulated wave signal, 12 a local oscillator that generates a local signal of the quadrature modulator 11, and 13 amplifies and transmits the modulated wave signal. An output power amplifier, 14 is a directional coupler for extracting a part of the transmission output, 15 is a demodulator for quadrature demodulating the output of the directional coupler 14, and 16 is a phase shifter for the local signal of the local oscillator 12. Phase shifters 17 and 18 are supplied to the quadrature demodulator 15, subtractors 17 and 18 are feedback amplifiers 19 and 20 that multiply the outputs of the subtractors 17 and 18 by K, and adders 21 and 22 are adders.

The quadrature modulator 11 quadrature modulates the local signal of the local oscillator 12 with the input baseband signals I and Q to generate a modulated wave signal such as a π / 4 shift QPSK modulation signal. The power amplifier 13 amplifies the modulated wave signal to generate a transmission output.

A part of the transmission output is extracted through the directional coupler 14, and the extracted modulated wave signal is quadrature demodulator 15
In the phase shifter 1
Quadrature detection is performed using the signal that has undergone phase shift via 6 to obtain a demodulation output composed of an in-phase component and a quadrature component.

In the subtractor 17, the in-phase component of distortion is obtained by obtaining the difference between the in-phase component having distortion from the quadrature demodulator 15 and the in-phase component I of the input baseband signal without distortion. Similarly, in the subtractor 18, the quadrature component of distortion is obtained by obtaining the difference between the quadrature component having distortion from the quadrature demodulator 15 and the quadrature component Q of the input baseband signal without distortion.

The in-phase component and the quadrature component of the obtained distortion are amplified by the feedback amplifiers 19 and 20 and multiplied by K, respectively, and the in-phase component I and the quadrature component Q of the input baseband signal are respectively added by the adders 21 and 22. Add to and. As a result, negative feedback is performed on the in-phase component and the quadrature component, respectively, so that both amplitude distortion and phase distortion are compensated.

[0012]

In the Cartesian power amplifier, the quadrature modulator that creates a modulated wave signal and the quadrature demodulator that demodulates the modulated wave signal are required in order to correctly perform distortion compensation of the power amplifier. , It is necessary that the local signals have the same phase, and if there is a phase difference between the local signals, the in-phase component and the quadrature component of the distortion will not be fed back correctly, so correct distortion compensation will be performed. Disappear.

For this reason, in the conventional Cartesian power amplifier shown in FIG. 10, a phase shifter 16 is provided between the output of the local oscillator and the quadrature demodulator so that the phases of both are matched. ..

However, since the phase rotation in the power amplifier greatly changes depending on the input level, the used frequency, the temperature, the secular change, etc., the fixed phase shifter cannot always obtain the optimum local phase. In the Cartesian power amplifier of, there is a problem that it is difficult to realize an ideal distortion compensation characteristic.

The present invention is intended to solve the problems of the prior art, and in a Cartesian power amplifier, the phase of a local signal supplied to a quadrature demodulator for quadrature detection of a modulated wave signal is always optimized. It is an object of the present invention to provide a power amplifier that can be controlled in various states and therefore can realize an ideal distortion compensation characteristic.

[0016]

FIG. 1 shows a principle configuration (1) of the present invention. The present invention has a quadrature modulator 1 that quadrature-modulates a transmission local signal synchronized with a reference signal by an input baseband signal expressed in quadrature coordinates and creates a modulated wave signal. The modulated wave signal is amplified and transmitted. In a power amplifier 3 that generates an output and feeds back a signal obtained by demodulating a transmission output through a quadrature demodulator 2 to an input baseband signal, a transmission output frequency is a transmission local signal and a reference signal is n (n is a positive integer). Frequency conversion means 4 for frequency-converting the first received local signal having a frequency difference from the frequency-divided signal to generate an intermediate frequency signal having a frequency of 1 / n of the reference signal; The phase difference between the input baseband signal and the input baseband signal is calculated by counting the reference signal as the master clock, A second reception local signal generating means 5 for generating a second reception local signal having a frequency of 1 / n of the reference signal having a difference is provided, and the quadrature demodulator 2 demodulates the second reception local signal. It is a thing.

In the present invention, the first received local signal transmits the frequency divider 27 which divides the reference signal by n, the n divided signal and a signal obtained by shifting this signal by 90 °. Image rejection for generating a first reception local signal having a frequency difference between the transmission local frequency and the frequency-divided n signal by multiplying and adding the local signal and a signal obtained by phase-shifting this signal by 90 °. And the mixer 28.

In the present invention, the transmitted local signal is produced by the synthesizer 26 which operates in synchronization with the reference signal.

Further, in this case, the present invention is provided with a changeover switch 34 at the input of the frequency conversion means 4 and inputs from the input side of the power amplifier 3 to the frequency conversion means 4 in the closed loop state.

In this case, the present invention provides a variable band type band pass filter 33 on the output side of the second received local signal generating means 5 to make the band wide at the time of burst rise and narrow the band thereafter.

FIG. 2 shows the basic configuration (2) of the present invention. The present invention has a quadrature modulator 1 that quadrature-modulates a local signal with an input baseband signal expressed in Cartesian coordinates to create a modulated wave signal, amplifies the modulated wave signal to generate a transmission output, and In the power amplifier 3 for feeding back a signal obtained by demodulating the transmission output through the quadrature modulator 2 to the input baseband signal, a variable phase shifter 6 for phase-shifting the local signal so as to have the same phase as the transmission output, A fixed phase shifter 7 for shifting the output of the variable phase shifter 6 by a fixed amount of phase is provided, and the quadrature demodulator 2 demodulates the output of the fixed phase shifter 7 as a local signal.

Further, the present invention provides the variable phase shifter 6 in this case.
Has an infinite phase shifter 45 for shifting the local signal, a mixer 47 for multiplying the output of the infinite phase shifter by the transmission output, and a maximum value detection circuit 48 for detecting the maximum value of the mixer output. The output of the maximum value detection circuit 48 controls the amount of phase shift of the infinite phase shifter 45.

Further, in the present invention, the maximum value detection circuit 48 generates the phase correction signal for the infinite phase shifter 45 at every constant calculation cycle, and the phase correction amount in the phase correction signal is set at the start of the phase control. It is controlled so that it becomes larger and becomes smaller thereafter.

[0024]

In the principle configuration (1), the transmission local signal synchronized with the reference signal is quadrature-modulated by the input baseband signal expressed in quadrature coordinates to create a modulated wave signal,
In a power amplifier that amplifies this modulated wave signal to generate a transmission output and feeds back a signal obtained by orthogonally demodulating the transmission output by a second reception local signal to an input baseband signal, the transmission output frequency is set to a transmission local signal and a reference signal. Is frequency-converted by a first received local signal having a frequency difference from a signal obtained by dividing n by (n is a positive integer),
An intermediate frequency signal having a frequency of 1 / n of the reference signal is created, and the phase difference between the phase of this intermediate frequency signal and the input baseband signal is calculated according to the phase difference obtained by counting the reference signal as a master clock. 1/1 of the reference signal having a predetermined phase difference from the phase of the input baseband signal by controlling
A second received local signal having a frequency of n is generated, and quadrature demodulation is performed by the second received local signal.

Thus, according to the principle configuration (1),
Since the phase of the local signal supplied to the quadrature demodulator for quadrature detection of the modulated wave signal can always be controlled to the optimum state, the quadrature demodulator always performs demodulation in the same phase as the quadrature modulator. Therefore, the feedback of the in-phase component and the quadrature component at the input of the quadrature modulator is correctly performed, and the distortion compensation of the power amplifier can be correctly performed.

In this case, the first received local signal is
The reference signal is frequency-divided by n, and the image rejection mixer 28 multiplies the frequency-divided n signal and the signal obtained by phase-shifting this signal by 90 ° to the transmission local signal and the signal obtained by phase-shifting this signal by 90 °, respectively. It can be created by adding and adding.

In this case, the synthesizer 26 which operates the transmission local signal in synchronization with the reference signal,
Can be created.

Further, in this case, by inputting the input from the input side of the power amplifier to the frequency converting means in the closed loop state to generate the intermediate frequency signal, the transition of the signal phase is measured even in the closed loop state, The phase shift of the second received local signal can be detected.

Further, in this case, a variable band type bandpass filter 33 is provided on the output side of the second received local signal so as to have a wide band at the time of burst rising and a narrow band thereafter, so that the second band is provided at the time of burst rising.
It is possible to prevent the deterioration of the spectrum by setting the phase of the reception local signal quickly and then setting the phase of the second reception local signal to be smooth.

In the principle configuration (2), the local signal is orthogonally modulated by the input baseband signal expressed in orthogonal coordinates to create a modulated wave signal, and this modulated wave signal is amplified to generate a transmission output. , In a power amplifier that feeds back a signal obtained by quadrature demodulating the transmission output to an input baseband signal, variably phase-shifts the local signal so that it has the same phase as the transmission output, shifts this output by a fixed amount of phase, and outputs this output locally. Quadrature demodulation is performed as a signal.

Therefore, even in the case of the principle configuration (2), the phase of the local signal supplied to the quadrature demodulator for quadrature detection of the modulated wave signal can always be controlled to the optimum state, and the quadrature demodulator can always control the phase. Since demodulation is performed in the same phase as the quadrature modulator, the in-phase component and the quadrature component at the input of the quadrature modulator are correctly fed back, and the distortion compensation of the power amplifier can be correctly performed.

In this case, the maximum value of the output obtained by multiplying the output of the infinite phase shifter for shifting the local signal and the transmission output by the mixer is detected, and the phase shift amount of the infinite phase shifter is controlled by this detected output. By doing so, the local signal can be phase-shifted to the same phase as the transmission output.

Further, in this case, when the maximum value for controlling the phase shift amount of the infinite phase shifter is detected, a phase correction signal for the infinite phase shifter is generated at every constant calculation cycle, and the phase correction signal is generated. By controlling the amount of phase correction in the correction signal to be large at the start of phase control and gradually decreasing thereafter, it is possible to satisfy the conflicting requirements of phase jitter and pull-in time when pulling the phase of the local signal to the optimum value. be able to.

[0034]

FIG. 3 shows an embodiment of the present invention, in which the same parts as those in FIG.
Reference numeral 5 is a reference oscillator, 26 is a synthesizer that operates in synchronization with the reference oscillator 25, and 27 is a frequency divider (1 / then that divides the output frequency of the reference oscillator 25 into 1 / n (n is a positive integer)).
n), 28 is an image rejection mixer, 29 is a mixer for converting a modulated wave signal into an intermediate frequency signal, 30 is an intermediate frequency amplifier for amplifying the intermediate frequency signal, 31 is a limiter, 32 is a 2nd local part, 33 is a variable band Shape band pass filter, 34 is a changeover switch.

The quadrature modulator 11 quadrature modulates the transmission local signal f _T from the synthesizer 26 with the input baseband signals I and Q to generate a modulated wave signal such as a π / 4 shift QPSK modulation signal. The power amplifier 13 amplifies the modulated wave signal to generate a transmission output. The synthesizer 26 uses the reference oscillator 2
The transmission local signal f _T is synchronized with the reference signal nf _{IF of} 5
To occur.

A part of the transmission output is extracted through the directional coupler 14, and the extracted modulated wave signal is converted into an intermediate frequency signal f _IF in the mixer 29 by using the 1st local signal f _T -f _IF. .. This signal is transferred to the intermediate frequency amplifier 30
Then, the quadrature demodulator 15 performs quadrature detection using the 2nd local signal f _IF to obtain a demodulation output including an in-phase component and a quadrature component.

In the subtractor 17, the in-phase component of distortion is obtained by obtaining the difference between the in-phase component having distortion from the quadrature demodulator 15 and the in-phase component I of the input baseband signal without distortion. Similarly, in the subtractor 18, the quadrature component of distortion is obtained by obtaining the difference between the quadrature component having distortion from the quadrature demodulator 15 and the quadrature component Q of the input baseband signal without distortion.

The in-phase component and the quadrature component of the obtained distortion are amplified by the feedback amplifiers 19 and 20 and multiplied by K, respectively, and the in-phase component I and the quadrature component Q of the input baseband signal are respectively added by the adders 21 and 22. Add to and. As a result, negative feedback is performed on the in-phase component and the quadrature component, respectively, so that both amplitude distortion and phase distortion are compensated.

At this time, the reference signal nf _{IF of the} reference oscillator 25
And a frequency divider 27 at 1 / n the divided signal f _IF, a signal which was 90 ° phase shift, by adding to the image rejection mixer 28, internally, a transmission local signal f _T , And the 1st local signal f _T −
Create f _IF .

In the 2nd local section 32, the phase of the signal obtained by limiting the amplitude of the intermediate frequency signal f _IF by the limiter 31 with reference to the input baseband signal is used as the reference signal nf _IF.
Is used as the master clock. The phase of the 2nd local signal f _IF given to the quadrature demodulator 15 is adjusted so that the phase of the intermediate frequency signal thus obtained matches the phase of the transmission modulated wave signal which is obtained in advance.

Thus, in this embodiment, the quadrature demodulator 15 is used.
Since the demodulation is always performed in the same phase as the quadrature modulator 11, the feedback of the in-phase component and the quadrature component at the input of the quadrature modulator 11 is correctly performed.

At this time, the modulated wave signal is converted into an intermediate frequency signal.
1st local signal to be converted and 2n to be given to the quadrature modulator
If the d local signal is asynchronous, the phase synchronization operation is always performed.
However, the reference signal nf_IFTransmit local synced to
Signal f_TAnd the reference signal nf _IFSignal f obtained by dividing_IFAnd
1st local signal f_T-F_IFIs creating
Therefore, such phase synchronization operation is unnecessary.

In this case, when simply multiplied to create this signal, the resulting signal is the signal f _IF at the DSB (Doub
le Side Band) The modulated signal has a frequency of f _T ± f _IF . Modulated wave signal is generally 1 GHz
Although it becomes a high frequency, the signal f _IF cannot be made too high because it has a digitally processable frequency. Therefore, both frequencies resulting from the multiplication are very close to each other, and it is difficult to separate them by using a filter. Is. Therefore, in the present embodiment, such a problem is solved by using the image rejection mixer.

FIG. 4 shows an image rejection mixer in which PS1 and PS2 are 90 ° phase shifters,
MIX1 and MIX2 are mixers, and ADD is an adder.
A signal f _IF and the signal f _T multiplied by a mixer MIX1, mixer and a signal was 90 ° phase shifted signal f _IF with a phase shifter PS1 and a signal signal the f _T and the 90 ° phase in the phase shifter PS2 MIX2
And the results of both multiplications are added by the adder ADD to obtain a signal having a frequency f _T −f _IF which is the difference between the two signals.

Further, the master clock for generating a signal f _IF is used a reference signal nf _IF synthesizer for generating a transmission local signal f _T. Therefore, since the 1st local signal f _T −f _IF , the signal f _IF, and the transmission local signal f _T are all synchronized, the 2nd local unit 3
It is sufficient that the measurement and the phase setting of the local signal phase in 2 are performed at a certain time interval to follow the temperature change.

In the 2nd local section 32, when the signal f _IF is digitally controlled during transmission, the phase of the 2nd local signal is one clock of the master clock nf _IF ,
Since it changes discontinuously, the transmission spectrum deteriorates.

2nd from the 2nd local section 32
By providing the band-pass filter 33 to the local signal output, the phase change of the 2nd local signal f _IF can be smoothly generated when the local phase is corrected.

On the other hand, when the burst rises, 2nd
If the phase change of the local signal f _IF is not rapid, the TDMA
Problems arise when the system transmits burst modulated waves. Therefore, in the present embodiment, the bandpass filter 33 is a variable bandpass bandpass filter, which is set to a wide band at the time of burst rise to quickly set the phase of the 2nd local signal and then narrow band to set the 2nd local. Make the signal phase smooth to prevent spectrum degradation.

When the Cartesian loop is closed, the loop operates so as to eliminate the phase error at the output of the quadrature demodulator, so that the local phase shift is compressed by the loop gain at the output of the power amplifier. It becomes impossible to measure the phase shift in the local part.

Therefore, in the closed loop state, the changeover switch 34 is changed over so that the modulated wave signal at the input side of the power amplifier 13 is fed back to the demodulating side. On the input side of the power amplifier 13, a signal in which the amount of phase rotation by the power amplifier 13 has been inversely corrected can be monitored. Therefore, the 2nd local phase shift is detected by measuring the signal phase transition at this point. can do.

FIG. 5 shows an example of the configuration of the 2nd local section, in which 35 is a presettable counter and 3 is a presettable counter.
Reference numerals 6 and 37 are latches, 38 is a load value calculation unit, 39 is a gate, 40 is a rising edge detection unit, 41 is a phase measurement timing generation unit, and 42 is a phase setting timing generation unit.

The presettable counter 35 has k bits.
Load value D_0,D_1,…_,D_k-1From the master clock n
f_IFTo count the k-bit count value Q_0,Q
_1,…, Q _k-1Is output and the key is
Output the second local signal.

The rising edge detecting section 40 is composed of a differentiator, and detects the edge of each rising edge of the intermediate frequency signal to generate an output. The latch 36 latches the count value of the presettable counter 35 according to the output of the rising edge detection unit 40.

The phase measurement timing generator 41 generates a signal indicating the timing at which the phase measurement should be performed, in accordance with the symbol clock indicating the timing of the input baseband signal. The latch 37 responds to this signal by the latch 36.
Latch the value of.

The load value calculator 38 presets a count value corresponding to the phase of the 2nd local signal corresponding to the signal point in the quadrature demodulator 15 when there is no phase shift of the 2nd local signal, and latches it. According to the output of 37, the load value of the presettable counter 35 necessary for compensating the phase shift of the 2nd local signal is calculated. Therefore, the load value is 0 when there is no phase shift of the 2nd local signal, but becomes a value corresponding to the phase shift when there is a phase shift.

The phase setting timing generator 42 generates a signal indicating the timing at which the load value should be set in the presettable counter 35 in response to the symbol clock. The gate 39 opens in response to this signal, and supplies the load value obtained by the load value calculator 38 to the presettable counter 35.

The presettable counter 35 starts counting the master clock nf _IF from the load value and outputs the 2nd local signal when the count is completed. In this way, the presettable counter 35 changes its count value according to the phase shift of the 2nd local signal for each cycle of the intermediate frequency signal, so that the presettable counter 35 has a correct phase.
The nd local signal can be output.

FIG. 6 explains the operation of phase measurement in the 2nd local portion. Now, when there is no phase shift of the 2nd local signal, the signal point A of the transmitted modulated wave
When the count value a obtained by measuring the signal with an accuracy of 360 ° / 2 ^k is known in advance, the count obtained by measuring the phase of the signal point B of the intermediate frequency signal with an accuracy of 360 ° / 2 ^k. By comparing the value b with the count value a and presetting the counter so that the count value b becomes equal to a, the phase of the 2nd local signal obtained by the counter is modified and the signal point B of the intermediate frequency signal is It comes to coincide with the signal point A of the transmitted modulated wave.

According to the embodiment shown in FIG. 3, since the limiter is used to input the intermediate frequency signal for generating the 2nd local signal to be supplied to the quadrature demodulator, the accuracy is improved even when the input signal is weak. A high phase measurement can be made, and thus the distortion at the transmit power can be correctly measured.

Further, since the phase measurement is performed by digital processing and counting the master clock, the measurement error is determined by ± 2π × intermediate frequency / master clock frequency, and no adjustment error occurs. (If the phase is detected from the demodulation baseband output, the DC drift of the demodulator and the adjustment accuracy at the input of the analog-digital converter have a great influence, which is not preferable.)

Since the phase of the local signal of the quadrature demodulator can be controlled by the presettable counter, the infinite phase shifter (E
This can be realized with a simple circuit as compared with the case of using PS (Endless Phase Shifter) or the like.

Further, since the quadrature demodulator is operated in the intermediate frequency band, the circuit can be constructed at a lower cost than the conventional one and the gain can be increased in the intermediate frequency band. Will be alleviated.

Further, by monitoring the signal before passing through the power amplifier, it is possible to detect the deviation of the local signal phase in the state where the Cartesian loop is closed, and the optimum control of the local phase becomes possible.

By varying the bandpass filter characteristic for the local signal during phase control in the closed loop state of the Cartesian loop, it is possible to achieve both transient characteristics for burst transmission and phase tracking during continuous transmission.

FIG. 7 shows another embodiment of the present invention, in which the same components as those in FIG. 10 are designated by the same reference numerals, 45 is an infinite phase shifter (EPS), and 46 is a fixed phase shifter. (Φ), 47 is a mixer, and 48 is a maximum value detection circuit.

In the quadrature modulator 11, the local signal of the local oscillator 12 is quadrature-modulated by the input baseband signals I and Q to generate a modulated wave signal such as a π / 4 shift QPSK modulated signal. This modulated wave signal is amplified and a transmission output is generated.

A part of the transmission output is extracted by the directional coupler 14, and the quadrature demodulator 15 uses the local signal obtained by delaying the output of the local oscillator 12 through the infinite phase shifter 45 and the fixed phase shifter 46. Quadrature detection is performed to obtain a detection output including an in-phase component and a quadrature component.

The subtractor 17 obtains the difference between the in-phase component from the quadrature demodulator 15 and the in-phase component I of the input baseband signal to obtain the in-phase component of distortion. Similarly, subtracter 1
8, the difference between the quadrature component from the quadrature demodulator 15 and the quadrature component Q of the input baseband signal is obtained to obtain the quadrature component of distortion.

The feedback amplifiers 19 and 20 amplify the in-phase component and the quadrature component of the distortion, respectively, and multiply by K, and the adders 21 and 22 add them to the in-phase component I and the quadrature component Q of the input baseband signal, respectively. By doing so, negative feedback is performed for the in-phase component and the quadrature component, respectively, and both amplitude distortion and phase distortion are compensated.

A part of the transmission output extracted through the directional coupler 14 is multiplied by the output of the infinite phase shifter 45 in the mixer 47. The maximum value detection circuit 48 includes a mixer 47.
When the output of is increased or decreased from that before, the amount of phase shift of the infinite phase shifter 45 is increased or decreased by the unit phase change amount Δφ, and further the increase or decrease of the output of the mixer 47 is judged, and the phase shift of ± Δφ is performed. By performing feedback control so that the amount change is repeated, the amount of phase shift of the infinite phase shifter 45 is controlled to the phase where the output of the mixer becomes maximum.

The fixed phase shifter 46 has a fixed amount of phase shift, and until the maximum value detection circuit 48 operates and the infinite phase shifter 45 reaches the maximum amount of phase shift, the local phase shifter 46 operates locally. It gives a fixed amount of phase shift to the signal.

When the maximum value detection circuit has reached a steady state, the output phase of the infinite phase shifter 45 matches the vector of the transmission signal point. Since the reference vector of the quadrature modulator can be obtained by inserting the fixed phase shifter 46 into the output of the infinite phase shifter 45 and subtracting the known signal point phase, the output of the fixed phase shifter 46 is orthogonalized. By supplying as a local signal of the demodulator 15, a correct demodulation result can be obtained.

In this case, the single phase change amount Δφ of the maximum value detection circuit 48 determines the phase jitter in the steady state and the pull-in time until the pull-in to the optimum value. It is desirable that both phase jitter and pull-in time are small,
For these two quantities, if one becomes smaller, the other becomes larger,
Since there is a contradictory relationship, both can be satisfied by increasing the phase change amount Δφ at the start of pulling in and gradually decreasing it as it approaches a steady state.

FIG. 8 shows an example of the configuration of the maximum value detection circuit. Reference numeral 51 is a sample hold circuit for sampling and holding the input from the mixer 47 in FIG. 7 according to a clock, and 52 is a sample. A comparator for comparing the output and the input of the hold circuit 51, 53 is a flip-flop, and 54 is an up / down counter.

The sample-hold circuit 51 holds the mixer output level one clock before, and the comparator 52 compares this with the current mixer output level. When the previous sample value is larger, the flip-flop 53 is inverted by the output of the comparator 51, but when the current value is larger, the state of the flip-flop 53 does not change.

The up / down counter 54 is controlled by the output of the flip-flop 53 to count up and down, and counts the same clock as the sample hold circuit 51. The up / down counter 54 is
If the mixer output is larger than before, it is counted in the same direction, and if the mixer output is smaller, it is counted in the opposite direction.

The output value of the up / down counter 54 is given to the infinite phase shifter 45 in FIG. 7 as a control signal to control the amount of phase shift. Therefore, the operation of the circuit of FIG.
Has its phase shift amount set such that the output of the mixer 47 is maximized.

FIG. 9 shows another configuration example of the maximum value detection circuit. The same components as those in FIG. 8 are shown by the same numbers, 55 is a switch box, and 56 is a frequency divider for dividing a clock. It is a vessel.

The clock is divided by the divider 56, and the opening and closing of the switch box 55 is controlled by the output of the divider 56. The switch box 55 thereby shifts the bits output from the up / down counter 54 to the infinite phase shifter 45 shown in FIG.

The frequency divider 56 is cleared before the phase measurement of the maximum value detection circuit is started, and the switch box is moved once every n clock time. Then, for example, assuming that the minimum phase change amount of the infinite phase shifter 45 is Δφ, the amount of phase correction once is ±
8Δφ → ± 4Δφ → ± 2Δφ → ± Δφ, which is the minimum value ±, which is changed by the output of the frequency divider 56.
When Δφ is reached, the operation of the frequency divider 56 is stopped.

According to the embodiment shown in FIG. 7, the phase of the demodulator local signal can be converged to the optimum value.
In addition, the analog-to-digital converter, ta, which is necessary for the conventional method of obtaining the phase shift from the baseband demodulated output signal,
Since the n ⁻¹ (Q / I) computing unit and the like are unnecessary, the circuit scale can be reduced.

[0082]

As described above, according to the present invention, in a power amplifier for performing Cartesian distortion compensation,
The local signal phase of the quadrature demodulator that demodulates the modulated wave signal can be controlled to an optimum state, and therefore the distortion component can be accurately fed back and the distortion compensation can be correctly performed.

[Brief description of drawings]

FIG. 1 is a diagram showing a principle configuration (1) of the present invention.

FIG. 2 is a diagram showing a basic configuration (2) of the present invention.

FIG. 3 is a diagram showing an embodiment of the present invention.

FIG. 4 is a diagram showing an image rejection mixer.

FIG. 5 is a diagram showing a configuration example of a 2nd local part.

FIG. 6 is a diagram illustrating an operation of phase measurement in a 2nd local part.

FIG. 7 is a diagram showing another embodiment of the present invention.

FIG. 8 is a diagram showing a configuration example of a maximum value detection circuit.

FIG. 9 is a diagram showing another configuration example of the maximum value detection circuit.

FIG. 10 is a diagram showing a conventional Cartesian power amplifier.

[Explanation of symbols]

 1 quadrature modulator 2 quadrature demodulator 3 power amplifier 4 frequency conversion means 5 second local signal generation means 6 variable phase shifter 7 fixed phase shifter 26 synthesizer 27 frequency divider 33 variable band type band pass filter 34 changeover switch 45 infinity Phase shifter 47 Mixer 48 Maximum value detection circuit

Claims (8)

[Claims] 1. A quadrature modulator (1) for quadrature-modulating a transmission local signal synchronized with a reference signal with an input baseband signal expressed in quadrature coordinates to create a modulation wave signal, and amplifying the modulation wave signal. In the power amplifier (3) for generating a transmission output and demodulating the transmission output through the quadrature demodulator (2) to the input baseband signal, the transmission output frequency is the same as the transmission local signal. A frequency of 1 / n (n is a positive integer) of the reference signal is converted by frequency conversion by the first received local signal having a frequency difference from a signal obtained by dividing the reference signal by n (n is a positive integer). A frequency conversion means (4) for creating an intermediate frequency signal having the phase, and a phase obtained by counting the phase difference between the phase of the intermediate frequency signal and the input baseband signal by counting the reference signal as a master clock. A second reception local signal generating means (5) for generating a second reception local signal having a frequency of 1 / n of the reference signal having a predetermined phase difference from the phase of the input baseband signal, A power amplifier characterized in that demodulation is performed in the quadrature demodulator (2) by a second received local signal. 2. The first received local signal is a frequency divider (27) that divides the reference signal by n and a signal obtained by phase-shifting the n divided signal and the signal by 90 degrees. Image rejection for generating the first received local signal having a frequency that is the difference between the transmitted local frequency and the frequency-divided n signal by multiplying and adding the signal and the signal obtained by phase-shifting the signal by 90 °. Power amplifier according to claim 1, characterized in that it is made by means of a mixer (28). 3. Power amplifier according to claim 1 or 2, characterized in that the transmitted local signal is produced by a synthesizer (26) operating in synchronization with the reference signal. 4. The power amplifier according to claim 1, further comprising a changeover switch (34) provided at an input of the frequency conversion means (4) so that the input side of the power amplifier (3) is closed in a closed loop state. A power amplifier which is inputted to the frequency conversion means (4). 5. The power amplifier according to claim 1, wherein a variable band type band pass filter (33) is provided at an output side of the second reception local signal generating means (5).
The power amplifier is characterized by providing a wide band at the time of burst rise and narrowing the band thereafter. 6. A quadrature modulator (1) for quadrature-modulating a local signal with an input baseband signal expressed in quadrature coordinates to create a modulated wave signal, and amplifying the modulated wave signal to generate a transmission output. In addition, in the power amplifier (3) for feeding back the signal obtained by demodulating the transmission output through the quadrature modulator (2) to the input baseband signal, the local signal is phase-shifted to have the same phase as the transmission output. And a fixed phase shifter (7) for shifting the output of the variable phase shifter (6) by a fixed amount of phase, the quadrature demodulator (2) A power amplifier characterized in that the output of a fixed phase shifter (7) is demodulated as a local signal. 7. The variable phase shifter (6) includes an infinite phase shifter (45) for shifting the local signal, and a mixer (47) for multiplying the output of the infinite phase shifter by the transmission output. A maximum value detection circuit (48) for detecting the maximum value of the mixer output, and controlling the phase shift amount of the infinite phase shifter (45) by the output of the maximum value detection circuit (48). The power amplifier according to claim 6, wherein the power amplifier is a power amplifier. 8. The maximum value detection circuit (48) generates a phase correction signal for the infinite phase shifter (45) at regular operation cycles, and the phase correction amount in the phase correction signal is set at the start of phase control. 8. The power amplifier according to claim 7, wherein the power amplifier is controlled so as to be larger and then gradually smaller.