JPH05153182A - Digitized orthogonal modulator - Google Patents

Abstract

PURPOSE:To dispense with the filter of a steep characteristic and adjusting work by D/A-converting the outputs of a first and a second digitized orthogonal modulators to modulate information signals I, Q, and after that, multiplying them with orthogonal local oscillator outputs, and summing and orthogonal- modulating these outputs. CONSTITUTION:The digitized orthogonal modulators A, B input the input information signals I, Q and the carrier waves omega1 orthogonal to each other to multipliers 1, 2, and input the information signals I, Q and the carrier waves whose phases are leading or lagging before/behind the carrier waves omega1 by pi/2 to the multipliers 11, 12, and multiply them respectively. The outputs S1, S2 obtained by summing these multiplied results by adders 4,14 are turned into analog signals by D/A converters 5, 15, and are inputted to mixers 20, 21 through BPFs 8, 18. The mixers 20, 21 multiply the outputs obtained by making the output of a local oscillator 26 orthogonal to each other by a phase shifter 19 together with the input signals, and the mixer 22 mixes and outputs these multiplied results. Thus, the filter of the steep characteristic and the adjusting work can be dispensed with.

Description

Detailed Description of the Invention

[0001]

FIELD OF THE INVENTION This invention relates to digitized quadrature modulators.

[0002]

2. Description of the Related Art A conventional analog modulator has information signals I and Q.
In the baseband signal of, since the information is from the DC component, the multiplication part is also DC coupled, temperature drift becomes a problem, a temperature compensating circuit is required, and a lot of adjustment is required. Therefore, a digitalized quadrature modulator has been proposed.

A conventional digital quadrature modulator has an information signal I and a reference carrier wave (cos ω ₁ t) as schematically shown in FIG.
Digital multiplier 1 for multiplying and
A digital multiplier 2 that multiplies a quadrature carrier (sin ω ₁ t) with a phase delay of ½ and an information signal Q, and a digital adder 4 that adds the output of the digital multiplier 1 and the output of the digital multiplier 2. There is. By storing the reference carrier wave and the quadrature carrier wave as data in the ROM in advance, a digital quadrature modulator is obtained. The frequency spectrum of the output of the digital adder 4 is as shown in FIG.

Further, the digitizing quadrature modulator is designed to extract an IF signal or an RF signal having a desired frequency.
The output of the digital adder 4 is converted into an analog signal by the D / A converter 5, the high frequency component is removed and taken out through the low pass filter 6, and the oscillation output (frequency ω ₂ ) from the local oscillator 26 and the low pass filter 6 Is multiplied by the mixer 7 and output. In this case, the frequency spectrum of the output signal of the mixer 7 is as shown in FIG.
There are two waves having center frequencies of (ω ₂ −ω ₁ ) and (ω ₂ + ω ₁ ).

[0005]

However, the above-mentioned conventional digitized modulator has the multiplication speed of the multiplier and the addition speed of the adder, the access time when the ROM is used, and the D / A converter set. Due to problems such as ring time, the carrier frequency is limited.

Another problem is the storage capacity of the ROM, and the carrier frequency and the information rate (signal band) are close to each other. In this case, if the frequency is converted into an IF signal or an RF signal having a desired frequency, two waves will appear in the vicinity of the local oscillation frequency ω ₂ . If this is expressed by an equation, the output S ₀ is as in the following equation (1).

[0007]

[Expression 1] Output S ₀ = I cosω ₁ t cosω ₂ t + Q sinω ₁ t cosω ₂ t = I {cos (ω ₂ −ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 + Q {sin (ω ₁ − ω ₂ ) t + sin (ω ₂ + ω ₁ ) t} / 2 = {I cos (ω ₂ −ω ₁ ) t−Q sin (ω ₂ −ω ₁ ) t} / 2 + {I cos (ω ₂ + ω ₁ ) t + Qsin (ω ₂ + ω ₁ ) t} / 2 ……… (1)

In order to separate two waves appearing in the vicinity of the local oscillation frequency ω _2, a bandpass filter is required to have a steep characteristic, and it is impossible to obtain a filter having a steep characteristic and a good group delay characteristic. There was a problem that it was difficult.

According to the present invention, an IF signal of a desired frequency can be obtained without using a filter having a sharp characteristic and a good group delay characteristic.
An object is to provide a digitized quadrature modulator that can be taken out as an RF signal.

[0010]

A digitized quadrature modulator of the present invention is a first digitization for multiplying an information signal I and an information signal Q by mutually orthogonal carriers and adding the multiplication results. The quadrature modulator and the phase of the carrier orthogonal to the information signal I and the information signal Q are further π /
Multiply the binary or delayed carrier respectively,
A second digitizing quadrature modulator for adding the multiplication results and first and second outputs for individually converting the outputs of the first and second digitizing quadrature modulators into analog signals.
Of the first and second band-pass filters, and the first and second band-pass filters that respectively input the converted outputs of the first and second D / A converters. It is characterized in that it is provided with first and second multiplying means for multiplying locally oscillated outputs which are orthogonal to each other respectively to the outputs, and adding means for adding the outputs of the first and second multiplying means.

[0011]

According to the digitized quadrature modulator of the present invention, the information signals I and Q are digitized quadrature modulated in the first and second digitized quadrature modulations respectively, and the first and second digitized quadrature modulations are performed. The outputs of the converters are converted into analog signals, which become orthogonal modulation signals, which are multiplied by the orthogonal local oscillation outputs, and the multiplication outputs are added, so that the addition means outputs the orthogonally modulated modulation signals. In this case, no information is added to the DC component at the outputs of the first and second digitizing quadrature modulators. Therefore, the latter part of the analog circuit can be AC coupled,
Eliminates the effects of temperature drift and eliminates the need for adjustment. Further, the quadrature-modulated modulated signal converted to a desired IF or RF frequency and output from the adding means has a center frequency which is either the sum or the difference between the carrier frequency and the local oscillation output frequency. Since it is only waves, there is no need for a filter with steep characteristics.

[0012]

EXAMPLES The present invention will be described below with reference to examples. FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention.

A digital multiplier 1 for multiplying the information signal I by a reference carrier (cosω ₁ t), π / 2 with respect to the reference carrier
A digital quadrature modulator including a digital multiplier 2 that multiplies a phase-delayed quadrature carrier (sin ω ₁ t) and an information signal Q, and a digital adder 4 that adds the output of the digital multiplier 1 and the output of the digital multiplier 2. Configure A. The digital quadrature modulator A is obtained by storing the reference carrier wave and the quadrature carrier wave as data in the ROM in advance. The frequency spectrum of the output of the digital adder 4 is shown in FIG.
It is as shown in (a).

A digital multiplier 11 for multiplying the information signal I by the quadrature carrier (sin ω ₁ t), and the quadrature carrier by π /
Digitization orthogonal by a digital multiplier 12 that multiplies a carrier wave (−cosω ₁ t) with a two-phase delay and an information signal Q, and a digital adder 14 that adds the output of the digital multiplier 11 and the output of the digital multiplier 12 The modulator B is configured. The digitized quadrature modulator B is obtained by storing the carrier wave (-cosω ₁ t) in the ROM in advance. The frequency spectrum of the output of the digital quadrature modulator B is also shown in FIG.
It is as shown in (a).

Further, an IF signal or R having a desired frequency
In order to extract as an F signal, the output of the digital adder 4 is converted into an analog signal by the D / A converter 5, the component of a predetermined frequency band is extracted through the bandpass filter 8, and the output of the digital adder 14 is output. The D / A converter 15 converts the signal into an analog signal and the bandpass filter 18 extracts a component in a predetermined frequency band. The output of the bandpass filter 8 and the output of the bandpass filter 18 are analog orthogonal modulation signals.

Furthermore, the output of the digitized quadrature modulator A that has passed through the band pass filter 8 is supplied to the mixer 20, and the oscillation output (frequency ω ₂ ) from the local oscillator 26 is supplied to the mixer π / 2 phase. Delayed sine wave (sinω
₂ t). The output of the digitized quadrature modulator B that has passed through the bandpass filter 18 is supplied to the mixer 21 and is multiplied by the local oscillation output (cosω ₂ t) of the frequency ω ₂ . Mixer 2
The output of 0 and the output of the mixer 21 are mixed by the mixer 22 and output. The frequency spectrum of the output signal of the mixer 22 is as shown in FIG. 2B with the center frequency of (ω ₂ + ω ₁ ).

In one embodiment constructed as described above, the output S ₁ of the digitized quadrature modulator A is as shown in equation (2).

[0018]

[Equation 2] _{S 1 = I cosω 1 t +} Q sinω 1 t ...... (2)

The output S ₂ of the digitized quadrature modulator B is as shown in equation (3).

[0020]

[Equation 3] _{S 2 = I sinω 1 t-} Q cosω 1 t ...... (3)

The outputs S ₁ and S ₂ are individually converted into analog signals by the D / A converters 5 and 15, respectively. As shown in FIG. 2 (a), the outputs S ₁ and S ₂ do not carry information on the DC component, so that the D / A converters 5 and 1 are provided.
From the 5th to the latter part, the analog part is improved by AC coupling.

[0022] Each D / A converted output S ₂ passing through the output S ₁ and D / A converted band-pass filter 18 through the band-pass filter 8 mixers 20 and 21
Is multiplied by the orthogonal local oscillator output of frequency ω ₂ . The local oscillator output to the mixer 20 and sin .omega ₂ t, when the local oscillator output to the mixer 21 and cos .omega ₂ t, a result of multiplication output is mixed, the mixed output S is expressed by equation (4).

[0023]

[Equation 4] Output _{S = S 1 sinω 2 t +} S 2 cosω 2 t = (I cosω 1 t + Q sinω 1 t) sinω 2 t + (I sinω 1 t-Q cosω 1 t) cosω 2 t = I {sin (ω 2 -ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ -ω ₁ ) t−cos (ω ₂ + ω ₁ ) t} / 2 + I {-sin (ω ₂ -ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 −Q {cos (ω ₂ −ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 = I sin (ω ₂ + ω ₁ ) t−Q cos (ω ₂ + ω ₁ ) t ……… (4)

As is clear from the equation (4), the I signal component leads the Q signal component by π / 2 phase. In other words, the output S is

[0025]

[Expression 5] Output S = I cos (ω ₂ + ω ₁ ) t + Q sin (ω ₂ + ω ₁ ) t (5)

The next, the frequency spectrum of the output signal will be only appears modulated signal (ω ₂ + ω ₁₎ A is as shown in FIG. 2 (b) having a center frequency of a frequency (ω ₂ + ω ₁₎ ..

Further, the reference carrier and sin .omega ₁ t, when the orthogonal carrier was cos .omega ₁ t output S ₁ and S ₂ becomes Equation (6) and (7).

[Formula 6] _{S 1 = I sinω 1 t +} Q cosω 1 t ...... (6)

[Equation 7] _{S 2 = -I cosω 1 t +} Q sinω 1 t ...... (7) , where the local oscillator output to the mixer 20 cos .omega ₂ t, when the local oscillator output to the mixer 21 and sin .omega ₂ t, the output S is expressed by equation (8).

[0028]

[Equation 8] output _{S = S 1 cosω 2 t +} S 2 sinω 2 t = (I sinω 1 t + Q cosω 1 t) cosω 2 t + (- I cosω 1 t + Q sinω 1 t) sinω 2 t = I {-sin (ω 2 -ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ -ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 −I {sin (ω ₂ -ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ −ω ₁ ) t−cos (ω ₂ + ω ₁ ) t} / 2 = −I sin (ω ₂ −ω ₁ ) t + Q cos (ω ₂ -ω ₁ ) t ……… (8)

Here, as in the case of equation (5), equation (8)
When S is transformed, the output S becomes as shown in Expression (9).

[0030]

[Expression 9] Output S = I cos (ω ₂ −ω ₁ ) t + Q sin (ω ₂ −ω ₁ ) t ………… (9)

The next, the frequency spectrum of the output signal is a spectrum having a center frequency (ω ₂ -ω _1), and only the modulating signal of a frequency (ω ₂ -ω ₁₎ appears.

Further, the quadrature modulator of the above-mentioned embodiment is
It can be applied to all quadrature modulators such as QPSK, QAM and MSK. Furthermore, a waveform shaping filter that adds a cosine roll-off characteristic can be inserted before the quadrature modulator for digital processing.

[0033]

As described above, according to the present invention, the information signals I and Q are digitized and quadrature-modulated in the first and second digitized quadrature modulations, respectively, and the first and second digitized quadrature modulators are provided. Since the output of is converted to an analog output is multiplied by the orthogonal local oscillation output and the multiplication outputs are added to perform quadrature modulation, the addition means outputs a quadrature-modulated modulation signal, and the first and second In the output of the digital quadrature modulator, the DC component does not carry information, and the analog circuit portion in the latter stage is AC coupled, which is effective in avoiding the influence of temperature drift and requiring no adjustment.

Furthermore, the IF signal or R of the desired frequency
The quadrature-modulated modulation signal converted into the F signal and output from the adding means is only one wave whose center frequency is either the sum or the difference between the carrier frequency and the local oscillation output frequency. Since the part is removed, there is an effect that a filter having a steep characteristic is not needed.

As a result, the carrier can be set to a frequency close to the information rate (signal band). Therefore, the multiplication speed, the addition speed, the storage capacity of the ROM and the access time of the ROM can be afforded, and an inexpensive device can be used.

Further, since there is a margin in the D / A converter, there is an effect that a D / A converter having a sufficiently large number of quantization bits can be used.

[Brief description of drawings]

FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention.

FIG. 2 is a schematic diagram showing a frequency spectrum used for explaining the operation of one embodiment of the present invention.

FIG. 3 is a block diagram showing a configuration of a conventional example.

FIG. 4 is a schematic diagram showing a frequency spectrum for explaining the operation of the conventional example.

[Explanation of symbols]

 A and B Digital quadrature modulator 1, 2, 11 and 12 Digital multiplier 4 and 14 Digital adder 5 and 15 D / A converter 8 and 18 Bandpass filter 19 Phase shifter 20 and 21 Mixer 22 Mixer 26 Local oscillator

[Procedure amendment]

[Submission date] April 6, 1992

[Procedure Amendment 1]

[Document name to be amended] Statement

[Correction target item name] 0007

[Correction method] Change

[Correction content]

Output S ₀ = I cosω ₁ t cosω ₂ t + Q sinω ₁ t cosω ₂ t = I {cos (ω ₂ −ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 + Q {sin (ω ₁ −ω ₂ ) t + sin (ω ₂ + ω ₁ ) t} / 2 = {I cos (ω ₂ −ω ₁ ) t−Q sin (ω ₂ −ω ₁ ) t} / 2 + {I cos (ω ₂ + ω ₁ ) t + Qsin (ω ₂ + ω ₁ ) t} / 2 ……… (1)

[Procedure Amendment 2]

[Document name to be amended] Statement

[Correction target item name] 0018

[Correction method] Change

[Correction content]

[0018] _{S 1 = I cosω 1 t +} Q sinω 1 t ...... (2)

[Procedure 3]

[Document name to be amended] Statement

[Correction target item name] 0020

[Correction method] Change

[Correction content]

[0020] _{S 2 = I sinω 1 t-} Q cosω 1 t ...... (3)

[Procedure amendment 4]

[Document name to be amended] Statement

[Name of item to be corrected] 0023

[Correction method] Change

[Correction content]

The output _{S = S 1 sinω 2 t +} S 2 cosω 2 t = (I cosω 1 t + Q sinω 1 t) sinω 2 t + (I sinω 1 t-Q cosω 1 t) cosω 2 t = I {sin (ω 2 - ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ -ω ₁ ) t−cos (ω ₂ + ω ₁ ) t} / 2 + I {-sin (ω ₂ -ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2−Q {cos (ω ₂ −ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 = I sin (ω ₂ + ω ₁ ) t−Q cos ( ω ₂ + ω ₁ ) t ……… (4)

[Procedure Amendment 5]

[Document name to be amended] Statement

[Name of item to be corrected] 0025

[Correction method] Change

[Correction content]

Output S = I cos (ω ₂ + ω ₁ ) t + Q sin (ω ₂ + ω ₁ ) t (5)

[Procedure correction 6]

[Document name to be amended] Statement

[Name of item to be corrected] 0027

[Correction method] Change

[Correction content]

Further, the reference carrier and sin .omega ₁ t, when the orthogonal carrier was cos .omega ₁ t output S ₁ and S ₂ becomes Equation (6) and (7). _{S 1 = I sinω 1 t +} Q cosω 1 t ...... (6) S 2 = -I cosω 1 t + Q sinω 1 t ...... (7) , where the local oscillation output to the mixer 20 cos .omega ₂ t, to the mixer 21 When the local oscillation output is sinω ₂ t, the output S is expressed by equation (8).

[Procedure Amendment 7]

[Document name to be amended] Statement

[Correction target item name] 0028

[Correction method] Change

[Correction content]

The output _{S = S 1 cosω 2 t +} S 2 sinω 2 t = (I sinω 1 t + Q cosω 1 t) cosω 2 t + (- I cosω 1 t + Q sinω 1 t) sinω 2 t = I {-sin (ω 2 - ω ₁ ) t + sin (ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ −ω ₁ ) t + cos (ω ₂ + ω ₁ ) t} / 2 −I {sin (ω ₂ −ω ₁ ) t + sin ( ω ₂ + ω ₁ ) t} / 2 + Q {cos (ω ₂ −ω ₁ ) t−cos (ω ₂ + ω ₁ ) t} / 2 = −I sin (ω ₂ −ω ₁ ) t + Q cos (ω ₂ -ω ₁ ) t ………… (8)

[Procedure Amendment 8]

[Document name to be amended] Statement

[Name of item to be corrected] 0030

[Correction method] Change

[Correction content]

Output S = I cos (ω ₂ −ω ₁ ) t + Q sin (ω ₂ −ω ₁ ) t (9)

Claims (1)

[Claims] 1. A first digitizing quadrature modulator for multiplying the information signal I and the information signal Q by mutually orthogonal carrier waves and adding the multiplication results, and the information signal I.
And a second digitizing quadrature modulator for multiplying the information signal Q by a carrier obtained by further π / 2 binary phase or lagging the phase of the orthogonal carrier and adding the multiplication result, and And the first and second D / A converters that respectively convert the outputs of the second digitized quadrature modulator into analog signals, and the converted outputs of the first and second D / A converters, respectively. First and second input separately
Band-pass filters, first and second multiplying means for respectively multiplying the outputs of the first and second band-pass filters by mutually orthogonal local oscillation outputs, and outputs of the first and second multiplying means. A quadrature digitizing modulator comprising: