JPH04269002A - Orthogonal detector - Google Patents

Abstract

PURPOSE:To realize the orthogonal detector with a small linear distortion by measuring an amplitude ratio and a correlation coefficient due to a distortion of an orthogonal detection signal and compensating the distortion based on the measured value. CONSTITUTION:A conventional orthogonal detector is provided with a measurement means 3 and a linear conversion means 4. A signal generating means 2 eliminates a DC component from an orthogonal detection output obtained by multiplying a reference signal and a modulation signal inputted from an input terminal 1 to generate an in-phase detection signal and an orthogonal detection signal. The measurement means 3 measures an amplitude ratio and a correlation ratio from the two signal components being the in-phase detection signal and the orthogonal detection signal. The linear conversion means 4 based on the said amplitude ratio and correlation ratio eliminate the correlation component from the orthogonal detection signal and makes the amplitude of the orthogonal detection signal with that of the in-phase detection signal to output the orthogonal detection signal subject to linear conversion.

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a quadrature detector, and more particularly to a quadrature detector for obtaining quadrature detection output signals with good orthogonality. 2. Description of the Related Art FIG. 4 shows a configuration diagram of a conventional quadrature detector. First, the conventional quadrature detector has an input terminal 41 and an output terminal 4.
2, 43, mixers 44, 45, low pass filter 46,
It is composed of a 47 and 90 degree phase shifter 48. Next, the operation of a conventional quadrature detector will be explained. A modulated signal S(t) is input from the input terminal 41. The modulated signal S(t) is S(t)=I(t)cos
ωc t −Q(t) sin ωc t
It is expressed as (1). Formula (1)
, I(t) and Q(t) on the right side are the amplitudes of the in-phase and quadrature detection components, respectively, and reflect the uncertain phase of modulation. This modulated signal S(t) is sent to two mixers 4
4, added to 45. These mixers 44 and 45 receive a reference signal C (
t) and its reference signal C(t) by 90° phase shifter 48.
A modulated signal S(t) phase-shifted by a degree is added. The reference signal C(t) is C(t)=Ac cos(ωc t)

(2), and the 90° phase shift modulation signal S
(t) is S(t)=-As sin(ωc t
−θ)
(3). Here, Ac and As in equations (2) and (3) are amplitudes, and θ is a deviation phase shift due to imperfection of the phase shifter 48. Low-pass filters (LPF1) 46 and (LPF2) 47 located after the mixers 44 and 45 in FIG. 4 are for removing high-frequency components from the product signals of the mixers 44 and 45. Outputs OUT1 and OUT of this configuration
2, the following outputs can be obtained from the output terminals 42 and 43. [0005] The output OUT1 is i(t)=Ac I(t)+δc

(4) Output OUT2 is q(t)=As [Q(t) cosθ−I
(t) sinθ]+δs
(5) However, the second terms δc and δs on the right side of equations (4) and (5) are DC offset components, and the mixers 44 and 45 and the low-pass filter 46,
This occurs due to the influence of the DC offset of 47. Originally, amplitude Ac = As, deviation phase shift θ = 0, D
Since the C offset component δc = δs = 0, the output OUT1 should be i(t) = Ac I(t) and the output OUT2 should be q(t) = Ac Q(t), but in general, equation (4 ), distortion components are added as shown in (5). [Problem to be Solved by the Invention] However, as shown in equations (4) and (5) above, if the output OUT1 is i(t)
= Ac I(t), output OUT2 is q(t) = Ac
Q(t) and is subject to distortion. FIG. 5 shows the state of linear distortion of a conventional quadrature detector. FIG. 5A shows a case where the amplitude, deviation phase shift, DC offset component, etc. are ideal. The ideal values are that the amplitude is A c = As, the DC offset component is δc = δs = 0, and the deviation phase shift is θ = 0. I(t
)=cos(ωo t), Q(t)=sin(ωo
When receiving the modulated wave of t), I(t) and Q(t)
It shows the locus of the circle drawn by. [0008] Figure (B) shows the case where there is a DC offset component, in which case the amplitude is Ac = As, DC
The offset component is δc≠0, δs≠0, and the deviation phase shift is θ=0. Therefore, the center of the circular locus is shifted by the DC offset components δc and δs. FIG. 2C shows a case where the amplitude is unbalanced. In this case, the amplitude is A c < As, the DC offset component is δc = δs = 0, and the deviation phase shift is θ = 0. Therefore, it becomes an elliptical locus extending in the Q direction. [0010] Figure (D) shows the case where the 90-degree phase shifter is incomplete, in which case the amplitude is Ac = As, the DC offset component is δc = δs = 0, and the deviation phase shift is θ>0.
It is. Therefore, the trajectory becomes a tilted ellipse. [0011] In this way, linear distortion greatly deteriorates detection characteristics, such as error rate characteristics in digital transmission, so manufacturing a quadrature detector requires careful selection of well-balanced components and fine tuning that requires skill. This method has disadvantages in that it requires adjustment work and expensive measuring instruments. The present invention has been made in view of the above points, and it is an object of the present invention to provide a quadrature detector with extremely small linear distortion without requiring various methods for preventing linear distortion when manufacturing the quadrature detector. purpose. [Means for Solving the Problems] FIG. 1 shows a diagram of the basic configuration of the present invention. A reference signal generating means inputs a modulated signal from the input terminal (1) and generates a reference signal, and removes a DC component from the quadrature detection output obtained by multiplying the reference signal and the modulation signal to generate an in-phase detection signal and a quadrature detection signal. a signal generating means (2) for generating a signal, a measuring means (3) for measuring an amplitude ratio and a correlation coefficient from two signal components, an in-phase detection signal and a quadrature detection signal; The linear conversion means (4) removes the correlation component from the orthogonal detection signal based on the correlation coefficient, and outputs the linearly converted orthogonal detection signal by matching the amplitude of the orthogonal detection signal with the in-phase detection signal. and configure. [Operation] The present invention includes a measurement means and a linear conversion means in the configuration of a conventional quadrature detector, and the measurement means eliminates distortion by using the amplitude ratio and correlation coefficient of the signal generated by the signal generation means. Measure. The linear conversion means compensates for distortion in the quadrature detection signal based on the measured value. This means that there is no correlation with each other,
Detection signals with equal levels can be obtained. [Embodiment] First, distortion compensation processing will be explained. First of all,
The equations (4) and (5) above can be transformed into the original signal I (
t) and Q(t), Ac I(t)=i(t)−δc

(6) Ac Q(
t)=[q(t)−δs]/A cos θ+[i(
t)−δc ]tan θ (7). However, A=As/Ac. That is, the orthogonal detection outputs i(t), q(t
) is processed by equations (6) and (7), signals A c times the original signals I(t) and Q(t) can be obtained. For this process, DC offset components δc, δs
, amplitude A, and deviation phase shift θ. These can be easily measured when the modulated wave satisfies the following conditions.
<I(t)>=0

(7a) <Q(t)>=0

(7b)
<I(t)Q(t)>=0

(7c) <I2(t)>
=〈Q2(t)〉=σ2
(7d) These mean that I(t) and Q(t) each have no DC offset component and are mutually uncorrelated, which is true under normal application conditions. At this time, the formula (4
), (5) from 〈i(t)〉=δc

(8a) <q(t)>=
δs
(8
b) Therefore, by averaging the orthogonal detection i(t) and q(t), the DC offset components δc and δs can be obtained. Therefore, these DC offset components are detected by quadrature detection i(t
) and q(t), i0(t) and q0
(t). i0(t) = i(t)−δc

(9a) q0(t
) = q(t)−δs

(9b) Here, when the root mean square of i0(t) and q0(t) with the DC offset component removed is measured, it is as follows. <i02(t)>= Ac 2 σ2

(10a) <q02(
t)〉= As 2 σ2

(10b) Therefore, A = [<i02(t)>/<q02(t)>
]1/2
The amplitude A is determined by (11). Next, i0(
When we measure the correlation coefficient ρ between t) and q0(t), we get
ρ=〈i0(t)q0(t)〉/[〈i02(t)〉〈
q02(t)〉]1/2 =-sin θ


(12), so θ=arcsin(-ρ)

The deviation phase shift θ is determined by (13). As explained above, the DC offset component δ
Since c, δs, amplitude A, and correlation coefficient ρ can be easily measured, the original signal I(
t), Q(t) can be obtained. FIG. 2 shows a configuration diagram of an embodiment of the present invention. In the figure, the same components as those in FIG. 4 are given the same reference numerals, and the explanation thereof will be omitted. In the figure, the offset measurement circuit 11 and the DC
The offset compensation circuit 13 includes low-pass filters 46 and 47.
The output from The amplitude ratio phase shift measuring circuit 12 is D
The output from the C offset compensation circuit 13 is supplied. The amplitude phase shift compensation circuit 14 has an output from the output signal q0(t) of the DC offset compensation circuit 13 and the amplitude ratio phase shift measurement circuit 12.
The measurement results from Outputs i(t) and q(t) obtained through the circuits up to the low-pass filters 46 and 47 are input to the offset measuring circuit 11. The offset measurement circuit 11 performs averaging processing using the above-mentioned equations (8a) and (8b),
DC offset components δc and δs are measured. Next, the DC offset components δc and δs and the outputs i(t) and q(t) from the low-pass filters 46 and 47 are
Inputted into the offset compensation circuit 13, the above equation (9a)
, (9b) is performed, and i0
(t) = Ac I(t) and q0(t) are obtained. Next, these i0(t) and q0(t) are input to the amplitude ratio phase shift measuring circuit 12. The amplitude ratio phase shift measurement circuit 12 calculates equations (10a) and (10b) for measuring the root mean square with the DC offset component removed, then calculates the amplitude A using equation (11), and then calculates the amplitude A by equation (11). 1
The deviation phase shift θ is determined by 2) and (13). The amplitude A and deviation phase shift θ obtained from these are i0(t)
and q0(t) are input to the amplitude phase shift compensation circuit 14. The amplitude phase shift compensation circuit 14 is based on the above equation (7a).
, (7b), (7c), and (7d), Ac q(t) in which the orthogonal component serves as distortion compensation is obtained. Also, since nothing is done about the in-phase component, it remains AC I(t). In this way, detection signals having the same amplitude and excellent orthogonal detection properties can be obtained. An example in which the orthogonal detector having the above configuration is specifically applied will be described. FIG. 3 is a graph showing the results of applying the present invention to mobile communications using an adaptive equalizer. The same figure (A
) shows an example of DC component compensation, and graph 21
It can be seen that the case with compensation shown in graph 22 has a considerable effect compared to the state without compensation. Similarly, FIG. 3B shows an example in which amplitude compensation is performed. In addition, FIG. 6(C) is an example in which phase shift compensation is performed, and it can be seen that when compensation is performed, a considerable difference occurs compared to the state without compensation. As described above, according to the present invention, the present invention can be easily realized by converting the DC detection outputs i(t) and q(t) into digital signals using an A/D converter. Also,
CMOS-IC Since it can be easily realized using semiconductor technology, power consumption can be reduced, and it can be widely used in transmission equipment such as mobile phones.

[Brief explanation of the drawing]

FIG. 1 is a diagram showing the principle configuration of the present invention.

FIG. 2 is a configuration diagram of an embodiment of the present invention.

FIG. 3 is a graph showing the results of applying the present invention to mobile communications using an adaptive equalizer.

FIG. 4 is a configuration diagram of a conventional quadrature detector.

FIG. 5 is a diagram showing the state of linear distortion of a conventional quadrature detector.

[Explanation of symbols]

1 Input terminal 2 Signal generation means 3 Measurement means 4 Linear conversion means 11 Offset measurement circuit 12 Amplitude ratio phase shift measurement circuit 13 DC offset compensation circuit 14 Amplitude phase shift compensation circuit 41 Input terminal 42, 43 Output terminal 44, 45 Mixer 46, 47 Low pass filter

Claims (1)

[Claims] 1. A reference signal generating means which inputs a modulated signal from an input terminal and generates a reference signal, and generates an in-phase detected signal by removing a DC component from a quadrature detection output obtained by multiplying the reference signal and the modulated signal. signal generating means for generating a quadrature detection signal; measuring means for measuring an amplitude ratio and a correlation coefficient from two signal components of the in-phase detection signal and the quadrature detection signal; linear conversion means for removing a correlation component from the orthogonal detection signal based on a relational coefficient and outputting a linearly converted orthogonal detection signal by making the amplitude of the orthogonal detection signal coincide with the in-phase detection signal; A quadrature detector featuring: