JP3036093B2 - Quadrature detector - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a quadrature detector and, more particularly, to a quadrature detector for obtaining a quadrature detection output signal having good orthogonality.

[0002]

2. Description of the Related Art FIG. 4 shows the configuration of a conventional quadrature detector. First, a conventional quadrature detector has an input terminal 41, output terminals 42 and 43, mixers 44 and 45, a low-pass filter 4
6, 47, and 90-degree phase shifters 48.

Next, the operation of the conventional quadrature detector will be described. The modulated wave S (t) is input from the input terminal 41. Modulated wave S (t) can be expressed as S (t) = I (t ) cos ω c t -Q (t) sin ω c t (1). In equation (1), I (t) and Q (t) on the right side
Are the amplitudes of the in-phase and quadrature components , respectively, and also reflect the uncertain phase of the modulation. This modulated wave S (t) is applied to two mixers 44 and 45. These mixers 44,
45 the reference signal C gamma (t) and the reference signal C gamma a (t) 90 ° phase shifter 4 that is outputted from the carrier oscillator OSC 49 in
8, a reference signal Sγ (t ) shifted by 90 degrees is added. Reference signal C gamma (t) is _{Cγ (t) = A c cos} (ω c t) (2), 90 ° phase-shifted reference signal S [gamma] (t) is _{Sγ (t) = - A s} sin (ω c t + Θ) (3). Here, A _c and A _s of formula (2) and (3) the amplitude,
θ is a phase shift deviation due to imperfection of the phase shifter 48.

The low-pass filters (LPF1) 46 and (LPF2) 47 at the subsequent stage of the mixers 44 and 45 in FIG.
This is for removing high frequency components from the product signal of 5. Outputs OUT1 and OUT2 having such a configuration are connected to output terminal 42,
43, the following output is obtained.

The output OUT1 is i (t) = A _c I (t) + δ _c (4) The output OUT2 is q (t) = A _s [Q (t) cos θ−I (t) sin θ] + δ _s (5) However, δ _c and δ _s of the second term on the right side of the equations (4) and (5) are DC offset components, which are generated due to the DC offset of the mixers 44 and 45 and the low-pass filters 46 and 47. Is what you do. Ideally , the amplitude A _c = A _s , the phase shift deviation θ = 0, and the DC offset component δ _c = δ _s = 0,
The output OUT1 is i (t) = A _c I (t). The output OUT2 should be q (t) = A _c Q (t). Generally, as shown in equations (4) and (5), A distortion component is added.

[0006]

SUMMARY OF THE INVENTION However, the above equation
As shown in (4) and (5), output OUT1 is i (t) = A _c I (t)
OUT2 does not satisfy q (t) = A _c Q (t) and is distorted.

FIG. 5 shows a state of linear distortion of a conventional quadrature detector. FIG. 6A shows a case where the amplitude, the phase shift deviation , the DC offset component, and the like are ideal. The ideal value of this amplitude is A _c = A _s, DC offset component δ _c = δ _s
= 0, and the phase shift deviation is θ = 0. I (t) = cos (ω _o t), Q
When (t) = sin (ω _ot ) modulated wave is received, I (t) and Q
The trajectory of the circle drawn by (t) is shown.

[0008] FIG. (B) shows a case where there is a DC offset component, In this case, the amplitude A _c = A _s, the DC offset component _{δ c ≠ 0, δ s ≠} 0, the phase shift deviation theta = 0. Accordingly, the center of the circular locus is shifted by DC offset component [delta] _c, [delta] _s.

[0009] FIG. (C) shows a case amplitude imbalance, In this case, the amplitude A _c <A _s, DC offset component δ _c = δ _s = 0, the phase shift deviation in theta = 0 is there. Therefore,
It becomes an elliptical locus extending in the q direction.

[0010] FIG. (D) shows a case is incomplete 90-degree phase shifter, In this case, the amplitude A _c = A _s, DC offset component δ _c = δ _s = 0, the phase shift deviation θ> 0. Therefore, it becomes an inclined elliptical locus.

As described above, detection characteristics such as error rate characteristics in digital transmission are greatly degraded when linear distortion is present. Therefore, in the manufacture of a quadrature detector, it is necessary to select a well-balanced component and to obtain skill. There were drawbacks that required adjustment work and expensive measuring instruments.

The present invention has been made in view of the above points, and an object of the present invention is to provide a quadrature detector having extremely small linear distortion without the need for various methods for preventing linear distortion in manufacturing the quadrature detector. Aim.

[0013]

FIG. 1 is a block diagram showing the principle of the present invention. A reference signal generating means for receiving a modulation signal from an input terminal (1) and generating a reference signal; removing a DC component from a quadrature detection output obtained by multiplying the reference signal and the modulation signal to remove an in-phase detection signal and a quadrature detection signal amplitude and signal generating means for generating (2), the amplitude ratio of the two signals <br/> in-phase detection signal and the quadrature detection signal and measuring means for measuring a correlation coefficient (3), by measuring means (3) Linear conversion means for removing a correlation component from the quadrature detection signal based on the ratio and the correlation coefficient, and outputting a quadrature detection signal linearly converted by matching the amplitude of the quadrature detection signal with the in-phase detection signal (4) And comprising.

[0014]

According to the present invention, a conventional quadrature detector is provided with a measuring means and a linear converting means in the configuration of the conventional quadrature detector, and the measuring means measures an amplitude ratio and a correlation coefficient due to distortion of a signal generated by the signal generating means.
I do . The linear conversion means performs distortion compensation on the quadrature detection signal based on the measured value. This makes it possible to obtain detection signals having no correlation and equal levels.

[0015]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First, the distortion compensation processing will be described . The expression before mentioned (4) the original signal I (t) the equations (5) and, Solving for _{Q (t), A c I} (t) = i (t) -δ c (6a) A c Q (t) = [q (t) −δ _s ] / A cos θ + [i (t) −δ _c ] tan θ (6b) . However, it is A = A _s / A _c.

That is, the quadrature detection outputs i (t), q
By performing the processing of equations (6a) and (6b) on (t), the original signal I
(t), a signal of _Ac times Q (t) is obtained. For this processing, DC offset components δ _c , δ _s , amplitude A, phase shift deviation θ
You need to know. These can be easily measured when the modulated wave satisfies the following conditions.

I (t)> = 0 (7a) <Q (t)> = 0 (7b) <I (t) Q (t)> = 0 (7c) <I ² (t)> = <Q ² (t)〉 = σ ² (7d) These mean that I (t) and Q (t) have no DC offset components and are uncorrelated with each other,
This is true under normal application conditions. At this time, equation (4)
(5) from <i (t)> = δ c (8a) <q (t)> = because it is [delta] _s (8b), DC offset if the average quadrature detection output i and (t) q (t), component [delta] _c and [delta] _s is obtained. So these D
Those obtained by removing the C offset component from the quadrature detections i (t) and q (t) are defined as i ₀ (t) and q ₀ (t).

I ₀ (t) = i (t) −δ _c (9a) q ₀ (t) = q (t) −δ _s (9b) where i ₀ (t) and q from which the DC offset component has been removed _When the root mean square of ₀ (t) is measured, it is as follows.

[0019] ^{_{<i 0 2 (t)>}} = A c 2 σ 2 (10a) <q 0 2 (t)> = A s 2 σ 2 (10b) ^{_{Therefore, A = [<i 0 2}} (t)> / <Q ₀ ² (t)>] ^1/2 (11) gives the amplitude A.

Next, i ₀ (t) from which the DC offset component has been removed is
When the correlation coefficient ρ of q ₀ (t) is measured, ρ = <i ₀ (t) q ₀ (t)> / [<i ₀ ² (t)><q ₀ ² (t)> ^1/2 (12), the phase shift deviation θ is obtained from θ = arcsin (−ρ) (13).

As described above, the DC offset component δ
_{Since c,} δ _s , amplitude A, and correlation coefficient ρ can be easily measured, the original signals I (t) and Q (t) can be obtained from equations (6a) and (6b) .

FIG. 2 is a block diagram showing one embodiment of the present invention.
4, the same components as those of FIG. 4 are denoted by the same reference numerals, and the description thereof will be omitted.

[0023] In the figure, the offset measurement circuitry 11 and the DC offset compensation circuit 13 outputs from the low-pass filter 46, 47 is supplied. DC is the amplitude-correlation measurement circuit 12
An output from the offset compensation circuit 13 is supplied. amplitude
The phase shift compensating circuit 14 is supplied with the output from the output signal q ₀ (t) of the DC offset compensating circuit 13 and the measurement result from the amplitude / correlation measuring circuit 12.

The low-pass output i obtained through the circuit to filter 46,47 (t), q (t ) is input to the offset measurement circuitry 11. The offset measuring circuit 11 uses the above formula (8a) (8
Mean treatment with b) is carried out, DC offset component [delta] _c and [delta] _s is measured. Next, DC offset components δ _c and δ _s and outputs i (t) and q (t) from low-pass filters 46 and 47 are obtained.
Is input to the DC offset compensating circuit 13, and the above equation (9a),
The DC component is removed by (9b), and i ₀ (t) = A
_c I (t) and q ₀ (t) are obtained.

Next, these i ₀ (t) and q ₀ (t) are inputted to the amplitude / correlation measuring circuit 12. The amplitude / correlation measurement circuit 12 calculates an equation (10) for measuring a root mean square from which a DC offset component has been removed.
a) and (10b) are calculated, then the amplitude A is obtained by equation (11), and the phase shift deviation θ is obtained by equations (12) and (13). The amplitude A and the phase shift deviation θ obtained from these are i
₀ (t) and q ₀ (t) are input to the amplitude / phase shift compensation circuit 14.

The amplitude / phase shift compensating circuit 14 obtains A _c q (t) in which the quadrature component is distortion-compensated by the above equation (6b) . Also,
Since nothing is done about the phase component remains A _C I (t). In this way, a detection signal having the same amplitude and excellent quadrature detection can be obtained.

The quadrature detector having the above configuration is specifically described.
Here is an example of application . FIG. 3 is a graph showing a result of applying the present invention to mobile communication using an adaptive equalizer. FIG. 7A shows an example in which DC component compensation is performed, and is a graph.
It can be seen that a considerable effect is obtained in the case with the compensation shown in the graph 22 as compared with the state without the compensation in 31 .

FIG. 2B shows an example in which amplitude compensation is performed. FIG. 11C shows an example in which phase shift compensation is performed, and it can be seen that the compensation is considerably improved as compared with the characteristics without compensation.

[0029]

As described above, according to the present invention, if the DC detection outputs i (t) and q (t) are converted into digital signals by an A / D converter, it can be easily realized. In addition, since it can be easily realized by CMOS-IC semiconductor technology, it is possible to reduce power consumption, and it can be widely used for transmission equipment such as mobile phones.

[Brief description of the drawings]

FIG. 1 is a principle configuration diagram of the present invention.

FIG. 2 is a configuration diagram of one embodiment of the present invention.

3 is a graph showing the result of applying the present invention to a mobile communication using the adaptive equalizer.

FIG. 4 is a configuration diagram of a conventional quadrature detector.

FIG. 5 is a diagram showing a state of linear distortion of a conventional quadrature detector.

[Explanation of symbols]

Reference Signs List 1 input terminal 2 signal generation means 3 measurement means 4 linear conversion means 11 offset measurement circuit 12 amplitude / correlation measurement circuit 13 DC offset compensation circuit 14 amplitude / phase shift compensation circuit 41 input terminals 42, 43 output terminals 44, 45 mixer 46, 47 Low Pass Filter

────────────────────────────────────────────────── ─── Continuing from the front page Examiner Hiroshi Yoshioka (56) References JP-A-57-57007 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03D 3/02-3 / 06 H04L 27/14

Claims (1)

(57) [Claims] 1. A reference signal generating means for inputting a modulation signal from an input terminal and generating a reference signal, and removing a DC component from a quadrature detection output obtained by multiplying the reference signal and the modulation signal to remove an in-phase detection signal. the phase relationship between the signal generating means for generating a quadrature detection signal, measuring means for measuring a correlation coefficient between the amplitude ratio of the two signals of the same phase detection signal and the quadrature detection signal, and the amplitude ratio by the measuring means A linear conversion unit that outputs a quadrature detection signal that is linearly converted by removing a correlation component from the quadrature detection signal based on a number and matching the amplitude of the quadrature detection signal with the in-phase detection signal. Characteristic quadrature detector.