JPH08340362A - Modulation circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a quadrature modulation circuit with the small possibility of the degradation of output signals for being hardly influenced for the jump-in of the signals from the outside. SOLUTION: First and second oscillators 1 and 6 respectively output frequencies ωc1 and ωc2 largely separated from a carrier frequency. First and second 90-degree phase shift circuits 2 and 60 input the output signals of the two oscillators and respectively generate the signals provided with the phase difference of 90 degrees from each other. First and second single sideband SSB modulation circuits 10 and 20 input the output signals of the 90-degree phase shift circuits and output the two signals provided with the phase difference of 90 degrees from each other of the carrier frequency ωc1 +ωc2 or ωc1 -ωc2 (or ωc2 -ωc1 ). A mixer circuit 30 obtains the respective products of the two signals and base band signals I and Q, then, obtains the sum and obtains quadrature modulated signals.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a quadrature modulation circuit used in a digital mobile communication system or the like.

[0002]

2. Description of the Related Art The spread of mobile phones has been remarkable in recent years, and it is about to shift from an analog system to a digital system for performing quadrature modulation. Mobile phones are expected to become even more popular. Transmission and reception frequencies are PDC (Personal Digital Cellular), IS5
4 ("Cellular system dual mode mobile unit-base station compatible standard" adopted by North American digital cellular), GS
M (European unified standard) 800MHz-1GHz, PD
800 MHz for C and PHS (simple mobile phone system)
z to 1.9 GHz.

The transmitter of these types of mobile phones requires a modulator for performing four-phase phase modulation or frequency modulation at these frequencies. As a method of performing 4-phase modulation, the modulation method is shown in FIG. 3.6 on page 28 of RCR Standard-28 (developed on December 20, 1993). FIG. 8 shows the configuration of the quadrature modulator.

In FIG. 8, the frequency ωc1 from the oscillator 1
Is input to the 90-degree phase shifter 2 to obtain cos (ωc1t) and −sin (ωc1t). The outputs of the 90-degree phase shifter 2 are input to the mixers 3 and 4, respectively, and are multiplied by the baseband signals I (t) and Q (t). The outputs of the mixers 3 and 4 are input to the adder 5 to obtain the output S (t). This output S
(T) is expressed by the following equation. S (t) = I (t) · cos (ωc1t) −Q (t) · sin (ωc1t)

Here, ωc1 is the frequency of the carrier. What is important here is how accurately the functions of FIG. 8 can be integrated. In this circuit, since the frequency ωc1 of the carrier and the frequency of the output signal S (t) are the same, a part of the signal output from the antenna becomes a radio wave to supply the integrated circuit and the carrier that configure the transmission circuit. If the output signal is fed back to the 90 ° phase shifter 2 or the like through the oscillator circuit, the parasitic capacitance in the integrated circuit, or the substrate, the signal quality of the output signal S (t) is greatly deteriorated (carrier field through). , Remixing).

As a method for preventing this, a method using an offset mixer has been announced. this is,
The transmitter is described on pages 5 and 6 of "Wireless Symposium 94" conducted by Hulett Packard Company, and its configuration is shown in FIG. In FIG. 9, the frequencies of the two input oscillators 1 and 6 are separated from the transmitted carrier frequency by 100 MHz or more. Therefore, even if an unnecessary signal such as a radio wave from the antenna is mixed in the oscillators 1 and 6 in the portion before the frequency conversion within the dotted line, the component is largely separated from the carrier frequency (849 MHz). Since it is converted to a frequency (700 MHz, 998 MHz, etc.), it is in a frequency band different from that of the output signal, and is unlikely to cause deterioration of the quality of the output signal.

[0007]

[Problems to be Solved by the Invention] PDC, IS54, G
In SM, etc., extremely high modulation accuracy is required. As an example, it is required to be within 1.5 to 3 degrees including manufacturing variations. In the prior art of FIG. 9, this accuracy is determined by the 90-degree phase shifter 7, and it is extremely difficult to satisfy this requirement at 849 MHz. As a circuit for solving this problem and performing high-accuracy quadrature modulation, reference 1 "19
93 International Solid-State Circuit Conferenc
The 90-degree phase shifter (Fig. 1) described on pages 140 to 141 of e "is widely known. However, since the circuit is complicated and the number of elements is large, the output signal S
When (t) goes around the 90 ° phase shifter 7 of FIG. 9 through the base of the IC and the power supply line, the 90 ° phase shifter 7 operates at the same frequency as the output S (t). As in the case of 8, the signal quality of S (t) is deteriorated (remixing). In addition, three 90 ° phase shifters (7, 8, 9), which are complicated circuits, are required, which increases the number of elements and power consumption, which is uneconomical.

Not only when this circuit is used, but after being converted to the same frequency as that of the carrier, a circuit having a large number of elements having gain as much as possible is not allowed to pass therethrough to reduce a possibility that an output signal is fed back. The number of elements should be as small as possible and the circuit scale should be kept small. The 90 degree phase shifter is preferably operated at a frequency different from that of the carrier, if possible. Further, since it is very difficult to realize a 90 ° phase shifter with high accuracy, it is desirable to have a circuit configuration that corrects an error and has high accuracy without depending on the accuracy of the 90 ° phase shifter. Further, the above-mentioned circuit uses three 90-degree phase shifters, and therefore the circuit scale is large and uneconomical.

The present invention is not easily affected by a jump in of a signal from the outside, the possibility of deterioration of the output signal is small, and even if the accuracy of the 90 ° phase shifter is insufficient, it is corrected, An object of the present invention is to provide a quadrature modulation circuit having good phase accuracy.

[0010]

In the quadrature modulation circuit according to the present invention, the first and second oscillators respectively output frequencies ωc1 and ωc2 different from the carrier frequency. First
The second 90-degree phase shifter inputs the output signals of the two oscillators and generates signals having a phase difference of 90 degrees from each other. The first and second single sideband (SSB) modulation circuits receive the output signal of the 90-degree phase shifter and input the frequency ωc1 + ωc2 or ωc1−ωc2 (or ωc2-ωc1).
2 signals having a phase difference of 90 degrees from each other are output.
The mixer circuit takes the product of these two signals and the baseband signal, respectively, and then outputs the sum to obtain a quadrature-modulated signal. Since the SSB modulation circuit and the mixers 31 and 32 can be coupled in the shortest distance without a signal processing circuit such as a 90 ° phase shifter, a 90-degree phase shifter that processes a signal of the same frequency as the carrier is unnecessary. Since the number of circuits in which a signal wraps around decreases, the possibility of deterioration of an output signal due to signal wraparound decreases. Further, when the SSB modulation circuits 10 and 20 are used together, even if the 90-degree phase shifters 2 and 6 are not accurate, the error is canceled and the accuracy is improved.
A signal having a phase difference of 0 degree can be supplied to the mixers 31 and 32.

[0011]

DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described with reference to FIG. In FIG. 1, a single sideband (SSB) modulation circuit 10,
20 is a signal of the frequency ωc1 from the oscillator 1 passing through the 90-degree phase shifter 2 and the oscillator 6 passing through the 90-degree phase shifter 60.
And outputs a signal having a phase difference of 90 degrees, which is converted into a frequency corresponding to the sum or difference of the frequencies ωc1 and ωc2.

This output signal is quadrature-modulated by inputting it to mixers 31 and 32 in the mixer circuit 30. Therefore, a 90-degree phase shifter for processing a signal having the same frequency as the carrier is not required, and the number of circuits around which the signal goes around is reduced, so that the possibility of deterioration of the output signal due to feedback from the substrate of the IC or the power supply line is reduced. . In addition, the three 90's required in the conventional circuit of FIG.
Since the number of phase shifters can be two, the configuration can be simplified. Further, 10 and 20 have a function of phase correction, and even if the accuracy required for the 90-degree phase shifters 2 and 60 is loosened, accurate quadrature modulation is possible.

The signals output from the SSB modulation circuits 10 and 20 have frequencies of ωc1 + ωc2, ωc1-ωc2, ωc2-ω
There are 3 ways of c1. Here, considering the case of ωc1> ωc2, x = ωc1t and y = ωc2t, and I (t) and Q
When (t) is simply displayed as I and Q, the output signal S (t) has the following four types. S (t) = I · cos (x−y) + Q · sin (x−y) (1) S (t) = I · cos (x−y) −Q · sin (x−y) (2) S (T) = I * cos (x + y) + Q * sin (x + y) (3) S (t) = I * cos (x + y) -Q * sin (x + y) (4)

90 degree SSB modulation circuits 10 and 20 in FIG.
A circuit specifically showing the above will be described with reference to FIGS. The SSB modulation circuits 10 and 20 in FIG. 1 perform multiplication and addition so as to output the signals having different phases by 90 degrees, and output the signals represented by the formulas (5) and (6) or the formulas (7) and (8). . 2 to 7, the inverting circuit 40 is used, but these are for inverting the polarities in order to perform the subtraction of the equations (5) and (8). cos (x + y) = cosxcosy-sinxsiny (5) sin (x + y) = sinxcosy + cosxsiny (6) cos (xy) = cosxcosy + sinxsiny (7) sin (x-y) = sinxcosy-8 cosxsin, FIG. 2 in FIG. The phase shifter 2 outputs a signal c which is 90 degrees out of phase with the signal of the frequency ωc1 output from the oscillator 1.
Generate osx and sinx. On the other hand, 90 degree phase shifter 6
0 generates signals cosy and siny whose phases are different from each other by 90 degrees from the signal of the frequency ωc2 output from the oscillator 6.

The mixer 11 receives the co signal from the 90-degree phase shifter 2.
Input sx and cosy from the 90-degree phase shifter 60.
The mixer 12 includes sinx and 90 from the 90-degree phase shifter 2.
Input siny from the phase shifter 60. Mixer 21
Is the sinx from the 90 degree phase shifter 2 and the 90 degree phase shifter 6
Enter the cosy from 0. The mixer 22 inverts cosx from the 90-degree phase shifter 2 (180-degree phase shifter) 4
-Cosx passed through 0 and si from 90 degree phase shifter 60
Enter ny.

The adder 13 includes a mixer 11 and a mixer 12.
Output signals are added and cos (xy) is output. The adder 23 adds the output signals of the mixer 21 and the mixer 22, and outputs sin (xy). The mixer 31 inputs the output signal of the adder 13 and the baseband signal I (t). The mixer 32 inputs the output signal of the adder 23 and the baseband signal Q (t). The output signals of the mixer 31 and the mixer 32 are added to the output of the adder 33, and S (t) = I · cos (xy) + Q · of the equation (1) is added.
sin (xy) is obtained.

In FIG. 3, the mixer 11 includes cosx from the 90-degree phase shifter 2 and cos from the 90-degree phase shifter 60.
Enter y. The mixer 12 receives the s signal from the 90-degree phase shifter 2.
Input inx and siny from the 90-degree phase shifter 60. The mixer 21 inputs −sinx obtained by passing the sinx from the 90-degree phase shifter 2 through the inverting circuit 40 and the cosy from the 90-degree phase shifter 60. The mixer 22 includes cosx from the 90-degree phase shifter 2 and siny from the 90-degree phase shifter 60.
Enter

The adder 13 includes the mixer 11 and the mixer 12.
Output signals are added and cos (xy) is output. The adder 23 adds the output signals of the mixer 21 and the mixer 22, and outputs -sin (xy). The mixer 31
The output signal of the adder 13 and the baseband signal I (t) are input. The mixer 32 inputs the output signal of the adder 23 and the baseband signal Q (t). The output signals of the mixer 31 and the mixer 32 are added to the output of the adder 33, and S (t) = I · cos (xy) −Q in the equation (2) is added.
-Sin (xy) is obtained.

In FIG. 4, the mixer 11 has cosx from the 90-degree phase shifter 2 and cos from the 90-degree phase shifter 60.
Enter y. The mixer 12 receives the s signal from the 90-degree phase shifter 2.
-sinx that passed inx through the inverting circuit 40 and siny from the 90-degree phase shifter 60 are input. The mixer 21 has 9
The sinx from the 0-degree phase shifter 2 and the cosy from the 90-degree phase shifter 60 are input. The mixer 22 is a 90-degree phase shifter 2
Input cosx from and the siny from the 90-degree phase shifter 60.

The adder 13 includes the mixer 11 and the mixer 12.
Output signals are added and cos (x + y) is output. The adder 23 adds the output signals of the mixer 21 and the mixer 22 and outputs sin (x + y). The mixer 31 inputs the output signal of the adder 13 and the baseband signal I (t). The mixer 32 inputs the output signal of the adder 23 and the baseband signal Q (t). The output signals of the mixer 31 and the mixer 32 are added to the output of the adder 33, and S (t) = I · cos (x + y) + Q · of the equation (3) is added.
sin (x + y) is obtained.

In FIG. 5, the mixer 11 passes cosx from the 90-degree phase shifter 2 to the inverting circuit 40 -cosx.
And cosy from the 90-degree phase shifter 60. The mixer 12 inputs sinx from the 90-degree phase shifter 2 and siny from the 90-degree phase shifter 60. The mixer 21 has 9
The sinx from the 0-degree phase shifter 2 and the cosy from the 90-degree phase shifter 60 are input. The mixer 22 is a 90-degree phase shifter 2
Input cosx from and the siny from the 90-degree phase shifter 60.

The adder 13 includes the mixer 11 and the mixer 12.
Output signals are added and -cos (x + y) is output.
The adder 23 adds the output signals of the mixer 21 and the mixer 22 and outputs sin (x + y). The mixer 31
The output signal of the adder 13 and the baseband signal I (t) are input. The mixer 32 inputs the output signal of the adder 23 and the baseband signal Q (t). The output of the adder 33, which is obtained by adding the output signals of the mixer 31 and the mixer 32, is passed through the inverting circuit 41 to obtain S (t) = I · cos of the equation (4).
(X + y) -Q * sin (x + y) is obtained.

The same output as the circuits shown in FIGS. 2 to 5 is not limited to this, and a modification thereof can be realized by inserting an inverting circuit at an appropriate place. 6 and 7 are modified examples thereof. In the configuration of FIG. 6, the inverting circuit 40
Is connected to the output terminal of the mixer 11 and the inverting circuit 41 is connected to the output side of the adder 33. In this case as well, the same output S (t) = I.cos ( x
+ Y) -Q.sin (x + y) is obtained. 2 to 5 show an example in which a 180 ° phase shifter is provided between the oscillator 1 and the mixer circuits 11, 12, 21, and 22, but instead,
Even if a 180 ° phase shifter is inserted between the oscillator 6 and the mixer circuits 11, 12, 21, 22, it is obvious that exactly the same operation can be performed.

In the configuration of FIG. 7, the inverting circuit 40 is connected to the mixer 2.
However, in this case as well, the same output S (t) = I.cos (x-
y) -Q.sin (xy) is obtained. A similar output can also be obtained by inserting an inverting circuit on the output side of the 90-degree phase shifter 60.

Next, the phase correction operation when there is a phase error in the 90-degree phase shifters 2 and 60 will be described. The outputs of the adders 13 and 23 of FIG. 2 are shown in equations (9) and (10), respectively. Ci = (cosxcosy + sinxsiny) (9) Cq = sinxcosy-cox (x) sin (y) (10) The case where both 90-degree phase shifters 2 and 60 have a phase error is shown. Assuming that the phase errors of the 90-degree phase shifters 60 and 2 from 90 degrees are φ degree and θ degree, respectively, equations (9) and (10)
Is expressed as in equations (11) and (12). Ci = cos (x + φ) cosy + sinxsin (y + θ) (11) Cq = sinxcosy−cos (x + φ) sin (y + θ) (12) Using the formula of sum and difference from products of trigonometric functions, Ci = 1/2 [cos ( x + y + φ) + cos (x−y + φ)] + 1/2 [−cos (x + y + θ) + cos (x−y−θ)] (13) Cq = 1/2 [sin (x + y) + sin (x−y)] + 1 / 2 [-sin (x + y + φ + θ) + sin (x−y + φ−θ)] (14) In addition, using the product formula from the sum and difference formulas, Ci = cos ((φ + θ) / 2) cos (x−y + (φ-θ) / 2) -sin ((φ-θ) / 2) sin (x + y + (φ + θ) / 2) (15) Cq = cos ((φ-θ) / 2) sin (xy− (φ −θ) / 2) − sin ((φ + θ) / 2) cos (x + y + (φ + θ) / 2) (16) Among the items of Ci and Cq, an expression ((y−y)) which is a component of frequency (x−y) It can be seen that the phase difference of the first term of 15) and (16) is always maintained at 90 degrees regardless of the values of the phase errors φ and θ. The difference in the amplitude of the first term of the equations (15) and (16) slightly occurs when there are phase errors φ and θ, but this effect is extremely small and the phase error of the two 90 degree phase shifters is small. For example, if the angle is 2 degrees and 5 degrees, the amplitude error is 0.15%. The conversion value to the phase error is
Since it is 0.043 degrees, it can be seen that there is a strong correction effect of the phase error. Adder circuits 13, 23 and mixer 31,
Amplitude equalizing means such as limiters (32 in FIG.
It is easy to add 34, 35) in the above to make the amplitudes equal, and in that case, the frequency (x
The component of -y) becomes an exactly quadrature modulated signal. Further, although the case where the amplitude values of the output signals of the 90-degree phase shifters 2 and 60 are all the same is shown, the values of the 90-degree phase shifters 2 and 60
Even if the two output amplitudes are not the same, it has a very strong phase correction action although a slight phase error occurs. This effect was confirmed by computer simulation. The sin (x + y) and cos (x + y) terms are converted to different frequencies that are far apart from the target frequency and become spurious, but they are far from the carrier frequency, so they are filtered as in the indirect modulation method. It is possible to remove by using such as.

According to the embodiments described above, the 90-degree phase shifter operating at the same frequency as the output signal (carrier) is not provided. Therefore, even if there is feedback (unnecessary leakage) from the output signal, the number of electronic circuits that operate at the same frequency as the carrier that requires attention to signal wraparound will be smaller than in the past, which is a factor that deteriorates the quality of the output signal. Is less. According to the embodiment of the present invention, the number of the three 90-degree phase shifters in FIG. 9 according to the prior art is reduced to two, so that the circuit scale can be reduced and the circuit can be simplified. Further, since the SSB modulation circuits 10 and 20 have the effect of mutually correcting the phases, it is possible to perform accurate quadrature modulation without using the 90-degree phase shifters 2 and 60 with good accuracy.

[0027]

As described above, according to the present invention, there is provided a quadrature modulation circuit having good phase accuracy which is not easily affected by a jump in of a signal from the outside and has a small possibility of deterioration of an output signal. Can be provided. Further, as compared with the prior art, the circuit connected in series is reduced by 90 ° phase shifter 7, so that the number of stages is reduced, which is advantageous in terms of noise.

[Brief description of drawings]

FIG. 1 is a block diagram showing a configuration of a quadrature modulation circuit according to an embodiment of the present invention.

FIG. 2 is a block diagram showing a first specific example of the circuit of FIG.

FIG. 3 is a block diagram showing a second specific example of the circuit of FIG.

FIG. 4 is a block diagram showing a third specific example of the circuit of FIG.

5 is a block diagram showing a fourth specific example of the circuit of FIG.

6 is a block diagram showing a modified example for obtaining the same output as the circuit of FIG.

7 is a block diagram showing a modified example for obtaining the same output as the circuit of FIG.

FIG. 8 is a block diagram showing a configuration of a quadrature modulation circuit according to a conventional technique.

FIG. 9 is a block diagram showing a configuration of a transmitter according to the related art.

[Explanation of symbols]

 1/6 Oscillator 7/8/9 90 degree phase shifter 2/60 90 degree phase shifter 10/20 SSB modulation circuit 30 Mixer circuit 11/12/21/22/31/32 mixer 13/23/33 Adder 34.35 Amplitude equalizing means 40/41 Inversion circuit

Claims (2)

[Claims] 1. Means (1, 6) for providing a signal of a first frequency (ωc1) and a second frequency (ωc2), and using these signals, of the first and second frequencies First and second single sideband modulation circuits (10) that output first and second signals having a phase difference of 90 degrees with respect to the sum or difference frequency (ωc1 + ωc2 or ωc1-−ωc2 or ωc2-ωc1), respectively. , 20) and a first mixer circuit (3) for inputting a first signal to a first input terminal and a third signal (I) to a second input terminal.
1) and a second mixer circuit (3) for inputting a second signal to the first input terminal and inputting a fourth signal (Q) to the second input terminal.
2) and the output signals of the first and second mixer circuits are added to the first
Quadrature modulation circuit having an adding circuit (33). 2. A first single sideband modulation circuit (10) comprises a third and a fourth mixer circuit (11, 12) and a second adder circuit (13) for summing these outputs. The second single sideband modulation circuit (20) includes the fifth and sixth mixer circuits (21, 22) and a third addition circuit (23) that sums these, and the oscillator (1 Or 6) and the mixer circuit (11, 12, 2)
The quadrature modulation circuit according to claim 1, further comprising a 180 ° phase shifter disposed between the quadrature modulation circuit and the quadrature modulation circuit.