JPH04101667A - Inverter unit - Google Patents

Abstract

PURPOSE:To suppress switching loss and noise by providing a control circuit for setting the switching timing of a switching element so that the capacitive component of the switching element can be charge and discharged appropriately with the residual energy in a load circuit thereby turning the switching element OFF. CONSTITUTION:At time t0, control signal Vc2 has High level and a switching element SW2 is turned ON to feed a current I2. The timing for turning the switching element SW2 OFF at time t0 is set at a time point where the current I2 is slightly larger than 0. After the time t0, the current I2 decreases gradually and a capacitive component Csw1 is discharged whereas a capacitive component Csw2 is charged. Consequently, the potential V0 at the joint of the switching elements SW1, SW2 increases gradually. According to the constitution, the switching elements SW1, SW2 are prevented from being fed with a whisker-like current through charge/discharge of the capacitive components Csw1, Csw2, resulting in control of noise or switching loss.

Description

[Detailed description of the invention] [Industrial application field]

The present invention relates to an inverter device that includes a switching element that is controlled to be turned on and off, converts a DC power source into an AC output, and supplies the AC output to a load circuit.

[Conventional technology]

Conventionally, as this type of inverter device, for example,
As shown in FIG. 13, a pair of switching elements sw,
, sw2 is inserted between both ends of the power supply E, and both switching elements SW+ and SW2 are turned on and off alternately. Each switching element sw,, sw
2, the on period is controlled by the control circuit 1. The control circuit 1 includes an oscillation circuit 2 that outputs a pair of contradictory rectangular waves having a constant period as shown in FIGS. 14(a) and 14(b). One output V of the oscillation circuit 2 is converted into a voltage via the level shift circuit 3 (FIG. 14(C)), and then sent to the drive circuit 4. A control signal Vc to one switching element SW is generated via the switching element SW (FIG. 14(d)). In addition, the other output ■ of the oscillation circuit 2 generates a control signal V c 2 to the other switching element SW2 via the drive circuit 42 (FIG. 14(,)). As shown in Fig. 14(f),
This is because the voltage (2)0 across the switching element SW2 changes, and the reference voltage of the switching element SW+ changes. A load circuit 5 is connected between both ends of the switching element SW2. The load circuit 5 connects an inductance L and a capacitor Ca in series, and also connects an inductance L and a capacitor Ca in series.
DC cut capacitor cb and load between both ends of a! Which series circuit is connected? By the way, both switching elements S W 1.3 W 2
is the driver circuit 41.4 shown in FIGS. 14(d) and (e).
The on/off control is performed by the control signals V C1 and V CH from 2 and turned on and off alternately. In other words, the potential at the connection point of both switching elements SW + and SW 2 ■
. has a rectangular wave shape as shown in FIG. 14(f). When the switching element SW + is turned on at time t1, the current tends to flow due to the energy stored in the load circuit 5, so the direction of the current ■ flowing through the switching SW1 is as shown in FIG. 14 (h). After going in the opposite direction to the arrow in FIG. 13, it reverses. Furthermore, when the switching element SWt is turned on at time t2, the direction of the current I2 flowing through the switching element SW2 due to the accumulated energy of the load circuit 5 is opposite to the arrow in FIG. 13, as shown in FIG. 14(g). After becoming
Reverse. By repeating such operations, a load current (2) having an alternating current waveform as shown in FIG. 14(i) flows through the load (1). Here, in FIG. 14, the oscillation circuit 2
It is assumed that the output frequency of the load circuit 5 is higher than the resonant frequency of the load circuit 5. That is, the phase of the resonant current is delayed with respect to the output of the oscillation circuit 2, and the direction of the currents I, I2 flowing when the switching element 5WSW2 is switched is in the negative direction.
The direction is opposite to that shown by the arrow in Figure 3). If the phase relationships are reversed, as shown in FIG. 15, current will flow in the positive direction when switching elements SW, SW2. In this case, the state changes from a negative current flowing through one switching element to a positive current flowing through the other switching element, and instantaneously the switching element through which the negative current was flowing changes. Since a voltage is applied, a reverse recovery current, ie, a recovery current, flows due to the accumulation of carriers when a negative current flows. As shown in Section 15), this current becomes a whisker-like current and causes switching loss and noise. Therefore, it is normal to set the output frequency of the oscillation circuit 2 higher than the resonant frequency of the load circuits, but even if the output frequency of the oscillation circuit 2 is higher than the resonant frequency of the load circuit 5, as shown in FIG. If the operation is as shown in , the switching elements sw, ,
When switching sw2, the current and voltage change rapidly, causing switching loss and noise in the switching elements SW+ and SW2. Therefore, as shown in FIG. 16, it is conceivable to make the frequency of the output V of the oscillation circuit 2 approximately equal to the resonant frequency of the load circuit 5. In this way, the switching times of the switching elements sw, , sw2 almost coincide with the zero-cross point of the resonant current of the load circuit 5, reducing switching loss and reducing noise. On the other hand, if you try to control the energy supplied to the load l while maintaining this state, as in the case of dimming a lighting load, the output frequency of the oscillation circuit 2 and the load circuit 5
It is necessary to change the resonant frequency in conjunction with the resonant frequency. Although it is easy to adjust the output frequency of the oscillation circuit 2,
Adjusting the resonant frequency of the load circuit 5 causes problems such as an increase in the number of parts and a complicated circuit configuration, which is not practical. Furthermore, if the energy supplied to the load is to be changed continuously, the resonant frequency of the load circuit 5 must also be changed continuously, which is difficult to realize. Therefore, by using a load circuit with a configuration in which the resonant frequency changes depending on the supplied energy and adjusting the power supply voltage, the switching timing of the switching element can be made to approximately coincide with the zero-crossing point of the resonant current of the load circuit. It is conceivable to control the That is, as shown in FIG. 17, a pair of switching elements SW+,
A series circuit of SW2 and impedance Z is connected between the output terminals of the voltage control circuit 6, and both switching elements SW1SW
2 are alternately turned on and off by the control circuit 1 as described above.
Control to turn off. The control circuit 1 operates each switching element SW + in synchronization with the rectangular wave output of the oscillation circuit 2.
, a drive circuit 41 that turns on SW2 alternately.
．． 42, an oscillation circuit 2 and a drive circuit 4. A level shift circuit 3 for converting voltage is inserted between the two. A load circuit 5 is connected between both ends of the series circuit of one switching element SW2 and the impedance Z. The load circuit 5 includes a series circuit of an inductance L and a capacitor Ca, and has a configuration in which a series circuit of a DC cut capacitor cb and a load p is connected between both ends of the capacitor Ca. Loads here! Assume that a discharge lamp such as a fluorescent lamp is used. On the other hand, the voltage across the impedance Z corresponding to the current flowing through the switching element SW2 is input to the zero cross detection circuit 7, the zero cross point is detected, and the voltage across the impedance Z corresponding to the current flowing through the switching element SW2 is input to the zero cross detection circuit 7, and the zero cross point is detected.
The voltage control circuit 6
control the output voltage of the That is, the voltage control circuit 6 is
The power supply E is used as an input, and the output voltage is changed according to the output of the zero-cross detection circuit 7. The configuration shown in FIG. 17 operates as shown in FIG. 18. 1st
The left half of FIG. 8 shows the operation when the output frequency of the oscillation circuit 2 is increased in order to reduce the energy supplied to the load 1 without providing the voltage control circuit 6 and the zero-cross detection circuit 7. Since the load 1 is a discharge lamp such as a fluorescent lamp, it exhibits negative characteristics in the lit state, and as the lamp current increases, the lamp voltage decreases. That is, it has a property that the impedance changes depending on the supplied energy. Therefore, when the output frequency of the oscillation circuit 2 increases, the impedance of the load circuit 5 increases,
The energy supplied to load l decreases. Further, as shown in FIGS. 18(g) and (h), the switching elements sw1. sW
The current flowing through the load circuit 5 is delayed in phase with respect to the resonant current of the load circuit 5, and the current flowing to the load l decreases.
Since the switching times of I and SW2 are shifted, switching loss increases. The right half of FIG. 18 shows the operation when the voltage control circuit 6 and zero-cross detection circuit &+87 are activated. As shown in FIG. 18(j), when the output voltage V1 of the voltage control circuit 6 is reduced, the energy supplied to the load circuit 5 is reduced, so that the impedance of the load 1 is increased. As a result, the resonant frequency of the load circuit 5 increases and becomes approximately equal to the output frequency of the oscillation circuit 2. In this way, the switching timing of the switching elements SW+ and SW2 and the resonance of the load circuit 5 can be made to substantially coincide with the zero-crossing point of the flow. In short, when increasing the output frequency of the oscillation circuit 2 to decrease the load current, the output voltage of the voltage control circuit 6 is decreased, and when decreasing the output frequency of the oscillation circuit 2 to increase the load current, the voltage control circuit If the output voltage of 6 is increased, the switching elements S W l, S W 2
It is possible to adjust the energy supplied to the load l while keeping the switching timing of the load circuit 5 substantially coincident with the zero-crossing point of the resonant current of the load circuit 5.

[Problem to be solved by the invention]

By the way, as shown in FIG. 19, the switching element S
W, , SW7 has capacitance components C5wt and C5w2. Therefore, the current ■2 flowing through the switching element SW2 is detected, and the switching element SW2 is detected at the zero cross point.
2 is turned off, the current from the load circuit 5 disappears, so the potential at the connection point of both switching elements SW, SW2. It becomes an opening. At this time, the switching element S
When W1 is turned on, the potential V. is raised to a high level. As a result, as shown at time t0 in FIG. 20, the potential V. Due to the rise in C, a charging current flows rapidly into the capacitance components Csw, , and whisker-like currents are generated as currents I+ and Ih. Also, at time 1+, when the switching element SW2 is turned on at the zero point of the current ■1, the potential V0 becomes a low level, and a whisker-like current that charges the capacitance component Csw flows. In this way, the zero crossing point of the current is detected and the load circuit 5
When trying to reduce noise and switching loss by controlling the energy supplied to the switching elements sw,
, sw2 due to the presence of capacitive components Csw, , C5w2, both switching elements SW1 . A situation occurred in which current simultaneously flowed through SW2, and noise and switching loss could not be reduced in some cases. The present invention aims to solve the above-mentioned problems, and is designed so that when the switching element is turned off, the energy of the load circuit can cancel out the charge accumulated in the capacitance component of the switching element in just the right amount. The present invention aims to provide an inverter device in which switching loss and noise are reduced by turning off the switching elements at the point when the switching element is turned off.

[Means to solve the problem]

In order to achieve the above object, the present invention provides an inverter device that includes a switching element that is controlled on and off, converts a DC power source into an AC output, and supplies the AC output to a load circuit. The load circuit is configured so that the resonant frequency changes, and a zero-crossing detection circuit detects the vicinity of the zero-crossing point of the resonant current of the load circuit, and a capacitive component of the switching element remains in the load circuit based on the output of the zero-crossing detection circuit. A control circuit is provided to set the switching timing of the switching elements so that the battery can be charged and discharged with just the right amount of energy.

[Effect]

According to the above configuration, the load circuit is configured so that the resonant frequency changes according to the supplied energy, and the zero-cross detection circuit detects the vicinity of the zero-cross point of the resonant current of the load circuit, and the A control circuit is provided to set the switching timing of the switching element so that the capacitance component of the switching element can be charged and discharged with just the right amount of energy remaining in the load circuit, so when the switching element is turned on and off, The capacitive component of the switching element can be charged or discharged with the energy remaining in the load circuit, and generation of whisker-like current due to the charging current to the capacitive component of the switching element can be prevented. As a result, noise and switching loss during on/off switching of the switching element can be reduced.

Embodiment 1 FIG. 1 shows the basic configuration of the present invention, and FIG. 2 shows operating waveforms of each part in FIG. 1. Each switching element sw,, sw
2 is the control signal V, as shown in FIG. 2 (aHb).
c, , Vc2 to turn on and off alternately, and both control signals V (1°V c2 have a period (t
+ to) and (tz tz) are provided. Since the control signal Vc2 is at a high level until time t0, the switching element SW2 is on, and the timing at which the switching element SW is turned off at time t0, when the current is flowing, is as shown in FIG. As in d), the current ■2
is set at a point in time when is slightly larger than 0. FIG. 3 shows the equivalent circuit at this point. The current I2 gradually decreases after time t0, discharging the electric charge of the capacitive component Csw of the switching element SW+, and charging the capacitive component C5w2 of the switching element SW2. Therefore, the potential at the connection point of both switching elements SW + and SW 2 is ■. increases over time. In the present invention, the control signal MCI
The potential V becomes high level until time t1 when the switching element SW is turned on. The lowest value of the current ■2 is set so that the voltage is approximately equal to the voltage of the power source E, and the potential ■2 is set during the period (t s - t2). Set the minimum value of the current (1) so that it becomes almost 0. In other words, the switching elements SW1 . SW
2, it is possible to prevent whisker-like current from flowing, and noise and switching loss can be suppressed. Switching elements SW+, S whose specific circuit is shown in Figure 4
A field effect transistor is used for W2. The oscillation circuit 2 is a timer a known as r555J.
The oscillation circuit 2 is constructed by externally attaching peripheral components to an m circuit, and as shown in FIG. 5(a), the output V of the oscillation circuit 2 provides high and low binary voltages alternately at a constant period. The output V of the oscillation circuit 2 is
The resistor R
Applied across I+, resistor R+2, capacitor CI2
, a drive circuit 4 consisting of a buffer via a rise delay circuit consisting of a diode D,2. is input as a voltage V. In the rise delay circuit, the drive circuit 41
When the input waveform rises, the rise is delayed by the resistor R12 and the capacitor CI2, and when the input waveform falls, the charge in the capacitor C12 is rapidly discharged through the diode DI2 so that there is no delay in the fall. This is what I did. In addition, the oscillation circuit 2
The output V is inputted to the drive circuit 42 consisting of an inverting circuit via a rise delay circuit consisting of a resistor R11, a capacitor C13, and a diode D11.
2, the rise of the input waveform is also delayed. The output of each drive circuit 41, 42 is a control signal Vc, which turns on/off the switching elements SW, SW2, respectively.
, Vc2. In this way, by delaying the rise of the input waveforms to each of the drive circuits 41 and 42, it is possible to provide a period in which the control signals VC1 and VC2 are simultaneously turned off. Note that the delay operation may be performed using a logic circuit instead of the rise delay circuit. The voltage control circuit 6 is a step-up chopper circuit in which a series circuit of a choke coil CH and a transistor T1, which is a switching element, is inserted between both ends of a power supply E, and a diode D and a capacitor are inserted between the collector and emitter of the transistor T1. Connect the series circuit with C1, transistor T1
A control signal is input to the base of capacitor C.
The output voltage Vi is obtained between both ends of 5. The transistor T1 is controlled by the output of an on-period control circuit 8, which is constructed by externally attaching peripheral components to a timer integrated circuit known as r555J. The on-period control circuit 8 is configured to be able to change the on-period of the transistor T by adjusting the impedance of the series circuit of the resistor R8 and the transistor T2. Further, the impedance of transistor T3.1 is adjusted by adjusting the impedance of transistor T3. T, and resistors R4 to R9. That is, in the reference state, transistor T3 is off and transistor T4 is off.
is on, and transistor T2 is almost completely on. As the on-period of the transistor Tz increases, the impedance of the transistor T2 decreases, the on-period of the transistor T1 becomes shorter, and the output voltage Vi of the voltage control circuit 6 becomes lower. Furthermore, if the off-period of the transistor T4 increases, the impedance of the transistor T2 increases, the on-period of the transistor T1 becomes longer, and the output voltage Vi of the voltage control circuit 6 becomes higher. Resistor R2, capacitor C2, Zener diode ZD2
is the drive circuit 4. The power supply circuit, resistor R, capacitor C, and Zener diode ZD3 are the control circuit 1.
This is the power supply circuit for other parts. The zero cross detection circuit 7 divides the voltage across the load circuit 5 using resistors R- and Rs! The vicinity of the zero-crossing point of the resonant current of the load circuit 5 is detected based on the voltage and the voltage across the impedance Z connected in series with the switching element SW. The configuration and operation of the zero-cross detection circuit 7 will be explained below with reference to FIGS. 5 to 7. FIG. 5 shows the operation when the zero-cross point of the resonant current of the load circuits and the switching timing of the switching elements sw, , sw2 almost coincide with each other. In this case, Fig. 5<h>
The voltage Va across the impedance 2 shown in is the reference voltage E of the comparator 11. lower, period (t, −t o
), (t, -t2) has a capacitance component CSwl+C
Charge and discharge sw2, and the potential becomes ■. The output level vb of the comparator 11 shown in FIG. 4(i) becomes "L" while having time for inversion. Therefore, the output level Ve of the AND circuit 12 shown in FIG. 5(1) is "L", and the fifth
The output level Vf of the inversion circuit 13 shown in the figure (-) is "H"
become. Therefore, the transistor T is turned off and the transistor T is turned on, resulting in the above-mentioned reference state, and the output voltage V1 of the power supply control circuit 6 is maintained as it is. Next, load! When the output frequency of the oscillation circuit 2 is increased to reduce the energy supplied to the
), when the switching elements SW, SW2 are switched, a large current II2 flows in the opposite direction to the arrow in FIG. As shown in figure (h), voltage across impedance 2 V a #J current 11. It occurs in the same direction as I2. Further, a falling detection circuit is configured by the inverting circuit 14, the capacitor C3o, and the resistor Rho, and when the voltage across the load circuit 5 falls, as shown in FIG. Since the input voltage Vc also falls, capacitor C1° and resistor R1o
The potential Vd at the connection point with , increases over time as shown in FIG. 6(k). At this time, the output level vb of the comparator 11 also temporarily becomes "H" 6 Therefore, as shown in FIG.
As shown in 1), the output level Vc of the AND circuit 12 temporarily becomes "H", the transistor T is turned on, and the impedance of the transistor T2 is reduced. On the other hand, when the potential at the connection point between capacitor C1o and resistor R3o rises and the input level of AND circuit 12 is "H", the input terminal of AND circuit 16 becomes "L" through inverting circuit 15 and is interlocked. Therefore, the output level of the AND circuit 17 is also "L", and the inverting circuit 1
The output level Vf of No. 3 is kept at "H". therefore,
The transistor T4 continues to be in the on state in the reference state. As described above, the on-period control cycle F! shown in No. 65 (n) is completed! The output Vg of @8 reduces the on period of the transistor T, and the output voltage vi of the voltage control circuit 6 shown in FIG. 6(0) decreases. As a result, the energy supplied to the load circuit 5 decreases, the impedance of the load p increases, and the resonant frequency increases, as shown in FIG. 6(f) (g>(h
) As shown by the broken line in (i), the switching element SW+
, SW2 changes, and the switching timing of the switching elements SW, , SW2 almost coincides with the zero-crossing point of the resonant current of the load circuit 5. Next, a case where the output frequency of the oscillation circuit 2 is reduced so as to increase the energy supplied to the load l will be explained based on FIG. 7. In this case, the switching element SW
,, the current II2 flowing through SW2 is the 71st M(f)(F
As shown in i), the switching elements SW, , SW
2, a current flows in the direction shown by the arrow in FIG. Therefore, as shown in FIG. 7(h),
The voltage Va across the impedance Z also has a polarity corresponding to the currents II and I2. Input voltage Va to comparator 11
When becomes negative, the output level vb of the comparator 11 becomes “
In addition, as shown in FIG. 7(e), the load circuit 5
During the period when the voltage V0 across is 0, the output level of the inverting circuit 18 is "H", and in the latter half of this period, as shown in FIG. 7(k), the capacitor C3° and Since the potential Vd at the connection point with the resistor R10 is 0 and the output level of the inverting circuit 15 is "H", the output level of the AND circuit 16 is "H'". Therefore, the output level of AND circuit ]7 becomes "H", and the output level of FIG.
The output level Vf of the inverting circuit 13 shown in is temporarily “L”.
become. As a result, the transistor T is turned off, the output Vg of the on-period control circuit 8 becomes as shown in FIG. It becomes expensive. That is, as the energy supplied to the load l increases, the impedance of the load p decreases, and the resonant frequency of the load circuit 5 decreases, causing the current ■1. I2 changes. As described above, if the output frequency of the oscillation circuit 2 is changed to adjust the energy supplied to the load circuit 5, the output voltage of the voltage control circuit 6 is changed, thereby changing the switching timing of the switching elements sw, , sw2. However, load circuit 5
is controlled so that it almost coincides with the zero-crossing point of the resonant current. Moreover, one of the switching elements SW, SW2
Due to the residual energy in the load circuit when is off,
Capacitive components Csw of switching elements SW+ and SW2,
After charging and discharging Csw2 to invert the potential V0, the other switching element sw2. sw is turned on, and noise and switching loss can be reduced. In the above embodiment, a field effect transistor is used as the switching element 5WSW2, but a switching element such as a transistor or a thyristor with a diode connected in antiparallel may also be used. Further, although a step-up type chopper circuit is used for the voltage control circuit 6, it may be a step-down type, and is not limited to the type of the embodiment as long as the output voltage can be varied.

Embodiment 2 This embodiment shows an example in which the output is controlled by controlling the duty ratio of the control signals V C1 and V C2. In this case as well, as shown in FIG. 8, the point in time when each switching element SW, SW2 is turned off is set to the point in time when energy remains in the load circuit 5, and the connection point between both switching elements SW+ and SW2 is set. The potential of■. By setting the capacitance component C5w1°C3ll+2 to the minimum value that allows charging and discharging when is reversed, the same effect as in Example 1 is obtained.

[Embodiment 3] This embodiment is a case where a voltage control circuit is not used, and as shown in FIG. By connecting in series a load 2' that changes in impedance, the change in impedance is canceled out. Even in this case, one of the switching elements sw, , sw2
When turns off, the capacitive component Cswl, C3112
Charge and discharge the electric charge to create the electric potential■. After reversing , the other switching elements S W 2 and S W t are turned on, whereby the same effect as in the first embodiment can be obtained. Note that the loads Z and Z' may be connected in parallel.

[Embodiment 4] In this embodiment, as shown in FIG. 10, instead of controlling the switching elements sw, sw2 by the output of the oscillation circuit 2, the inductance of the load circuit 5 is controlled by the choke coil CH, Inductance La with feedback winding
The automatic configuration is such that the switching elements sw, , sw2 are controlled by the output of the feedback winding. A switch SW is connected in parallel to the choke coil CH, and by opening and closing the switch SW, the energy supplied to the load I is adjusted. Also in this configuration, the same effects as in the first embodiment can be obtained.

[Embodiment 5] In this embodiment, as shown in FIG. 11, the switching element SW2 is on/off controlled by the oscillation circuit 2 as in the first embodiment, and the switching element SW + is controlled by the load circuit. It is controlled by the output of a feedback winding connected to an inductance of 5. Even with this configuration, the effect of reducing switching loss is
This is the same as in Example 1.

[Embodiment 6] As shown in FIG. 12, this embodiment is an example in which the alternating current power supply AC is controlled at a stage before the rectification and smoothing circuit 11 that obtains the power supply E, and is effectively an embodiment of the present invention. It is the same as 1. Although the basic configuration of the embodiment shown above is a half-bridge type inverter device, even a full-bridge type or single-stone type can be configured to switch the switching elements at the zero-crossing point of the resonant current of the load circuit. If you have something,
The technical idea of the present invention is applicable to any inverter device having any configuration. Further, the power source E may be a rectified and smoothed AC power source, and the AC power source may only be rectified for input to the voltage control circuit 6 having the configuration shown in the first embodiment. Furthermore, the impedance Z for detecting the zero-crossing point of the resonant current is not limited to the position of the embodiment.

【Effect of the invention】

As described above, the present invention configures a load circuit so that the resonant frequency changes depending on the supplied energy, and includes a zero-cross detection circuit that detects the vicinity of the zero-cross point of the resonant current of the load circuit, and an output of the zero-cross detection circuit. A control circuit is provided to set the switching timing of the switching element so that the capacitance component of the switching element can be charged and discharged with just the right amount of energy remaining in the load circuit. At times, the capacitive component of the switching element can be charged or discharged with the energy remaining in the load circuit, and generation of whisker-like current due to charging current to the capacitive component of the switching element can be prevented. As a result, it is possible to reduce noise and switching loss when the switching element is turned on and off.

[Brief explanation of the drawing]

FIG. 1 is a schematic configuration diagram showing the basic configuration of Embodiment 1 of the present invention, FIG. 2 is an explanatory diagram of the same operation, FIG. 3 is an equivalent circuit diagram when the switching element is switched, and FIG. 4 is the same as the above. 5 to 7 are operation explanatory diagrams of the same as above, FIG. 8 is an operation explanatory diagram of the second embodiment of the present invention, and FIG. 9 is a schematic diagram showing the third embodiment of the present invention. FIG. 10 is a circuit diagram showing Embodiment 4 of the present invention, and FIG. 11 is a circuit diagram showing Embodiment 5 of the present invention.
12 is a circuit diagram of a main part showing a sixth embodiment of the present invention, FIG. 13 is a circuit diagram showing a conventional example, FIGS. 14 to 16 are operation explanatory diagrams of the same, and FIG. 17 18 is a circuit diagram showing another conventional example, FIG. 18 is an explanatory diagram of the operation of the same, FIG. 19 is an equivalent circuit diagram showing the problem of the above, and FIG. 20 is an explanatory diagram of the operation of the same. 5...Load circuit, 7...Zero cross detection circuit, E.
・・Power supply, sw, , sw2 ・・Switching element, l
··load. Agent Patent Attorney Ishi 1) Chief 7 Figure 3 Figure 5 Figure 6 Figure 8 Figure 9 Figure 7 (o) Vi -111111 Figure 10 Figure 1 Figure 2 Figure 13 Figure 5 Figure 14 Figure 17 Figure 19 Figure 20 Procedure amendment (voluntary) December 8, 1990

Claims (1)

[Claims] (1) Equipped with a switching element that is controlled on and off,
In an inverter device that converts DC power into AC output and supplies AC output to a load circuit, the load circuit is configured so that the resonant frequency changes depending on the supplied energy, and and a control circuit that sets the switching timing of the switching element so that the capacitance component of the switching element can be charged and discharged with just enough energy remaining in the load circuit based on the output of the zero-crossing detection circuit. An inverter device comprising: