JP5042880B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply device that outputs a DC stabilized voltage / current that is insulated from an input power supply side, and more particularly, to a device that achieves a reduction in switching element loss and a reduction in size and cost.

  Conventionally, a composite resonance type series converter circuit is known as a small-sized and high-efficiency switching power supply as described above. FIG. 10 shows a main circuit configuration according to the prior art, and FIG. 11 shows waveforms of main parts.

  In this switching power supply device, a series circuit of switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of the DC input power supply E, and a capacitor C0 is connected. A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 Alternatively, a capacitor C2 is connected in parallel with either Q2 (in FIG. 10, one end of the DC input power supply E is connected to the low voltage side, and the capacitor C2 is connected in parallel to the power MOSFET Q2). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively (in many cases, they are also used as body diodes for power MOSFETs Q1 and Q2).

  Further, an intermediate tap is provided in the output winding of the output transformer T and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. The power MOSFETs Q1 and Q2 are alternately turned ON / OFF at a preset frequency in consideration of the complex resonance condition by the control unit 1 shown as a block. Therefore, the control unit 1 has a high-frequency oscillation function, a function of alternately driving the two power MOSFETs Q1 and Q2, and a function of setting a dead time period for turning off the two power MOSFETs Q1 and Q2. A feedforward, feedback control function and output variable function for controlling voltage, current, and power are provided.

  Referring to FIG. 11, Vg1 and Vg2 indicate drive signals for power MOSFETs Q1 and Q2 set in advance by control unit 1. A dead time period in which both are turned ON / OFF alternately and both are turned OFF is set. VQ1, IQ1 and VQ2, IQ2 indicate the drain-source voltage and drain current of the power MOSFETs Q1, Q2. When the drive signal Vg1 is High, the drain current IQ1 flows through the power MOSFET Q1, and when it is Low, the voltage VQ1 substantially equal to the DC input power source E is applied (the same applies to the power MOSFET Q2). In the dead time period, the drain-source voltages VQ1 and VQ2 have rising and falling waveforms with an arbitrary slope due to the effect of the capacitor C2, the inductor L, and the excitation inductance of the output transformer T. The drain currents IQ1 and IQ2 have a series resonance current waveform set approximately by the inductor L and the capacitor C1, and these combined currents are the series resonance of the inductor L, the primary winding L11 of the output transformer T, and the capacitor C1. It becomes the current of the circuit. VC1 represents a voltage waveform of the capacitor C1, and has a waveform delayed in phase from the current of the series resonance circuit. ID3 and ID4 indicate current waveforms of the output rectifier diodes D3 and D4. The relationship between the drive frequency of the power MOSFETs Q1 and Q2 and the series resonance frequency of the inductor L and the capacitor C1 is expressed as “resonance frequency> drive frequency. By satisfying this condition, it is possible to set the current of one of the diodes D3 and D4 to flow and then the other current starts to flow. During the period when both diode currents do not flow, power is transmitted to the output side. Not. That is, during the period when the currents of the diodes D3 and D4 do not flow, the secondary side of the output transformer T is considered to be unloaded, and the primary side exciting inductance L of the transformer T is inserted in series into the primary side series resonance circuit. As a result of the switching of the series resonance condition, inflection points are also found in the waveforms of the drain currents IQ1 and IQ2.

  In such a complex resonance type series converter, ZVS (zero voltage switching), that is, a condition setting such that a current starts to flow after the applied voltage of the switching elements Q1 and Q2 is reduced is possible, and switching loss is extremely small. Since the recovery loss of the secondary side rectifier diodes D3 and D4 can be avoided, high frequency can be achieved with high efficiency. Further, since the voltage / current waveform at the time of switching is stable and the ringing of the secondary side rectifier diodes D3 and D4 can be suppressed, the noise is also excellent.

  The above-described prior art is a so-called separately-excited switching power supply that drives the two switching elements Q1 and Q2 by setting the frequency in advance with an oscillator while having such various features. A level shifter is required for the potential side switching elements Q1 and Q2. From the viewpoint of frequency followability and loss, there is a technical problem with high frequency, and from a cost viewpoint, for example, Patent Documents 2 to 4 Self-excited studies are also being conducted as shown.

  FIG. 12 is an electric circuit diagram showing a known example of a current feedback type self-excited composite resonance series converter, which is shown in Patent Document 2. Although the main circuit configuration is substantially the same as that in FIG. 10, a commercial power supply is used for full-wave rectification as an input power supply, and the secondary side of the output transformer T02 is a full-wave rectification circuit using a rectification bridge DB. The switching elements TR1 and TR2 are composed of bipolar transistors, and the driving is performed using the secondary windings LB1 and LB2 of the current feedback transformer T01, and the dead time is secured by the reverse bias means and the driving conditions are improved.

  On the other hand, FIG. 13 is an electric circuit diagram showing a known example of a voltage feedback type self-excited composite resonance series converter, which is shown in Patent Document 3. A feedback signal to the bipolar transistors TR1 and TR2 as switching elements is obtained by providing a feedback winding NB2 in the output transformer PIT, and feedback control for load variation countermeasures and output stabilization is added.

  These conventional examples of the self-excited composite resonant series converter shown in FIGS. 12 and 13 suggest the possibility of simplifying the drive circuit. However, the number of separately excited composite resonant series converters described with reference to FIG. It is thought that it is difficult to realize the same feature from the following points. First, it is the time constant and delay element of the feedback circuit that determines the driving frequency, and it seems difficult to maintain the optimum switching conditions such as ZVS against fluctuations in external factors such as load fluctuations. Next, a bipolar transistor is assumed as a switching element, which is not suitable for high frequency operation.

FIG. 14 is a block diagram showing a known example of a voltage feedback type self-excited separately excited composite resonance series converter using a power MOSFET, which is shown in Patent Document 4. In this prior art, power MOSFETs Q1 and Q2 are used as switching elements, and the power source of the main control circuit 4 and the sub control circuit 5 is secured from the auxiliary winding L12 of the output transformer T ′, and this winding voltage is used as a signal source. The main control circuit 4 controls ON / OFF of the low-voltage side power MOSFET Q2 so that the supply voltage to the load Load becomes constant, and the sub-control circuit 5 reduces the voltage between the terminals of the high-voltage side power MOSFET Q1 below the reference voltage. Is turned on to maintain the ZVS of the high-voltage power MOSFET Q1.
Japanese Patent No. 2734296 Japanese Patent No. 3371595 JP 2002-262568 A JP 2006-129548 A

  In the above-described prior art, a specific example as shown in FIG. 15 is shown in the sub-control circuit 5. The drain-source voltage of the power MOSFET Q 1 is divided by the resistors r 01 and r 02, and the reference voltage is output by the comparator 7. Compared with FIG. 8, the power MOSFET Q <b> 1 is turned on by charging and discharging the capacitor c <b> 1 of the integration circuit. In such a configuration, the ON timing is determined while directly determining the drain-source voltage of the power MOSFET Q1 on the high voltage side. Therefore, the ZVS may be realized, but the following problems are encountered. Conceivable. First, it is necessary to divide the drain-source voltage of the power MOSFET Q1, and there is a concern about the loss due to the resistors r01 and r02. On the other hand, when these resistors r01 and r02 are configured with high resistance, a delay time due to the input capacitance of the comparator 7 is assumed, and it is considered difficult to increase the frequency. Next, the drive control circuits 4 and 5 become complicated, and no superiority over the separate excitation system is recognized.

  The object of the present invention is to greatly simplify the circuit and reduce the cost of a self-excited composite resonant series converter of voltage feedback type compared to other types of excitation while maintaining the original characteristics of low loss and low noise. It is to provide a switching power supply device that can be realized.

  In the switching power supply device of the present invention, a series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and the connection point between the first and second switching elements and the DC input power supply A series circuit including an inductor, a capacitor, and a primary winding of a transformer is connected between one terminal and a secondary side induced current of the transformer obtained by switching of the first and second switching elements. Rectified and smoothed by a smoothing capacitor and output, and the first and second switching elements are respectively supplied with voltages induced in the first and second auxiliary windings of the transformer by the first and second control circuits. A switch consisting of a voltage feedback self-excited complex resonance series converter that is continuously switched on and off. In the power supply apparatus, the first and second control circuits include third and fourth switching elements for turning off the first and second switching elements, respectively, and the first and second auxiliary windings. A peak hold circuit for peak-holding the induced voltage generated at the time, and when the induced voltage falls below a predetermined level from the hold voltage by the peak hold circuit, the third and fourth switching elements are turned on to turn the first and And a comparator for turning off each of the second switching elements.

  According to said structure, the series circuit which consists of a 1st and 2nd switching element is connected between both terminals of DC input power supply, and the connection point of said 1st and 2nd switching element and one of said DC input power supplies A series circuit comprising an inductor, a capacitor, and a primary winding of a transformer is connected between the first and second terminals, and voltage feedback using the induced voltage of the first and second auxiliary windings for ON / OFF driving of the switching element. Switching power supply comprising a self-excited composite resonance series converter of the type, the first and second switching circuits for turning ON / OFF the switching element from the induced voltage of the auxiliary winding are the first and second switching circuits. The induced voltage generated in the third and fourth switching elements for turning off the elements and the first and second auxiliary windings is reduced. A peak hold circuit for holding, and when the induced voltage is lowered by a predetermined level or more from a hold voltage by the peak hold circuit, the third and fourth switching elements are turned on to turn on the first and second switching elements. Each of them is configured to include a comparator that is turned off.

  Accordingly, the first and second control circuits indicate that the secondary-side induced current stops flowing due to the completion of charging of the secondary-side smoothing capacitor. Detecting from the drop, the ON side of the first and second switching elements is driven OFF, and the other voltage increase is detected to detect OFF of the first and second switching elements. The switching is continued by turning on the side that has been turned on. Thus, in the voltage feedback type self-excited complex resonance series converter, appropriate switching conditions can be realized with a simple configuration, while maintaining the low loss and low noise that are the original features, and a circuit that is significantly larger than the other excitation type. Simplification and cost reduction can be realized. Moreover, it can be adapted to higher frequency, smaller size and higher efficiency.

  In the switching power supply of the present invention, the third and fourth switching elements are bipolar transistors.

  According to the above configuration, by using a bipolar transistor as the third and fourth switching elements, the hold charge of the peak hold circuit can be discharged by the base current of the bipolar transistor, and a separate discharge unit is provided. Even if the third and fourth switching elements are turned on for a moment, the disappearance time of minority carriers can be used for an appropriate delay time, and function as a delay circuit for adjusting dead time. You can also.

  Preferably, the peak hold circuit includes a series circuit of a diode and a capacitor for peak-holding an induced voltage generated in the first and second auxiliary windings in a direction to turn on the first and second switching elements. The first and second control circuits further include switch means for discharging the charge of the capacitor after the comparator turns on the third and fourth switching elements.

  Furthermore, in the switching power supply device of the present invention, the first and second auxiliary windings of the transformer are connected to the first and second switching elements via a gate resistor, and the first and second The second control circuit is connected to a peak hold circuit and a comparator.

  According to the above configuration, the voltage induced in the first and second auxiliary windings of the transformer is applied to the first and second switching elements via the gate resistance to turn them on, The voltage induced in the first and second auxiliary windings of the transformer is also supplied to the peak hold circuit and the comparator of the first and second control circuits via the gate resistance.

  Therefore, the rise of the gate voltage of the first and second switching elements can be moderated, which is suitable for forming a dead-off time in which both the first and second switching elements are turned off.

  In the switching power supply device of the present invention, the comparator includes a MOSFET.

  According to the above configuration, the fact that the induced voltage generated in the first and second auxiliary windings has decreased by a predetermined level or more from the hold voltage by the peak hold circuit is determined using the gate threshold voltage of the MOSFET. Can be detected.

Furthermore, in the switching power supply apparatus of the present invention, a load connected between the terminals of the smoothing capacitor, characterized in that an LED.

  According to the above configuration, the current does not flow until a voltage equal to or higher than the forward voltage of the LED is applied, and is in a so-called no-load state at the time of startup. As well as improving the above, there is an effect of suppressing the rush current to the LED generated when the LED load is attached / detached.

  As described above, the switching power supply according to the present invention is a switching power supply comprising a voltage feedback type self-excited complex resonance series converter that uses the induced voltage of the auxiliary winding for ON / OFF driving of the switching element. The first and second control circuits for turning on / off the switching element from the induced voltage of the two auxiliary windings, and the third and fourth switching elements for turning off the first and second switching elements, respectively. A peak hold circuit for peak-holding the induced voltage generated in the first and second auxiliary windings, and the third and second when the induced voltage falls below a predetermined level from the hold voltage by the peak hold circuit. 4 switching elements are turned on and the first and second switching elements are turned off. Configure a that comparator.

  Therefore, the first and second control circuits indicate that the secondary-side induced current does not flow due to the completion of charging of the secondary-side smoothing capacitor, and one of the first and second auxiliary windings Detecting from a voltage drop, the ON side of the first and second switching elements is driven OFF, and detecting the other voltage increase to detect OFF of the first and second switching elements. The switching is continued by turning on the side that has been turned on. Thus, in the voltage feedback type self-excited complex resonance series converter, appropriate switching conditions can be realized with a simple configuration, while maintaining the low loss and low noise that are the original features, and a circuit that is significantly larger than the other excitation type. Simplification and cost reduction can be realized. Moreover, it can be adapted to higher frequency, smaller size and higher efficiency.

  In the switching power supply device of the present invention, the third and fourth switching elements are bipolar transistors.

  Therefore, the hold charge of the peak hold circuit can be discharged by the base current of the bipolar transistor, no additional discharge means is required, and even if the third and fourth switching elements are turned on for a moment. The minority carrier disappearance time can be used as an appropriate delay time, and can function as a delay circuit for adjusting the dead time.

  Furthermore, in the switching power supply device of the present invention, as described above, the voltage induced in the first and second auxiliary windings of the transformer is applied to the first and second switching elements via the gate resistance. The voltage induced in the first and second auxiliary windings of the transformer is also supplied to the peak hold circuit and the comparator of the first and second control circuits via the gate resistance. Given.

  Therefore, the rise of the gate voltage of the first and second switching elements can be moderated, which is suitable for forming a dead-off time in which both the first and second switching elements are turned off.

  In the switching power supply device of the present invention, as described above, the comparator includes a MOSFET.

  Therefore, it is possible to detect using the gate threshold voltage of the MOSFET that the induced voltage generated in the first and second auxiliary windings is lower than the hold voltage by the peak hold circuit by a predetermined level or more. it can.

Furthermore, the switching power supply apparatus of the present invention, as described above, the LED of the load.

  Therefore, current does not flow until a voltage higher than the forward voltage of the LED is applied, and it is in a so-called no-load state at the time of start-up, so that the start-up property of the voltage feedback type self-excited complex resonance series converter is improved. There is an effect of suppressing the rush current to the LED generated when the LED load is attached / detached.

  FIG. 1 is a block diagram showing an electrical configuration of a switching power supply device 11 according to an embodiment of the present invention. This switching power supply device 11 is an improved voltage feedback type self-excited composite resonance series converter, and maintains the low loss and noise reduction of the switching elements Q1 and Q2, which are the original features, compared to the separate excitation type. This greatly simplifies the circuit and lowers the cost, and further enables higher frequency operation.

  In FIG. 1, a series circuit of first and second switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of a DC input power source E, and a capacitor C0 is connected. . A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T1, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 , Q2 and a capacitor C2 are connected in parallel (FIG. 1 shows an example in which one end of the DC input power supply E is connected to the low voltage side and the capacitor C2 is connected in parallel to the power MOSFET Q2). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively. Capacitor C2 may be substituted by the junction capacitance of power MOSFETs Q1 and Q2, and diodes D1 and D2 may also be used as body diodes of power MOSFETs Q1 and Q2.

  Further, an intermediate tap is provided in the output winding of the output transformer T1 and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. Further, an auxiliary winding L12 is provided in the output transformer T1, and the reverse polarity side of the primary main winding L11 is connected to the gate of the power MOSFET Q2 via the gate resistor R12, and the power MOSFET Q2 is generated by the voltage generated in the auxiliary winding L12. Is configured to be driven. Similarly, for the gate drive of the power MOSFET Q1 on the high voltage side, the second auxiliary winding L13 is provided in the output transformer T1, and the same polarity side as the primary main winding L11 is connected to the gate of the power MOSFET Q1 via the gate resistor R11. And the power MOSFET Q1 can be driven by the voltage generated in the second auxiliary winding L13. Thus, the power MOSFETs Q1 and Q2 are alternately turned on and off by the feedback voltages from the two auxiliary windings L13 and L12, and self-oscillate.

Here, it should be noted that in the present invention, the power MOSFETs Q1 and Q2 are provided with control circuits Cont1 and Cont2 for setting the OFF timing thereof. Note that these control circuits Cont1 and Cont2 have the same configuration, and in order to simplify the drawing, only the control circuit Cont2 is shown in FIG. 1 as a specific configuration. These control circuits Cont1 and Cont2 have a switch element SW1 which is a third and fourth switching element for short-circuiting the gates and sources of the power MOSFETs Q1 and Q2 and turning off the power MOSFETs Q1 and Q2, and a diode D5. And a capacitor C4 for detecting the reverse voltage of the diode D5 and detecting the reverse voltage of the diode D5 and turning on the switch element SW1, and for discharging the charge of the capacitor C4 in preparation for the next cycle. Delay circuit 12 and switch SW2. These control circuits Cont1 and Cont2 perform the following self-excited operation.

  The circuit operation will be described with reference to FIGS. In the figure, VQ1 and VQ2 are drain-source voltages of the power MOSFETs Q1 and Q2, IQ1 and IQ2 are drain currents of the power MOSFETs Q1 and Q2, VC1 is a voltage of the capacitor C1, ID3 and ID4 are diode currents of the output rectifier diodes D3 and D4, VL11 is a voltage of the primary winding L11 of the output transformer T1, VL12 and VL13 are voltages of the auxiliary windings L12 and L13 of the output transformer T1, and cont1 and cont2 are for driving the switch element SW1 in the control circuits Cont1 and Cont2. Each of the signals is represented. However, FIG. 2 is not an operation waveform diagram of the self-excited composite resonance series converter of the present embodiment shown in FIG. 1, but an operation waveform diagram shown in FIG. 11 of the separately excited composite resonance series converter shown in FIG. This shows in detail the voltages VL11 and VL12 of the windings L11 and L12.

  Referring to FIG. 2, the voltage waveform applied to primary winding L11 of output transformer T1 is a waveform like VL11, and similar waveforms are generated in auxiliary windings L12 and L13. In the case of current feedback, a sinusoidal resonance current is fed back, whereas in the voltage feedback from the output transformer T1, a rectangular wave voltage such as VL11 is fed back, so that the power MOSFETs Q1 and Q2 are driven. It is understood that it is suitable. However, there is a step as indicated by reference symbol P in VL11 corresponding to a section in which the currents ID3 and ID4 of the secondary diodes D3 and D4 are interrupted. The present invention pays attention to the step P, and aims at the following effects by setting the timing of turning off the switching elements Q1 and Q2 when the step P is generated.

  That is, the period until the step P is generated depends on the series resonance frequency of the inductor L and the capacitor C1, and a stable operating frequency design is possible. Further, the generation of the step P is a result of the interruption of the diode currents ID3 and ID4 on the output side, and the recovery of the output diodes D3 and D4 is suppressed at the switching OFF operation point that is ideal in the separate excitation type. Is possible. Furthermore, an appropriate dead-off time can be generated by the rectifying / smoothing circuit used as the detecting means for detecting the step P, which can be used for realizing the ZVS operation. These effects will be described in detail below.

FIG. 3 is an operation waveform diagram of the voltage feedback self-excited composite resonance series converter according to the present embodiment. Referring to FIG. 3, VL1 2 represents the feedback voltage waveform generated in the auxiliary winding L1 2, VC4 denotes the voltage of the capacitor C4 for the peak hold capacitor C4 between the feedback voltage VL1 2 is positive The battery is charged and held up to a voltage lower by the ON voltage of the diode D5. VL1 2 moment the step occurs, VC4> VL1 2 state reverse voltage occurs the diode D5 is applied. At time t1, the comparator Comp detects this reverse voltage at a certain threshold δ, and the switch MOSFET SW2 connected between the gate and source of the power MOSFET Q2 is turned on by the output of the comparator Comp, so that the power MOSFET Q2 Turn off. Thereafter, after delaying by the delay circuit 12 by the time t3 in preparation for the next ON cycle, the charge of the capacitor C4 is discharged by the switch element SW2, so that VC4 is reset to zero.

Thus, the auxiliary winding voltage VL1 time the second step occurs in (the time t1) in the power MOSFET Q 2 gate - by ON the switching element SW1 provided between the source results in a OFF the power MOSFET Q 2, the auxiliary winding voltage VL1 2 is rapidly decreased from the time t2, the auxiliary winding L1 3 voltage VL1 3 rise through an appropriate dead-off time, to turn ON the power MOSFET Q 1 via the gate resistor R1 1, inductor L and capacitor A step is generated after a certain period determined by the resonance frequency with C1, and the same operation is repeated, so that the drain-source voltage VQ1, the drain current IQ1 of the power MOSFET Q1 and the drain-source of the power MOSFET Q2 can be obtained even in the self-excited type. Voltage VQ2, drain current IQ2, and diode De current ID3, ID4 is waveform closely approximates the separately excited shown in FIG. 11 that is obtained is understood. Thus, as described above, while maintaining low loss and low noise of the switching elements Q1 and Q2, the circuit can be greatly simplified and cost-reduced compared to the separately excited type, and the operation can be performed at a higher frequency (for example, 500 kHz) may be possible.

Hereinafter, specific examples of development of the present invention will be sequentially described. Parts corresponding to those in FIG. 1 are denoted by the same reference numerals. Control circuit Cont1a, Cont2a of the switching power supply device 11a in FIG. 4, the power MOSFETs Q1, Q2 of the gate in FIG. 1 - as Luz switch SW1 to short-circuit the source is a specific circuit using the transistor Q11 of the npn type. A reverse current blocking diode D11 is connected in series with the transistor Q11, and a capacitor C11 is connected between the base and emitter of the transistor Q11 to prevent malfunction of the transistor Q11 due to the drive signals cont1 and cont2 from the comparator Comp. Is connected.

  The operation of the switching power supply device 11a configured as described above is substantially the same as that of the switching power supply device 11 shown in FIG. Further, in this example, the capacitor C4 is discharged at the same time.

In contrast, the control circuit of the switching power supply device 11b of FIG. 5 Cont1b, Cont2b the gate of the power MOSFETs Q1, Q2 in FIG. 1 - as Luz switch SW1 to short-circuit the source, specifically using a transistor Q12 of the pnp Circuit. The reverse current blocking diode D11 is connected in series with the transistor Q12, and the capacitor for preventing malfunction of the transistor Q12 due to the drive signals cont1 and cont2 from the comparator Comp is connected between the base and emitter of the transistor Q12. C11 is connected.

  Since the switch SW1 is a pnp transistor Q12, the diode D5 and the capacitor C4 constituting the peak hold circuit are interchanged. Also by this, when the transistor Q12 serving as the switch SW1 is turned on by the charge of the capacitor C4 when the step is generated, the charge of the capacitor C4 can be discharged at the same time.

  6 is an example in which the voltages of the auxiliary windings L13 and L12 are applied to the control circuits Cont1 and Cont2 via the gate resistors R11 and R12, and the gate resistors R11 and R12 and the peak hold circuit are provided. The rise of the gate voltage of the power MOSFETs Q1 and Q2 can be moderated by the effect of the integrating circuit by the capacitor C4, which is suitable for forming a dead-off time in which both the power MOSFETs Q1 and Q2 are turned off. The main operation is substantially the same as in FIGS.

Switching power supply device 11d of FIG. 7, the control circuit Cont1d in FIG 6, in Cont2d, the gate of the power MOSFETs Q1, Q2 - as Luz switch SW1 to short-circuit the source, an example using a MOSFET Q 21, generally, the above-described FIG. 4 and FIG. 5, but the MOSFET is a voltage-driven type element. Therefore, the MOSFET Q21 is directly driven by the drive signals cont1 and cont2 from the comparator Comp, and the capacitor SW is used by the switch SW2. The charge of C4 is directly discharged.

Switching power supply apparatus 11e in FIG. 8, performs through the Gate drive power MOSFETs Q1, Q2 in FIG. 4, and the gate resistors R11, R12, control circuit Cont1e, a diode D5 in Cont2e, high frequency Accordingly, there is a possibility that the capacitor C4 can be substituted by the gate capacitance Ciss of the power MOSFETs Q1 and Q2.

FIG. 9 is a circuit diagram of control circuits Conta and Contb, which are other specific configuration examples of the control circuits Cont1 and Cont2 of FIG. In the control circuit Conta shown in FIG. 9A, the voltage feedback signal input between the terminals P11-P13; P21-P23 is divided by the resistors r1 and r2, and the divided voltage is -terminal of the comparator IC. To supply. The divided voltage charges the capacitor c2 via the diode d5 and the Zener diode zd, and supplies the charging voltage to the + terminal of the comparator IC. With such a configuration, when the feedback voltage drops below a predetermined value, the potential at the voltage dividing point also drops, the potential at the − terminal becomes lower than the potential at the + terminal, and the output of the comparator IC becomes high level. As a result, the MOSFET q1 is turned ON, the terminals P12-P13; P22-P23 are short-circuited, that is, the power MOSFETs Q1, Q2 can be turned OFF. In the figure, the diode d2 and the capacitor c1 form a power supply for the comparator IC, and the diodes d3 and d4 and the resistor r3 form a circuit for extracting the charge of the capacitor c2. Such a diode d5 and Zener diode zd increase the degree of freedom in setting the threshold value of the comparator IC.

  Further, in the control circuit Contb shown in FIG. 9B, the voltage feedback signal input between the terminals P11-P13; P21-P23 is accumulated in the capacitor c11 via the resistor r11 and the diode d11, and the feedback When the voltage drops below the gate threshold voltage of the MOSFET q11, the MOSFET q11 is turned on to release the charge of the capacitor c11 and the switch element q12 composed of a pnp bipolar transistor is turned on by the discharge current, Terminals P12-P13; P22-P23 are short-circuited, that is, the power MOSFETs Q1, Q2 are turned OFF. The correspondence between the terminals P11 to P13; P21 to P23 is shown in FIG.

  In the above description, the input power source has been described as the DC power source E, but a commercial power source may be rectified and smoothed. The secondary side of the output transformer T1 has been described as an example in which a center tap is provided to configure a rectifier circuit. Further, the capacitor C2 forming the composite resonance circuit is provided in at least one of the first and second switching elements Q1 and Q2. When the operating frequency is high, the parasitic capacitance (Crss) of the power MOSFETs Q1 and Q2 is used. ) Can be substituted.

  The load Load is preferably an LED. The current does not flow until a voltage higher than the forward voltage Vf of the LED (n times that in the case of the series configuration) is applied, and is in a so-called no-load state at start-up. This is because the startability of the resonant series converter is improved and the rush current to the LED generated when the LED load is attached / detached is suppressed.

1 is a block diagram showing an electrical configuration of a switching power supply device according to an embodiment of the present invention. It is a figure which shows in detail the operation | movement waveform of the conventional other excitation type | mold composite resonance series converter shown in FIG. FIG. 2 is an operation waveform diagram of the voltage feedback self-excited composite resonance series converter according to the embodiment of the present invention shown in FIG. 1. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a figure which shows the specific circuit example of the converter shown in FIG. It is a circuit diagram which shows the basic composition of a separately excited composite resonance series converter. FIG. 11 is an operation waveform diagram of FIG. 10. It is a circuit diagram which shows the example of the conventional self-excitation (current feedback) type | mold composite resonance series converter. It is a circuit diagram which shows the example of the conventional self-excitation (voltage feedback) type | mold composite resonance series converter. It is a circuit diagram which shows the other example of the conventional self-excitation (voltage feedback) type | mold composite resonance series converter. It is a block diagram which shows the specific example of the control circuit of the converter shown in FIG.

Explanation of symbols

11; 11a, 11b, 11c, 11d, 11e Switching power supply device 12 Delay circuit C0, C1, C2, C4, C11 Capacitor C3 Smoothing capacitor Comp Comparator Cont1, Cont2; Cont1a, Cont2a Control circuit Cont1b, Cont2b; Circuits Cont1e, Cont2e; Conta, Contb Control circuit D1, D2, D3, D4, D5, D11 Diode E DC input power supply L Inductor Load DC load Q1, Q2 Switching element (power MOSFET)
Q11, Q12 Transistor Q21 MOSFET
SW1, SW2 switch element T1 output transformer

Claims (6)

A series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and between the connection point of the first and second switching elements and one terminal of the DC input power supply, A series circuit composed of an inductor, a capacitor, and a primary winding of the transformer is connected, and the secondary induced current of the transformer obtained by switching the first and second switching elements is rectified and smoothed by a diode and a smoothing capacitor. And the first and second control circuits turn on and off the first and second switching elements with voltages induced in the first and second auxiliary windings of the transformer, respectively. In a switching power supply device comprising a voltage feedback type self-excited composite resonance series converter that is continued,
The first and second control circuits are:
Third and fourth switching elements for turning off the first and second switching elements, respectively;
A peak hold circuit for peak-holding an induced voltage generated in the first and second auxiliary windings;
A comparator that turns on the third and fourth switching elements and turns off the first and second switching elements, respectively, when the induced voltage is lowered by a predetermined level or more from the hold voltage by the peak hold circuit; A switching power supply device comprising:   2. The switching power supply device according to claim 1, wherein the third and fourth switching elements are bipolar transistors. The peak hold circuit comprises a series circuit of a diode and a capacitor for peak-holding an induced voltage in a direction to turn on the first and second switching elements generated in the first and second auxiliary windings,
The first and second control circuits further include switch means for discharging the charge of the capacitor after the comparator turns on the third and fourth switching elements. 3. The switching power supply device according to 1 or 2.   The first and second auxiliary windings of the transformer are connected to the first and second switching elements via a gate resistor, and the peak hold circuit and the comparator of the first and second control circuits The switching power supply device according to claim 1, wherein the switching power supply device is connected to the switching power supply device.   The switching power supply device according to claim 1, wherein the comparator includes a MOSFET. The switching power supply according to claim 1, wherein a load connected between terminals of the smoothing capacitor is an LED.

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