JP5042882B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply device that outputs a DC stabilized voltage / current that is insulated from an input power supply side, and more particularly, to a device that achieves a reduction in switching element loss and a reduction in size and cost.

  Conventionally, a composite resonance type series converter circuit is known as a small-sized and high-efficiency switching power supply as described above. FIG. 7 shows a main circuit configuration according to the prior art, and FIG. 8 shows waveforms of main parts.

  In this switching power supply device, a series circuit of switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of the DC input power supply E, and a capacitor C0 is connected. A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 Alternatively, a capacitor C2 is connected in parallel with either Q2 (FIG. 7 shows an example in which one end of the DC input power supply E is connected to the low voltage side and the capacitor C2 is connected in parallel to the power MOSFET Q2). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively (in many cases, they are also used as body diodes for power MOSFETs Q1 and Q2).

  Further, an intermediate tap is provided in the output winding of the output transformer T and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. The power MOSFETs Q1 and Q2 are alternately turned ON / OFF at a preset frequency in consideration of the complex resonance condition by the control unit 1 shown as a block. Therefore, the control unit 1 has a high-frequency oscillation function, a function of alternately driving the two power MOSFETs Q1 and Q2, and a function of setting a dead time period for turning off the two power MOSFETs Q1 and Q2. A feedforward, feedback control function and output variable function for controlling voltage, current, and power are provided.

  Referring to FIG. 8, Vg1 and Vg2 indicate drive signals for power MOSFETs Q1 and Q2 set in advance by control unit 1. A dead time period in which both are turned ON / OFF alternately and both are turned OFF is set. VQ1, IQ1 and VQ2, IQ2 indicate the drain-source voltage and drain current of the power MOSFETs Q1, Q2. When the drive signal Vg1 is High, the drain current IQ1 flows through the power MOSFET Q1, and when it is Low, the voltage VQ1 substantially equal to the DC input power source E is applied (the same applies to the power MOSFET Q2). In the dead time period, the drain-source voltages VQ1 and VQ2 have rising and falling waveforms with an arbitrary slope due to the effect of the capacitor C2, the inductor L, and the excitation inductance of the output transformer T. The drain currents IQ1 and IQ2 have a series resonance current waveform set approximately by the inductor L and the capacitor C1, and these combined currents are the series resonance of the inductor L, the primary winding L11 of the output transformer T, and the capacitor C1. It becomes the current of the circuit. VC1 represents a voltage waveform of the capacitor C1, and has a waveform delayed in phase from the current of the series resonance circuit. ID3 and ID4 indicate current waveforms of the output rectifier diodes D3 and D4. The relationship between the drive frequency of the power MOSFETs Q1 and Q2 and the series resonance frequency of the inductor L and the capacitor C1 is expressed as “resonance frequency> drive frequency. By satisfying this condition, it is possible to set the current of one of the diodes D3 and D4 to flow and then the other current starts to flow. During the period when both diode currents do not flow, power is transmitted to the output side. Not. That is, during the period when the currents of the diodes D3 and D4 do not flow, the secondary side of the output transformer T is considered to be unloaded, and the primary side exciting inductance L of the transformer T is inserted in series into the primary side series resonance circuit. As a result of the switching of the series resonance condition, inflection points are also found in the waveforms of the drain currents IQ1 and IQ2.

  In such a complex resonance type series converter, ZVS (zero voltage switching), that is, a condition setting such that a current starts to flow after the applied voltage of the switching elements Q1 and Q2 is reduced is possible, and switching loss is extremely small. Since the recovery loss of the secondary side rectifier diodes D3 and D4 can be avoided, high frequency can be achieved with high efficiency. Further, since the voltage / current waveform at the time of switching is stable and the ringing of the secondary side rectifier diodes D3 and D4 can be suppressed, the noise is also excellent.

  The above-described prior art is a so-called separately-excited switching power supply that drives the two switching elements Q1 and Q2 by setting the frequency in advance with an oscillator while having such various features. A level shifter is required for the potential side switching elements Q1 and Q2. From the viewpoint of frequency followability and loss, there is a technical problem with high frequency, and from a cost viewpoint, for example, Patent Documents 2 to 4 Self-excited studies are also being conducted as shown.

  FIG. 9 is an electric circuit diagram showing a publicly known example of a current feedback type self-excited composite resonance series converter, which is shown in Patent Document 2. Although the main circuit configuration is substantially the same as that in FIG. 7, a commercial power supply is used for full-wave rectification as the input power supply, and the secondary side of the output transformer T02 is a full-wave rectification circuit using a rectification bridge DB. The switching elements TR1 and TR2 are composed of bipolar transistors, and the driving is performed using the secondary windings LB1 and LB2 of the current feedback transformer T01, and the dead time is secured by the reverse bias means and the driving conditions are improved.

  On the other hand, FIG. 10 is an electric circuit diagram showing a known example of a voltage feedback type self-excited composite resonance series converter, which is shown in Patent Document 3. A feedback signal to the bipolar transistors TR1 and TR2 as switching elements is obtained by providing a feedback winding NB2 in the output transformer PIT, and feedback control for load variation countermeasures and output stabilization is added.

  The conventional examples of the self-excited composite resonant series converters shown in FIGS. 8 and 9 suggest the possibility of simplifying the drive circuit. However, the number of separately excited composite resonant series converters described with reference to FIG. It is thought that it is difficult to realize the same feature from the following points. First, it is the time constant and delay element of the feedback circuit that determines the driving frequency, and it seems difficult to maintain the optimum switching conditions such as ZVS against fluctuations in external factors such as load fluctuations. Next, a bipolar transistor is assumed as a switching element, which is not suitable for high frequency operation.

FIG. 11 is a block diagram showing a known example of a voltage feedback type self-excited separately excited composite resonance series converter using a power MOSFET, which is shown in Patent Document 4. In this prior art, power MOSFETs Q1 and Q2 are used as switching elements, and the power source of the main control circuit 4 and the sub control circuit 5 is secured from the auxiliary winding L12 of the output transformer T ′, and this winding voltage is used as a signal source. The main control circuit 4 controls ON / OFF of the low-voltage side power MOSFET Q2 so that the supply voltage to the load Load becomes constant, and the sub-control circuit 5 reduces the voltage between the terminals of the high-voltage side power MOSFET Q1 below the reference voltage. Is turned on to maintain the ZVS of the high-voltage power MOSFET Q1.
Japanese Patent No. 2734296 Japanese Patent No. 3371595 JP 2002-262568 A JP 2006-129548 A

  In the above-described prior art, a specific example as shown in FIG. 12 is shown in the sub-control circuit 5, the drain-source voltage of the power MOSFET Q 1 is divided by the resistors r 01 and r 02, and the reference voltage is output by the comparator 7. Compared with FIG. 8, the power MOSFET Q <b> 1 is turned on by charging and discharging the capacitor c <b> 1 of the integration circuit. In such a configuration, the ON timing is determined while directly determining the drain-source voltage of the power MOSFET Q1 on the high voltage side. Therefore, the ZVS may be realized, but the following problems are encountered. Conceivable. First, it is necessary to divide the drain-source voltage of the power MOSFET Q1, and there is a concern about the loss due to the resistors r01 and r02. On the other hand, when these resistors r01 and r02 are configured with high resistance, a delay time due to the input capacitance of the comparator 7 is assumed, and it is considered difficult to increase the frequency. Next, the drive control circuits 4 and 5 become complicated, and no superiority over the separate excitation system is recognized.

  The object of the present invention is to greatly simplify the circuit and reduce the cost of a self-excited composite resonant series converter of voltage feedback type compared to other types of excitation while maintaining the original characteristics of low loss and low noise. It is to provide a switching power supply device that can be realized.

  In the switching power supply device of the present invention, a series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and the connection point between the first and second switching elements and the DC input power supply A series circuit including an inductor, a capacitor, and a primary winding of a transformer is connected between one terminal and a secondary side induced current of the transformer obtained by switching of the first and second switching elements. Rectified and smoothed by a smoothing capacitor and output, and the first and second switching elements are respectively supplied with voltages induced in the first and second auxiliary windings of the transformer by the first and second control circuits. A switch consisting of a voltage feedback self-excited complex resonance series converter that is continuously switched on and off. In the power supply apparatus, the first control circuit includes a third switching element for turning off the first switching element, a peak hold circuit for peak-holding an induced voltage generated in the first auxiliary winding, And a first comparator that turns on the third switching element and turns off the first switching element when the induced voltage drops by a predetermined level or more from the hold voltage by the peak hold circuit. The second control circuit forward biases the fourth switching element for turning off the second switching element and the second switching element corresponding to the voltage induced in the second auxiliary winding. A triangular wave generation circuit that generates a triangular wave starting from the point in time when it is generated in a direction to generate, and a reference voltage source that generates a variable reference voltage And a second comparator that compares the triangular wave voltage with a reference voltage and turns on the corresponding fourth switching element when the triangular wave voltage becomes higher than the reference voltage. It is characterized by comprising.

  According to said structure, the series circuit which consists of a 1st and 2nd switching element is connected between both terminals of DC input power supply, and the connection point of said 1st and 2nd switching element and one of said DC input power supplies A series circuit comprising an inductor, a capacitor, and a primary winding of a transformer is connected between the first and second terminals, and voltage feedback using the induced voltage of the first and second auxiliary windings for ON / OFF driving of the switching element. Switching power supply comprising a self-excited composite resonance series converter of the type, the first control circuit among the first and second control circuits for turning on / off the switching element from the induced voltage of the auxiliary winding, A third switching element for turning off the first switching element and a peak for holding an induced voltage generated in the first auxiliary winding. A hold circuit; and a first comparator that turns on the third switching element and turns off the first switching element when the induced voltage is lower than a predetermined level by a hold voltage by the peak hold circuit. The second control circuit includes a fourth switching element for turning off the second switching element, and a second switching circuit corresponding to a voltage induced in the second auxiliary winding. A triangular wave generating circuit that generates a triangular wave starting from the time when the element is generated in the forward bias direction, a reference voltage source that generates a variable reference voltage, the voltage of the triangular wave and the reference voltage are compared, and the voltage of the triangular wave And a second comparator that turns on the corresponding fourth switching element when becomes higher than a reference voltage. It is formed.

  Therefore, the first control circuit detects from the voltage drop of the first auxiliary winding that the secondary-side induced current stops flowing due to the completion of charging of the secondary-side smoothing capacitor, and detects the third switching element. The first switching element that has been turned on is turned off, and the second switching element that has been turned off is turned on by a forward induced voltage generated in the second auxiliary winding (ZVS operation). . On the other hand, the second control circuit can add an output variable function or an output stabilization function by changing the reference power supply. When the fourth switching element is turned on by the output adjustment circuit and the second switching element that has been turned on is turned off, a forward induced voltage is generated in the first auxiliary winding, and the first switching element is turned on. 1 switching element is turned on, and a stable switching operation is continued by repeating these operations.

  With such a configuration, an appropriate switching condition (ZVS operation) can be realized with a simple configuration in a self-excited composite resonant series converter of voltage feedback type, while maintaining low loss and low noise, which are the original features, Compared with the separate excitation type, the circuit can be greatly simplified and the cost can be reduced, and an output variable function or an output stabilization function can be added. Moreover, it can be adapted to higher frequency, smaller size and higher efficiency.

  Also, the switching power supply device of the present invention feeds back the detection result of the load detection circuit to the primary side, and changes the reference voltage of the reference voltage source. And a feedback circuit.

  According to the above configuration, feedback control can be performed so that the load current or output voltage on the secondary side becomes a constant current or a constant voltage.

  Furthermore, in the switching power supply device of the present invention, the peak hold circuit is a series circuit of a diode and a capacitor for peak-holding an induced voltage in the direction of turning on the first switching element generated in the first auxiliary winding. The plurality of diodes are connected in series, or Zener diodes are connected in series.

  According to said structure, the freedom degree of the threshold value setting of a said 1st comparator can be raised with the stage number of the said diode, or the threshold voltage of a Zener diode.

  The switching power supply device of the present invention is characterized in that a reverse bias circuit is provided on the output side of the second comparator.

  According to said structure, the malfunctioning and semiconductive state of a 4th switching element can be avoided, and operation | movement can be stabilized.

Furthermore, in the switching power supply device of the present invention, the load connected between the terminals of the smoothing capacitor is an LED.

  According to the above configuration, the current does not flow until a voltage equal to or higher than the forward voltage of the LED is applied, and is in a so-called no-load state at the time of startup. As well as improving the above, there is an effect of suppressing the rush current to the LED generated when the LED load is attached / detached.

  Preferably, the reference voltage source includes a dimmer, and changes the reference voltage in accordance with the dimming amount.

  As described above, the switching power supply according to the present invention is a switching power supply comprising a voltage feedback type self-excited complex resonance series converter that uses the induced voltage of the auxiliary winding for ON / OFF driving of the switching element. Among the first and second control circuits that perform ON / OFF driving of the switching element from the induced voltage of the two auxiliary windings, the first control circuit is configured to turn on the first switching element. A switching element, a peak hold circuit for peak-holding an induced voltage generated in the first auxiliary winding, and the third switching element when the induced voltage is lowered by a predetermined level or more from a hold voltage by the peak-hold circuit. And a first comparator for turning off the first switching element and turning on the first switching element. The control circuit is generated in a direction in which the fourth switching element for turning off the second switching element and the voltage induced in the second auxiliary winding forward bias the corresponding second switching element. The triangular wave generation circuit that generates a triangular wave starting from the point of time, the reference voltage source that generates a variable reference voltage, the voltage of the triangular wave and the reference voltage are compared, and the voltage of the triangular wave becomes higher than the reference voltage. And an output adjusting circuit including a second comparator for turning on the fourth switching element.

  Therefore, in a voltage feedback self-excited composite resonant series converter, appropriate switching conditions (ZVS operation) can be realized with a simple configuration, while maintaining the original characteristics of low loss and low noise while being separately excited. In comparison with this, circuit simplification and cost reduction can be realized, and an output variable function or output stabilization function can be added. Moreover, it can be adapted to higher frequency, smaller size and higher efficiency.

  In addition, as described above, the switching power supply device of the present invention feeds back the detection result of the load detection circuit that detects the secondary side load current or voltage to the primary side and the reference voltage source. And a feedback circuit for changing the reference voltage.

  Therefore, feedback control can be performed so that the load current or output voltage on the secondary side becomes a constant current or a constant voltage.

  Furthermore, as described above, the switching power supply device of the present invention includes a diode and a capacitor for peak-holding the induced voltage in the direction of turning on the first switching element generated in the first auxiliary winding by the peak hold circuit. A plurality of the diodes are connected in series, or Zener diodes are connected in series to the diodes.

  Therefore, the degree of freedom in setting the threshold value of the first comparator can be increased by the number of stages of the diodes and the Zener diode threshold voltage.

  Further, as described above, the switching power supply device of the present invention is provided with the reverse bias circuit on the output side of the second comparator.

  Therefore, it is possible to avoid a malfunction or a semiconducting state of the fourth switching element and to stabilize the operation.

Furthermore, the switching power supply apparatus of the present invention, as described above, the LED of the load.

  Therefore, current does not flow until a voltage higher than the forward voltage of the LED is applied, and it is in a so-called no-load state at the time of start-up, so that the start-up property of the voltage feedback type self-excited complex resonance series converter is improved. There is an effect of suppressing the rush current to the LED generated when the LED load is attached / detached.

  FIG. 1 is a block diagram showing an electrical configuration of a switching power supply device 11 according to an embodiment of the present invention. This switching power supply device 11 is an improved voltage feedback type self-excited composite resonance series converter, and maintains the low loss and noise reduction of the switching elements Q1 and Q2, which are the original features, compared to the separate excitation type. This greatly simplifies the circuit and lowers the cost, and further enables higher frequency operation.

  In FIG. 1, a series circuit of first and second switching elements Q1 and Q2 (hereinafter referred to as power MOSFET) is connected between both terminals of a DC input power source E, and a capacitor C0 is connected. . A series resonant circuit of an inductor L, a primary winding L11 of the output transformer T1, and a capacitor C1 is formed between the connection point of the power MOSFETs Q1 and Q2 and one end of the DC input power supply E, and the power MOSFET Q1 , Q2 and a capacitor C2 are connected in parallel (FIG. 1 shows an example in which one end of the DC input power supply E is connected to the low voltage side and the capacitor C2 is connected in parallel to the power MOSFET Q2). Also, diodes D1 and D2 are connected in antiparallel to power MOSFETs Q1 and Q2, respectively. Capacitor C2 may be substituted by the junction capacitance of power MOSFETs Q1 and Q2, and diodes D1 and D2 may also be used as body diodes of power MOSFETs Q1 and Q2.

  Further, an intermediate tap is provided in the output winding of the output transformer T1 and divided into two (L21, L22), and a full-wave rectifier circuit is formed by diodes D3 and D4 that rectify their outputs, and between the intermediate taps. A smoothing capacitor C3 and a DC load Load are connected. Further, an auxiliary winding L12 is provided in the output transformer T1, and the opposite polarity side of the primary main winding L11 is connected to the gate of the power MOSFET Q2 via the gate resistor R2, and the power MOSFET Q2 is generated by the voltage generated in the auxiliary winding L12. Is configured to be driven. Similarly, for the gate drive of the high-voltage side power MOSFET Q1, a second auxiliary winding L13 is provided in the output transformer T1, the same polarity side as the primary main winding L11, the gate resistor R1, the capacitor C4, and the Zener diode. The power MOSFET Q1 is connected to the gate of the power MOSFET Q1 through a parallel circuit composed of ZD so that the power MOSFET Q1 can be driven by the voltage generated in the second auxiliary winding L13. A starting resistor R4 is connected between the gate of the power MOSFET Q1 and the other end (high voltage side) of the DC input power source E. Thus, the power MOSFETs Q1 and Q2 are alternately turned on and off by the feedback voltages from the two auxiliary windings L13 and L12, and self-oscillate.

  Here, it should be noted that in this configuration, the power MOSFETs Q1 and Q2 are provided with control circuits Cont1 and Cont2 for setting the OFF timing thereof. First, the control circuit Cont1 short-circuits between the gate and the source of the power MOSFET Q1, and the switch element q1 and the diode d1, which are third switching elements for turning off the power MOSFET Q1, and the voltage dividing resistors r1, r2, diodes a peak hold circuit composed of d5, a Zener diode zd and a capacitor c2, a first comparator Comp1 for detecting the reverse voltage of the diode d5 and the Zener diode zd and turning on the switch element q1, and in the next cycle And diodes d3 and d4 and a resistor r3 for discharging the charge of the capacitor c2, and a diode d2 and a capacitor c1 for creating a control power source.

  On the other hand, the control circuit Cont2 includes a triangular wave generation circuit 23 that generates a triangular wave starting from the time when the voltage induced in the auxiliary winding L12 is generated in the direction in which the corresponding switching element Q2 is forward-biased, and a variable reference voltage VTH. The reference voltage source 24 for generating the reference voltage, the voltage of the triangular wave and the reference voltage VTH are compared, and when the voltage of the triangular wave becomes higher than the reference voltage VTH, the comparison circuit 25 that short-circuits between the gate and the source of the corresponding switching element Q2. And an output adjustment circuit. In the output adjustment circuit 22, a control power supply Vc is created from the non-ground side terminal of the auxiliary winding L12 of the output transformer T1 by the diode Dc1 and the capacitor Cc3. These control circuits Cont1 and Cont2 perform the following self-excited operation.

  The circuit operation of the control circuit Cont1 will be described based on FIGS. In the figure, VQ1 and VQ2 are drain-source voltages of the power MOSFETs Q1 and Q2, IQ1 and IQ2 are drain currents of the power MOSFETs Q1 and Q2, VC1 is a voltage of the capacitor C1, ID3 and ID4 are diode currents of the output rectifier diodes D3 and D4, VL11 is the voltage of the primary winding L11 of the output transformer T1, VL12 and VL13 are the voltages of the auxiliary windings L12 and L13 of the output transformer T1, and cont1 and cont2 drive the switch elements q1 and Qc5 in the control circuits Cont1 and Cont2. The signals for performing are shown respectively. However, FIG. 2 is not an operation waveform diagram of the self-excited complex resonance series converter of the present embodiment shown in FIG. 1, but an operation waveform diagram of the separately excited complex resonance series converter shown in FIG. This shows in detail the voltages VL11 and VL12 of the windings L11 and L12.

  Referring to FIG. 2, the voltage waveform applied to primary winding L11 of output transformer T1 is a waveform like VL11, and similar waveforms are generated in auxiliary windings L12 and L13. In the case of current feedback, a sinusoidal resonance current is fed back, whereas in the voltage feedback from the output transformer T1, a rectangular wave voltage such as VL11 is fed back, so that the power MOSFETs Q1 and Q2 are driven. It is understood that it is suitable. However, there is a step as indicated by reference symbol P in VL11 corresponding to a section in which the currents ID3 and ID4 of the secondary diodes D3 and D4 are interrupted. This configuration pays attention to the step P, and aims at the following effects by setting the timing of turning off the switching element Q1 when the step P is generated.

  That is, the period until the step P is generated depends on the series resonance frequency of the inductor L and the capacitor C1, and a stable operating frequency design is possible. Further, the generation of the step P is a result of the interruption of the diode current ID4 on the output side, and at the switching OFF operation point that is ideal in the separate excitation type, the recovery of the output diode D4 is suppressed and the switching element Q1 ZVS operation is possible. Furthermore, an appropriate dead-off time can be generated by the rectifying / smoothing circuit used as the detecting means for detecting the step P, which can be used for realizing the ZVS operation. These effects will be described in detail below.

  FIG. 3 is an operation waveform diagram of the voltage feedback type self-excited complex resonance series converter according to the present configuration. Referring to FIG. 3, VL13 indicates a feedback voltage waveform generated in the auxiliary winding L13, Vc2 indicates the voltage of the peak hold capacitor c2, and the capacitor c2 is connected to the diode d5 while the feedback voltage VL13 is positive. The voltage is charged and held to a voltage lower by the ON voltage and the Zener voltage of the Zener diode zd. At the moment when the step is generated in VL13, a state of Vc2> VL13 occurs, and a reverse voltage is applied to the diode d5 and the Zener diode zd. At time t1, the comparator Comp1 detects this reverse voltage with a certain threshold value δ, and the switch MOSFET q1 connected between the gate and the source of the power MOSFET Q1 is turned on by the output cont1 of the comparator Comp1, whereby the power MOSFET Q1 Turns off. Thereafter, in preparation for the next ON cycle, after delaying by the time t3 when the feedback voltage VL13 becomes negative, the charge of the capacitor c2 is discharged by the diode d4, the resistor r3, and the diode d3, so that Vc2 is reset to zero. The resistor r3 is a discharge resistor, and the diode d3 is a diode for preventing reverse charging of the capacitor c2.

  Thus, at the time when a step occurs in the auxiliary winding voltage VL13 (the time t1), the switch element q1 provided between the gate and the source of the power MOSFET Q1 is turned on and the power MOSFET Q1 is turned off. As a result, the auxiliary winding voltage VL13 is The voltage VL12 of the auxiliary winding L12 rises rapidly after time t2 and passes through an appropriate dead-off time, and the power MOSFET Q2 is turned on via the gate resistor R2. When the power MOSFET Q2 is turned off after an arbitrary ON time by a second control circuit Cont2, which is an output adjusting circuit described later, a forward induced voltage is generated in the auxiliary winding L13, and the power MOSFET Q1 is turned on. A similar switching operation continues. As a result, the power MOSFET Q1 side performs a switching operation at an optimal timing and maintains a low loss, while performing the PWM control that can arbitrarily set the ON time on the power MOSFET Q2 side. It can be understood that the drain-source voltage VQ1, the drain current IQ1, and the diode current ID4 of the power MOSFET Q1 can obtain waveforms that are very close to the separately excited type shown in FIG. As a result, while maintaining the low loss and low noise of the switching element Q1, it is possible to greatly simplify the circuit and reduce the cost as compared with the separately excited type, and to enable a higher frequency of operation (for example, 500 kHz). Can do.

  On the other hand, FIGS. 4 and 5 are waveform diagrams for explaining the operation of the second control circuit Cont2. In the triangular wave generating circuit 23, a voltage dividing circuit is formed by resistors Rc1 and Rc2 from the non-ground side terminal of the auxiliary winding L12 of the output transformer T1, and the connection point is connected to the gate terminal of the FET Qc1. A capacitor Cc2 in parallel with the resistor Rc1 is used for speeding up the polarity reversal. Therefore, as shown in FIG. 4, when the voltage VL12 of the auxiliary winding L12 is generated in the direction in which the gate of the power MOSFET Q2 is forward-biased, the FET Qc1 is represented by the reference sign VQC1 (drain-source voltage). The transistor Qc2 forming the inverting circuit is turned off as indicated by the reference symbol VQC2.

  On the other hand, transistors Qc3 and Qc4 are mirrored from the control power supply Vc, and a circuit for supplying a reference current to the resistor Rc4 connected to the diode-structured transistor Qc3 and charging the capacitor Cc1 connected to the opposite transistor Qc4 is provided. At the same time, the transistor Qc2 is connected between the terminals of the capacitor Cc1 as a discharge circuit. Therefore, when the voltage VL12 of the auxiliary winding L12 is generated in the direction to reverse-bias the gate of the power MOSFET Q2, that is, when the transistor Qc2 is ON, charging by the current mirror circuit is not performed and the auxiliary winding When the voltage VL12 of L12 is generated in a direction in which the gate of the power MOSFET Q2 is forward-biased, that is, when the transistor Qc2 is turned off, charging is performed, and the terminal voltage of the capacitor Cc1 has a triangular waveform as indicated by reference numeral VCC1. To rise.

  On the other hand, the reference voltage source 24 is connected to the parallel circuit of the variable resistor VR and the phototransistor TR10 of the photocoupler PC from the control power source Vc via the resistor Rc5. The divided voltage value VTH of the parallel circuit and the capacitor A comparator comp2 for comparing the triangular wave voltage of Cc1 is provided. Cc4 is a noise prevention or delay capacitor.

  The comparator comp2, when the triangular wave voltage of the capacitor Cc1 connected to the positive input terminal exceeds the divided voltage VTH of the negative input terminal, the output of the comparator comp2 is at a high level as indicated by the reference symbol VCP. As described above, the FET Qc5, which is the fourth switching element, is turned on, the gate and the source of the power MOSFET Q2 are short-circuited via the diode Dc2, and the power MOSFET Q2 is turned off. By changing the resistance value of the variable resistor VR and changing the reference voltage VTH of the comparator Comp2 in the range of ΔVTH, the turn-off timing of the power MOSFET Q2 is shown in FIG. 4 (VTH = VTH1). The control circuit Cont2 After the high level timing (full duty), the speed can be increased as shown in FIG. 5 (VTH = VTH2).

  Corresponding to such a second control circuit Cont2, a load detection circuit 33 is provided on the secondary side for creating the feedback signal described above. In the load detection circuit 33, a series circuit of resistors R21 and R22 and a series circuit of resistors R23 and R24 are connected in parallel with the output smoothing capacitor C3, and a current detection resistor R25 is provided between the two series circuits. By inputting the divided voltage of the resistors R21 and R22 to the negative input terminal of the comparator Comp3 and inputting the voltage at the connection point of the resistors R23 and R24 to the positive input terminal, the voltage drop due to the current detection resistor R25 is amplified. Then, the photocoupler PC is driven via the transistor Q5 provided at the output of the comparator Comp3, and the luminance of the light emitting diode D10 of the photocoupler PC increases as the load current increases. The potential at the connection point of the resistors R23 and R24 also varies depending on the output voltage of the capacitor C3, and the luminance of the light emitting diode D10 of the photocoupler PC increases as the output voltage increases.

  Therefore, as the luminance of the light emitting diode D10 increases, that is, as the load current increases or the output voltage increases when there is no load, the reference voltage VTH decreases, and the triangular wave voltage generated in the capacitor Cc1 decreases. At that time, the output of the comparator Comp2 becomes high level, the FET Qc5 is turned on, and as a result, the ON duty of the power MOSFET Q2 is reduced and the output can be reduced. Thus, feedback control is performed via the photocoupler PC so that the voltage drop of the current detection resistor R25 becomes a constant value and the output voltage also becomes a substantially constant range. The output can be adjusted by changing the ON width.

  With this configuration, the first control circuit Cont1 has a voltage drop in the first auxiliary winding L13 corresponding to the fact that the secondary-side induced current stops flowing due to the completion of charging of the secondary-side smoothing capacitor C3. , The third switching element is turned on, the first switching element Q1 that has been turned on is turned off, and the first induced voltage generated in the second auxiliary winding L12 is turned off by the forward induced voltage. Since the switching is continued by turning on the switching element Q2 of No. 2, an appropriate switching condition (ZVS operation) can be realized with a simple configuration in the voltage feedback self-excited composite resonance series converter, which is an original feature. While maintaining low loss and low noise, the circuit can be greatly simplified and the cost can be reduced compared to the separate excitation type. Moreover, it can be adapted to higher frequency, smaller size and higher efficiency.

  Further, the second control circuit Cont2 is a triangular wave that generates a triangular wave starting from the time when the voltage induced in the second auxiliary winding L12 is generated in the direction of forward biasing the corresponding second switching element Q2. The generation circuit 23, the reference voltage source 24 for generating the variable reference voltage VTH, the voltage of the triangular wave and the reference voltage VTH are compared, and when the voltage of the triangular wave becomes higher than the reference voltage VTH, the switching element Q2 Since the output adjustment circuit includes the comparison circuit 25 that short-circuits between the gate and the source, an output variable function or an output stabilization function can be added by changing the reference voltage VTH. Thus, in the voltage feedback type converter, it is possible to realize an output variable function or an output stabilization function while maintaining an appropriate switching condition with a simple configuration.

  Further, the load detection circuit 33 for detecting the secondary side load current is further provided with a photocoupler PC which is a feedback circuit for feeding back the detection result to the primary side and changing the reference voltage VTH of the reference voltage source 24. Feedback control can be performed so that the load current on the secondary side becomes a constant current.

  Furthermore, in the peak hold circuit of the first control circuit Cont1, the Zener diode zd is provided in series with the diode d5 that supplies the charging current to the capacitor c2. Therefore, the first comparator is controlled by the threshold voltage of the Zener diode zd. The degree of freedom for setting the threshold value of comp1 can be increased. Note that the same effect can be obtained by connecting a plurality of diodes d5 in series. In this case, the threshold voltage can be adjusted by the number of stages of the diodes d5.

  In the comparison circuit 25 of the second control circuit Cont2, a reverse bias circuit including a capacitor Cc6, resistors Rc6 and Rc7, a diode Dc3, and a Zener diode ZD1 is provided on the output side of the second comparator comp2. . As a result, a reverse bias voltage is obtained with respect to the gate of the FET Qc5 in the capacitor Cc6, and the malfunction of the FET Qc5 due to noise or the like can be avoided while the output of the comparator comp2 is at a low level, thereby stabilizing the switching operation. Can be made. The reverse bias voltage generated in the capacitor Cc6 can be adjusted by the Zener voltage of the Zener diode ZD1 provided in parallel.

  Furthermore, the induced voltage of the auxiliary winding L13 is applied to the power MOSFET Q1 via a startup compensation circuit including a resistor R1 and a parallel circuit of a capacitor C4 and a Zener diode ZD. In this case, by setting the capacitance value of the capacitor C4 to be sufficiently larger than the capacitance value of the resonance capacitor C1, an additional startup resistor for charging the resonance capacitor C1 is not required for the startup. At this time, the weak feedback signal generated in the auxiliary winding L13 can be raised by the charge voltage of the capacitor C4, so that the starting can be facilitated.

  The Zener diode ZD has an effect of suppressing an overvoltage to the power MOSFET Q1 at the start-up and preventing an influence on the switching operation due to the AC impedance of the capacitor C4 in the switching operation after the start-up. Further, after starting, the DC potential of the capacitor C4 is discharged every cycle by the operation of the switching element q1, so that the switching operation is not affected.

  Here, the feedback is not limited to the load current / voltage as described above, but the load current detection circuit 33a in FIG. 6A and the load current detection circuit 33b in FIG. Either one is acceptable. By feedback, the load voltage detection circuit 33a in FIG. 6A can perform constant voltage control, and the load current detection circuit 33b in FIG. 6B can perform constant current control.

  In the above description, the input power source has been described as the DC power source E, but a commercial power source may be rectified and smoothed. The secondary side of the output transformer T1 has been described as an example in which a center tap is provided to configure a rectifier circuit. Further, the capacitor C2 forming the composite resonance circuit is provided in at least one of the first and second switching elements Q1 and Q2. When the operating frequency is high, the parasitic capacitance (Crss) of the power MOSFETs Q1 and Q2 is used. ) Can be substituted.

It is a block diagram which shows the electric constitution of the switching power supply device of one Embodiment of this invention. It is a figure which shows in detail the operation | movement waveform of the conventional other excitation type | mold composite resonance series converter shown in FIG. FIG. 2 is an operation waveform diagram of the voltage feedback type self-excited composite resonance series converter according to the configuration shown in FIG. 1. FIG. 2 is an operation waveform diagram of PWM in the voltage feedback type self-excited complex resonance series converter shown in FIG. 1. FIG. 2 is an operation waveform diagram of PWM in the voltage feedback type self-excited complex resonance series converter shown in FIG. 1. It is a block diagram which shows the structure of the secondary side in the switching power supply device of the other embodiment of this invention. It is a circuit diagram which shows the basic composition of a separately excited composite resonance series converter. FIG. 8 is an operation waveform diagram of FIG. 7. It is a circuit diagram which shows the example of the conventional self-excitation (current feedback) type | mold composite resonance series converter. It is a circuit diagram which shows the example of the conventional self-excitation (voltage feedback) type | mold composite resonance series converter. It is a circuit diagram which shows the other example of the conventional self-excitation (voltage feedback) type | mold composite resonance series converter. It is a block diagram which shows the specific example of the control circuit of the converter shown in FIG.

Explanation of symbols

11 switching power supply device 23 triangular wave generation circuit 24 reference voltage source 25 comparison circuit 33 load detection circuit C3 smoothing capacitor C0, C1, C2 capacitor Comp1, Comp2 comparator Comp3 comparator Cont1, Cont2 control circuit D1, D2, D3, D4 diode E DC Input power supply L Inductor Load DC load PC Photocoupler Q1, Q2 Switching element (Power MOSFET)
R25 Current detection resistor T1 Output transformer

Claims (6)

A series circuit composed of first and second switching elements is connected between both terminals of the DC input power supply, and between the connection point of the first and second switching elements and one terminal of the DC input power supply, A series circuit composed of an inductor, a capacitor, and a primary winding of the transformer is connected, and the secondary induced current of the transformer obtained by switching the first and second switching elements is rectified and smoothed by a diode and a smoothing capacitor. And the first and second control circuits turn on and off the first and second switching elements with voltages induced in the first and second auxiliary windings of the transformer, respectively. In a switching power supply device comprising a voltage feedback type self-excited composite resonance series converter that is continued,
The first control circuit includes:
A third switching element for turning off the first switching element;
A peak hold circuit for peak-holding an induced voltage generated in the first auxiliary winding;
And a first comparator that turns on the third switching element and turns off the first switching element when the induced voltage drops by a predetermined level or more from the hold voltage by the peak hold circuit. ,
The second control circuit includes:
A fourth switching element for turning off the second switching element;
A triangular wave generating circuit for generating a triangular wave starting from a point in time when a voltage induced in the second auxiliary winding is generated in a direction in which the corresponding second switching element is forward-biased;
A reference voltage source for generating a variable reference voltage;
It comprises an output adjustment circuit comprising a second comparator that compares the triangular wave voltage with a reference voltage and turns on the corresponding fourth switching element when the triangular wave voltage becomes higher than the reference voltage. The switching power supply device characterized by the above-mentioned. A load detection circuit for detecting a secondary side load current or voltage;
2. The switching power supply device according to claim 1, further comprising a feedback circuit that feeds back a detection result of the load detection circuit to a primary side and changes a reference voltage of the reference voltage source.   The peak hold circuit includes a series circuit of a diode and a capacitor for peak-holding an induced voltage in the direction of turning on the first switching element generated in the first auxiliary winding, and a plurality of the diodes are connected in series. 3. The switching power supply device according to claim 1, wherein the switching power supply devices are connected or Zener diodes are connected in series.   4. The switching power supply device according to claim 1, further comprising a reverse bias circuit on an output side of the second comparator. 5. The switching power supply according to any one of claims 1 to 4, wherein a load connected between terminals of the smoothing capacitor is an LED.   6. The switching power supply device according to claim 5, wherein the reference voltage source includes a dimmer and changes the reference voltage in accordance with the dimming amount.

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