JPH02261058A - Power conversion device - Google Patents

Abstract

PURPOSE:To permit the reduction of the pulsation of an output current remarkably by a method wherein the title device is provided with a PWM converter, a PWM inverter and the like and an AC load, driven by the sum of the output voltage of a cyclo converter and the output voltage of an output transformer, is impressed. CONSTITUTION:The title device is provided with a pulse width modulation control converter (PWM converter) CONV, a pulse width modulation control inverter (PWM inverter) INV, an output transformer OTR, a cyclo converter CC and the like. The PWM converter CONV controls an input current, supplied from an AC power source, or that a DC voltage, impressed on a DC smoothing capacitor Cd, becomes constant substantially. The PWM inverter INV employs the smoothing capacitor Cd as a DC power source and outputs a variable voltage and variable frequency power through the output transformer OTR. The cyclo converter CC is connected directly to an AC load and the sum of the output voltage of the cyclo converter and the output voltage of the output transformer OTR of the PWM inverter INV is impressed on the AC load. According to this method, the pulsation of the output current of the power converter may be reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to a large-capacity power conversion device that supplies variable voltage, variable frequency power to a load such as an AC motor.

(Prior Art) In recent years, large-capacity self-extinguishing elements (for example, gate turn-off thyristors: GTOs, etc.) have been actively developed and are being used in power conversion devices such as inverters. In particular, pulse width modulation controlled inverters have come to be widely used as drive power sources for AC motors because they can provide a positive arc wave output with variable voltage and variable frequency.

Furthermore, as the capacity of AC motors increases, inverter outputs need to have high voltage and large current, and high-capacity inverters using series-parallel connections of elements or multiple-connected inverters using output transformers have been proposed.

FIG. 8 shows the configuration of a conventional power conversion device, in which multiple connections are made using output transformers.

In the figure, Vd is a DC voltage source, INV-1 and INV-2 are first and second inverters with output transformers, TRU
Engineering.

TRυ2 represents an output transformer, INV-3 represents an output 1 to direct-coupled inverter without a lance, and Lυ+Run vcu represents the inductance, resistance, and back electromotive force of the load, respectively.

The figure shows one output phase (U phase), and the U phase of the first inverter INV-1 is equipped with a self-extinguishing element (
For example, GTO, etc.) 81□ to S□4 and freewheel diodes D1□ to Di. The same applies to the ■ phase and the W phase. The second inverter INV-2 is similarly configured. Third inverter (direct inverter) I
NV-3 is connected in a three-phase bridge, of which the U phase is connected to self-extinguishing elements S11. It consists of S work 2, freewheel diode draw work □, and D□2.

A neutral point N of the load is connected to a midpoint Nd of the DC voltage vd. That is, Vd□=Vdz=Vd/2. If the load is a motor load, variable voltage, variable frequency power must be output from the converter.

When the output frequency is zero, no voltage can be obtained from an inverter with an output transformer. Therefore, power is output from the direct-coupled inverter INV-3, and the voltage drop due to the load resistance (U-RU) is supplied.

Output frequency f. There is a certain value fIl+. When it becomes larger, inverter INV-1,,I with output transformer
Voltages Vυ□ and ■υ2 are generated from NV-2. This power VU r VU2 has an output frequency f. The output transformer core is increased or decreased approximately in proportion to the output transformer to prevent saturation of the output transformer iron core.

(Problems to be Solved by the Invention) The conventional power conversion device described above can increase the capacity by increasing the number of inverters with output transformers, and also performs pulse width modulation control of the inverters, and multiplexes the inverters. It has the characteristic that by controlling it, harmonic components of the output voltage can be reduced. On the other hand, the direct-coupled inverter at the lowest stage must be configured with one unit and cannot be multiplexed. Therefore, the harmonic components of the output voltage of the direct-coupled inverter become large, resulting in a drawback that the pulsation of the load voltage υ becomes large. Also, if the voltage drop IU-RU due to load resistance is large, or if the output transformer TR□
, When the minimum frequency fmin at which TR2 can operate is high, the dependence on the direct-coupled inverter INV-3 increases, the capacity of the direct-coupled inverter increases, and as a result, series-parallel connection of elements becomes necessary. When the elements are connected in series and parallel, the snubber circuit becomes complicated and it becomes difficult to regenerate the spanner loss into the power supply. As a result, the switching frequency of the element must be kept low, which has the disadvantage of further increasing the pulsation of the load voltage υ.

The pulsations in the load current generate electromagnetic noise, which causes pulsations in the torque generated by the motor. Furthermore, the peak current value flowing through the elements constituting the inverter increases, and elements with larger capacities are required.

The present invention has been made in view of the above problems, and is a high-performance power source that reduces the pulsation of the output current of a power converter and supplies variable voltage, variable frequency power to a high voltage, large capacity AC load. The purpose is to provide a conversion device.

[Structure of the invention]

(Means for Solving the Problems) In order to achieve the above object, the present invention provides an AC power supply and
A pulse width modulation control converter (PWM converter) whose AC side terminal is connected to the AC power supply, a DC smoothing capacitor connected to the DC side terminal of the converter, and a pulse width modulation control converter (PWM converter) whose DC side terminal is connected to the smoothing capacitor. a modulation control inverter (PWM inverter), an output transformer connected to an AC side terminal of the inverter, and a cycloconverter whose long side terminal is connected to the AC power source;
It consists of an AC load driven by the sum of the output voltage of the cycloconverter and the output voltage of the output transformer.

(Function) The PIIIM converter controls the input current supplied from the alternating current so that the direct current voltage applied to the direct current smoothing capacitor remains approximately constant.

The PldM inverter uses a smoothing capacitor as a DC voltage source and outputs variable voltage and variable frequency power via an output transformer. Although one PWM inverter may be used, in order to increase the capacity of the converter, a plurality of PWM inverters are constructed and multiplexed by an output transformer.

A cycloconverter directly converts constant frequency AC power into variable frequency AC power, and is directly connected to an AC load without any output power. The sum of the output voltage of the cycloconverter and the output voltage of the output transformer of the PIIIM inverter is applied to the AC load.

Output frequency f. When is zero or low frequency, the cycloconverter generates an output voltage, and the output voltage of the PWM inverter is zero. Output frequency f. When the voltage exceeds a certain level, the PIJM inverter also outputs the voltage (2), and the voltage applied to the load is adjusted and controlled in conjunction with the output voltage of the cycloconverter. At this time, in order to prevent saturation of the iron core of the output transformer, the inverter output voltage is
The output frequency f is increased or decreased approximately in proportion to the output frequency f.

Since the cycloconverter operates with natural commutation, it is easy to increase the capacity, and even when the voltage drop due to the load resistance is large or the minimum output frequency at which the output transformer can operate becomes high, it is easy to increase the capacity. The capacity can be increased accordingly. In addition, by increasing the number of control phases of the cycloconverter, it is possible to reduce the pulsation component (harmonic component) of the output voltage, and it is possible to significantly reduce the pulsation of the output current, which has traditionally been a problem. It becomes possible.

Problems with cycloconverters are reactive power on the input side and harmonics of the input current, but the PWM converter is provided with an active filter function to compensate for the reactive power and harmonic current.

As a result, the input current supplied from the AC power supply is controlled to be a sine wave in phase with the power supply voltage, and the converter becomes an ideal converter with an input power factor of 1 and no harmonics.

(Embodiment) FIG. 1 is a configuration diagram showing an embodiment of a power conversion device of the present invention.

In the diagram, BUS is the electric line of the 3-phase AC power supply, TR construction,
2 is a power transformer, Ls is an AC reactor, C0NV is a PIIM converter, Cd is a DC smoothing capacitor, IN
V is the P-th inverter, OTR is the 3-phase output transformer, LO
AD is a three-phase AC load, Lυ+LV+Llil is the inductance of the load, Itu+RVI R11+ is the resistance of the load, VCVI VCVI VCLI+ is the back electromotive force of the load, CAP is a three-phase advance capacitor, and CC is a cycloconverter.

FIG. 2 is a block diagram showing in more detail one phase on the load side of the device shown in FIG.

In the figure, INV-U is the U phase of the PWM inverter INV,
TRυ indicates the U phase of the output transformer OTR, TRccu indicates the U phase of the power transformer TR2, and CC-U indicates the U phase of the cycloconverter. Other symbols follow the symbols in FIG.

Note that the power transformer TR□ is omitted.

The cycloconverter CC-U is a positive group converter SPU
, negative group converter S and DC reactor. □.

It consists of LO2.

The PWM inverter INV-U is composed of self-extinguishing elements (for example, gate turn-off thyristors) S to S4 and freewheel diodes D1 to D.

Also, CTCI CTLI CTcct CTDI C
TxU+cTU, CTPUICTNU stands for current detector, ISO stands for isolation amplifier.

PWM converter C0NV is DC smoothing capacitor c
The input current IC is controlled so that the DC voltage vd applied to d remains approximately constant.

The PWM inverter INV-U is connected to the smoothing capacitor c
d is a DC voltage source, which outputs variable voltage and variable frequency power through pulse width modulation control.

The output transformer TRU insulates the output voltage of the inverter INV-U and applies it to the load, and the sum of the output voltage of the cycloconverter CC-U, which will be described later, is applied to the load.

The cycloconverter CC-U performs direct power conversion from constant frequency alternating current power to variable voltage variable frequency alternating current power, and here a circulating current type cycloconverter is employed. Output voltage V of cycloconverter CC-U
LIz is given by the average value of the output voltage VPU of the positive group converter SPU and the output voltage VNυ of the negative group converter SNU. The output voltage Vυ1 of the output transformer TRU is added to the output voltage VU2 and applied to the load U phase.

Output frequency f. When is zero, the output voltage Vυ1 of the output transformer TRU cannot be expected. Therefore, voltage VU2 is generated from cycloconverter CC-υ, and U
Control the current charge current ■. The minimum frequency fIIl at which the output transformer TRυ can operate. Up to this point, the load current is similarly controlled only by the cycloconverter CC-U.

Output frequency f. becomes high and the frequency at which the output transformer TRU can operate is reached, the PWM inverter INV-
A voltage is generated from U, and a voltage of VU□10Vυ2 is applied to the load. At this time, to prevent saturation of the iron core of the output transformer TRU, the output voltage VU□ is set at the output frequency f. PIIIM inverter INV-U is approximately proportional to
control.

(2) The W-phase load current is similarly controlled.

cycloconverter cc (cc-u, cc-v,
(including CC-W) has the advantage of being able to convert power only by natural commutation operation, but on the other hand, it always lags behind the AC power supply and produces reactive power Q.

It has the disadvantage that it takes 0, and furthermore, this delayed reactive power Qcc has an output frequency f. There is a problem in that the voltage changes every moment in synchronization with the power supply system, causing voltage fluctuations in the power supply system. In addition, harmonic components of the input power supply of the cycloconverter CC also become a problem.
This can cause problems with guidance to communication lines.

Therefore, first, a circulating current type cycloconverter is adopted as the cycloconverter CC, and the circulating current is adjusted so that the delayed reactive current Qcc on the input side of the CC is controlled to be approximately constant.

In addition, a phase advancing capacitor CAP is installed at the power receiving end, and its advancing reactive power Q. ap and the delayed reactive power QCC cancel each other out to keep the input fundamental wave power factor at 1.

Also, regarding the harmonic components of the CC input current, P, 1
By making the 11M converter C0NV also serve as an active filter and controlling the input current ■c of the converter C0NV, harmonic components of the CC input current are compensated for. As a result, the electrician s supplied from the AC power source BUS can be operated without harmonic components and with an input power factor of 1.

FIG. 3 is a configuration diagram showing an embodiment of the control circuit of the U-phase PWM inverter INV-U.

In the figure, CAL is an arithmetic circuit, AD□, AD2 is an adder,
C□ is a comparator, GE(S) is a control compensation circuit, PWM
. is a pulse width modulation control circuit.

VIJl is the voltage V output from the output transformer TRυ
The command value for the U-engineer is the voltage drop Jω due to the motor's back electromotive force vcυ and inductance l, υ at the load.・Lυ・I
Set to output U. Namely.

■. is the load current command value ωo=2πf.

It is given as follows.

Simply put, by using one input υ1 of the adder AD2 as it is as the input signal eU for PWM control, a voltage υ proportional to eU'=Vυ□ is generated from the inverter INV-U. 1 occurs, and if the ratio of the primary and 72nd turns of the output transformer TRυ is 1, then the secondary winding of the transformer Vt1x
occurs. However, in reality, the transformer TRυ is
It is not certain whether the magnet is excited from the primary winding or from the secondary winding. Therefore, the output voltage V
A command value Eυ of the excitation current IEL+ for outputting IJx=VIJ is calculated by the calculator CAL, and the excitation current IEυ is controlled. That is, M determines the excitation current command value IEυ from the excitation inductance of the transformer, goes up to the adder AD□, adds the load current TU (secondary current of the transformer), and obtains the transformer's primary current command value ■□υ It is said that

Next, the comparator C1 compares the above command value with the back type electrician, υ
Compare the deviation εtu=It:rlυ with the control compensation circuit G. (s) and corrects the aforementioned PWM control input signal eU via the adder AD2.

eU = VU, +GE(S) ・E 1u
...++ (3) Note that adder AD1 and comparator C
1 together, the deviation ExU becomes equal to the deviation εEU"IEU-IEU from the relationship: ■Eυ=11U-Iυ.

Therefore, ■. When u > I Eυ, the deviation εEU
becomes a positive value, increasing the output voltage of inverter INV-U and increasing excitation current IEU. On the contrary, ku<
When IEu, the deviation εELI becomes a negative value, and e
υ decreases, lowering the output voltage of inverter INV-U and reducing IEU. Therefore, it is controlled so that ■Eυ=■correction.

This prevents the excitation electrician Eυ of the transformer TR-U from becoming excessive and prevents the iron core from becoming saturated.

FIG. 4 is a block diagram showing an embodiment of the control circuit of the U-phase microconverter CC-U.

In the figure, VR is a reactive power setting device, 02 to c4 are comparison circuits, IQ(S), Gou(s), Gu(s) are control compensation circuits, spc is a motor (load) speed control circuit, and OA is a Inverting proportional amplifier (gain -1), PHP, and PHN indicate phase control circuits, respectively.

First, a load current command value 1 is given from the speed control circuit spc. Also, the load voltage increases to the current detector CTυ. is detected and compared with the command value IU by the comparator C4. The deviation ευ = engineering υ - Iu is the control compensation circuit Gυ (
s), one is given to the phase control circuit circle (P) of the positive group converter SPU through the adder AD3, the other is inverted by the inverting amplifier OA, and the other is amplified by the negative group through the adder AD4. Phase control circuit P of converter SNU
Given to HN. The positive group converter SPU outputs a voltage VPU proportional to the PHP input signal αP. or,
The negative group converter SNU outputs a voltage VNU proportional to the inverted value of the input signal UαN of the PHN, with the direction of the arrow in FIG. 2 being positive.

When ■υ〉■υ, the deviation EU becomes a positive value and increases the output voltages VPU and vNυ of the positive group and negative group converters. Therefore, the output voltage VU2 of the cycloconverter
= (Vpu + VNU)/2 increases, the load current IU increases. Conversely, when ■υ〉■υ, the deviation εU becomes a negative value, the output voltage VU2 decreases, and the load current ■.

decreases, and finally settles down to ■υ=iu.

]5 When the current command value Iυ is changed in a sinusoidal manner, the actual current Iυ follows it and is controlled in a sinusoidal manner.

On the other hand, the reactive power Q on the input side of the cycloconverter CC.

0 is detected, and the command value Q from the reactive power setting device VR is detected by the comparator C2. Compare with C. The deviation εQ=QCCQ
CC is amplified or integrated by a control compensation circuit IIJs) to obtain a circulating current command value ■. shall be.

The comparator C3 connects the U-phase microconverter CC-U.
Detected circulating current value■. U and the above command value ■. Compare the deviation E. = Engineering. −I, υ as the following control compensation circuit G. υ(s)
amplify by. Here, the circulating current of CC-U is ■.

υ is the output current IPU of the positive group converter SPU and the negative group S
Using the output current INυ of UN, it is determined as follows.

Iou= (IPLI+INU IIPU+INUl
)/2 ・= Q) Control compensation circuit G. u(s)
The output signals of each adder AD1. The signal is applied to phase control circuits PNP and PHN via AD. That is, y ap ” Gu (s)・E u+Gou(s)・
[.

... ■ Z' α5=Gu(s)・fu〇. u(s)・t
o... Becomes 0.

Engineering. >Iou, the deviation ε. becomes a positive value,
The output voltage Vpυ of the positive group converter SPU is increased in the direction of the arrow in FIG. 2, and the output voltage v of the negative group converter SNU is increased.
Decrease Nυ. Therefore, DC reactor L. 1jLO2
A voltage of Vpu-VNU is applied to , and a circulating current ■. Increase υ. On the contrary, I. <IoU, the deviation ε.

becomes a negative value, decreasing VPU and increasing VNU, causing the circulating current ■, 1. Decrease l. Finally■. U
Benko. I yell and calm down. At this time, Vpu”FVNυ.

When Qcc>Qcc, the deviation εQ becomes a positive value, and the circulating current of the cycloconverter CC. Un IoV
T-engineering. If V is increased and the delayed reactive power Qcc taken by the CC is increased, and conversely, Qcc>Qcc, the circulating current of the CC is IoUy IoVt. By reducing V, it settles down to Qcc'' = Qcc.

In this way, the delayed reactive power Qcc taken by the cycloconverter CC can be controlled to be constant regardless of the load, which is accompanied by fluctuations in the reactive power. It becomes possible to eliminate fluctuations in power supply voltage.

In addition, a phase advance capacitor CAP is installed at the power receiving end, and the above-mentioned constant lagging reactive power Q is maintained. Leading reactive power Qc that cancels C
By taking ap, the input power factor can always be kept at 1.

FIG. 5 is a configuration diagram showing an embodiment of the control circuit of the PWM converter C0NV.

In the figure, C5 and C are comparators, AD5 to AD7 are adders/subtractors, GO(S) and GR(S) are control compensation circuits, 衚 is a multiplier, PldMR is a pulse width modulation control circuit of an R-phase PIdM converter, FF is a feedforward control circuit.

First, the insulation amplifier ISO detects the DC voltage vd applied to the smoothing capacitor cd, and inputs it to the comparator C5. The comparator C5 compares the DC voltage command value ■d and the voltage detection value vd, and the deviation εd=Vd−Vd is amplified or integrated by the control compensation circuit GD(S).

In addition, PWM inverter INV (INV-U, IN
(including V-V, INV-W) taken by Pd=v
d·Id is detected and input to the feedforward control circuit FF, and the following calculation is performed. However, VSm is the peak value of the power supply voltage.

■Confucian = 2Pd 3vs, n... ■The output signal lSm0 of the relevant FF and the output signal ΔISm of the previous 1EGD(s) are added by the adder AD5, and the peak value command of the power supply current ■s□I"l5IIIO+Δ■2□ This peak value command Δ■511+ is multiplied by a unit sine wave sjnω2t synchronized with the power supply voltage by a multiplier ML to obtain a power supply current command value ISR+ SS+ isT.

Only the R phase will be explained below. The S phase and T phase are similarly controlled.

The input current 'L' (ILRI
ILSy ILT) is detected, and the subtractor AD calculates the power supply type flow command value Is (Is+R+Iss+l5
T). AD, output ICR”ILRISR
is the R-phase input current command value of the PWM converter C0NV. The comparator C6 compares the R-phase input current detection value ICR of the converter and the above command value NCR, and calculates the deviation εO.
R"ICRICR is input to the control compensation circuit GR(S), where it is inverted and amplified, and passed through the adder AD7 to become the input signal eR of the pulse width modulation control circuit PυM. The other input signal ESR of the adder AD is This is to compensate the power supply voltage (R phase), and from the PIi1M converter C0NV in advance.
It generates a counter voltage corresponding to the power supply voltage VSR (R phase) and serves to improve the control response of the input current ICR.

The PWM converter C0NV generates a voltage VCR proportional to the input signal eR at the AC side terminal.

When ICR>ICR, the deviation εCR becomes a positive value and is inverted and amplified via GR(S), so that the input signal eB of the PWM control circuit decreases. Therefore, the AC side generated voltage VcR of the PWM converter C0NV decreases, and the voltage VSR applied to the AC reactor LSR (R phase component of LS) decreases.
-=20 VCR increases. As a result, the input terminal ■. R increases and is controlled so as to approach the command value ICH. On the contrary, I.C.
If R>Icrt, Plt1M converter C0
The generated voltage VCR of NV increases, the voltage applied to the LSR decreases, and the input current ICR decreases. As expected, ICR”
FICR is controlled.

FIG. 6 shows a voltage-current vector diagram of the R phase of the AC power supply.
It is assumed that the input current ILR is flowing through C and the phase advancing capacitor CAP with a lagging phase angle of 0.

Since the power supply command value ISR is given as a sine wave in phase with the power supply voltage VSR, the input current command value I of the PWM converter C0NV. R''ISRILR is given by the vector shown in the figure.As described above, the input current ICR of the PWM converter C0NV is controlled to match the command value NCR, so it becomes ICR''ICR. Therefore, the actual power supply current IS
Since R is the sum of ICIt and ILR, ISR”IC
R+ILR ICR+ILR=■SR-■LR+■LR=ISR...0. In other words, the power supply current ISR is equal to the power supply voltage VSR
It is controlled to a sine wave (with small harmonics) that is in phase with (input power factor = 1).

Next, the operation of compensating for harmonic components included in the ILR will be explained.

Now, suppose the output frequency is f. The operation will be described assuming that cycloconverter CC is low and power is supplied to load LOAD only from cycloconverter CC.

FIG. 7 shows waveforms of input currents at various parts. The input current ILR of the cycloconverter CC, including the input current of the phase advance capacitor CAP, has a power factor of 1 but includes a third high frequency component.

The power supply current command value ISR is given as a sine wave current synchronized with the power supply voltage VSR as shown by the broken line. Therefore, the input current command value of PldM converter C0NV ICR” ISR+
As shown, the ILR is applied so as to cancel the harmonic components contained in the ILR, and as a result, the power supply current ISR always matches the command value ISR and becomes a sine wave current containing no harmonics.

Now, returning to FIG. 5, the control operation of the DC voltage vd will be explained.

When Vd>Vd, the deviation εd becomes a positive value, and the peak value command IS+ of the power supply current. increase. the result,
PWM+inverter C0NV input current command ICR+I
C8+ICT increases and inrush current ■. R+I. Sr.
ICR also increases. As a result, the effective power PB = (3/2) ・Vsm'Ism supplied from the AC power supply increases, and its energy Ps・t increases as the DC smoothing capacitor Cd
It is accumulated as d(1/2)Cd-Vd2. Therefore,
DC voltage vd increases and approaches command value vd. On the contrary, vd
<vd, the deviation εd becomes a negative value, and the active power Ps supplied from the AC power supply decreases. Therefore, the energy stored in the DC smoothing capacitor Cd decreases, causing the DC voltage Vd to decrease, and is also controlled so that Vd=Vd.

Here, consider a case where the energy storage capacity of the DC smoothing capacitor cd is not very large.

When the load on the PIIIM inverter INV is suddenly increased, the feedback control of the DC voltage alone cannot respond in time, and the DC voltage vd decreases, causing the PWM converter C0NV or PWM inverter JNv to become uncontrollable due to the lack of DC voltage. There is a danger. or.

On the other hand, if the load is suddenly reduced, the DC voltage Vd will rise, which may lead to element broken lines due to overvoltage.

Therefore, the feedforward control circuit FF is useful.

When the load on the PWM inverter INV increases rapidly, the DC power (active power) Pa of the inverter also increases immediately,
Through the FF circuit, the power supply current peak value command Is expressed by the formula
Increase mo.

Assuming that □ is not very large in the output signal Δ of the DC type suppression # circuit Go (s), it becomes 4 IS - SmO, and the power supply current ISR + ISS + IST also increases on the far side, and the effective power Ps'' (3/ 2) ・Vsm' Is
Supply m=Pd. Therefore, it flows in and out of the DC smoothing capacitor cd. There is no change in active power, and DC voltage vd
will no longer decrease.

On the other hand, when the load suddenly decreases, the feedforward control circuit FF immediately decreases the active power Ps supplied from the AC power source, thereby preventing the DC voltage Vd from increasing.

In the embodiment shown in FIG. 1, one inverter is used as the PWM inverter INV, but two or more inverters are prepared, and each output terminal is isolated by an output transformer. Multiplexing control may be performed using pulse width modulation control.

It goes without saying that either a circulating current type or a non-circulating current type cycloconverter may be used.

〔Effect of the invention〕

As described above, according to the power conversion device of the present invention, by increasing the number of control phases of the cycloconverter directly connected to the load, the pulsation component (harmonic component) of the output voltage can be reduced. It becomes possible to significantly reduce the pulsation of the output current, which has been a problem in the past. In addition, since the cycloconverter operates with natural commutation, it is easy to increase the capacity, and even when the voltage drop due to load resistance is large, or the inverter's output transformer can operate or the minimum output frequency is high. The capacity can also be increased accordingly. Furthermore, the power supply current can be controlled to a sine wave in phase with the power supply electricity, and an ideal power conversion device with an input power factor of 1 and few harmonics can be provided.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing an embodiment of the power conversion device of the present invention, FIG. 2 is a block diagram of one phase on the output side to explain the operation of the device in FIG. 1, and FIGS. 5 and 5 are block diagrams showing the control circuit of the device shown in FIG. 1, FIGS. 6 and 7 are vector diagrams and current waveform diagrams for explaining the operation of the device shown in FIG. 1, and FIG. 1 is a configuration diagram of a conventional power conversion device. +3US...Electric line of 3-phase AC power supply TR□, TR2...Power transformer L8...AC reactor C0NV...Pulse width modulation control converter cd...DC smoothing capacitor INV...Pulse width modulation control inverter OTR...
Output transformer CAP... Phase advance capacitor CC... Cyclo comparator

Claims (2)

[Claims] (1) An AC power supply, a pulse width modulation control converter whose AC side terminal is connected to the AC power supply, a DC smoothing capacitor connected to the DC side terminal of the converter, and a DC side terminal connected to the smoothing capacitor. a pulse width modulation controlled inverter, an output transformer connected to an AC side terminal of the inverter, a cycloconverter whose input side terminal is connected to the AC power supply, an output voltage of the cycloconverter, and an output voltage of the output transformer. What is claimed is: 1. A power conversion device comprising: an AC load driven by the sum of (2) The pulse width modulation control converter is characterized by comprising means for controlling the input current of the converter so as to cancel part or all of the reactive power or harmonic current on the input side generated by the cycloconverter. The power conversion device according to claim 1.