JPH0157866B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a synchronous receiver using a synchronous carrier regeneration method, and an object of the present invention is to obtain two synchronous carrier waves having a phase difference of 90 degrees from each other over a wide frequency range and at high speed.

FIG. 1 shows a conventional example. An example of this is a synchronous carrier regeneration type synchronous receiver known as a Costas loop. This receiver has a first synchronous detector 1 that synchronously detects the in-phase component of a modulated carrier input, a second synchronous detector 2 that synchronously detects the orthogonal component, and the outputs of each of these two synchronous detectors 1 and 2. Low-pass filters 3 and 4 perform low-pass filtering, a phase comparator 5 that multiplies the outputs of these two low-pass filters 3 and 4 by voltage, and a low-pass filter 6 that performs low-pass filtering on the output of this phase comparator 5. , a voltage-controlled oscillator 7 controlled by the output voltage of this low-pass filter 6, and a 90° phase shifter 8 that shifts the phase of the output of this voltage-controlled oscillator 7 by 90°. From the in-phase and quadrature component signals obtained by synchronous detection by the synchronous detector 2 of 2, the modulated carrier wave input and the voltage controlled oscillator 7
It detects the phase error with the output of the carrier, that is, the reproduced carrier wave, and operates to minimize this error.

In order to apply the synchronous receiver shown in Figure 1 to a television receiver or a radio receiver, a broadband
Requires 90° phase shifter. The present invention attempts to use a phase lock loop (PLL) that synchronizes when the phase difference between two input signals is 90 degrees, instead of this 90 degree phase shifter.

FIG. 2 shows a block diagram of an embodiment of the present invention. Blocks 1 to 7 in FIG. 2 each show the same elements as 1 to 7 in FIG. 1, and their operations are also the same. Phase comparator 9, low pass filter 1
0, the loop consisting of voltage controlled oscillator 11 is PLL
This corresponds to the 90° phase shifter 8 in FIG. The operation of this loop is as follows.

The output υ ₀ (t) of the first voltage controlled oscillator 7 is υ ₀ (t) = A ₀ cos [ω ₀ t+ ₀ (t)] ......(1) The output υ _Q of the second voltage controlled oscillator 11 (t) as υ _Q (t)=A _Q cos [ω _Q t+ _Q (t)] ......(2)
Add to. However, ω ₀ and ₀ (t) indicate the frequency and phase of the oscillation output of the first voltage controlled oscillator 7, and _Q and _Q (t) indicate the frequency and phase of the oscillation output of the second voltage controlled oscillator 1.
1 shows the frequency and phase of the oscillation output.

Since the phase comparator 9 is a voltage multiplier, its output υ _d (t) is υ _d (t)=υ ₀ (t)・υ _Q (t) = A ₀ A _Q cos [ω ₀ t+ ₀ (t )〕cos〔ω _Q t+ _Q (t
)] =A ₀ A _Q /2 {cos[(ω ₀ +ω _Q )t+ ₀ (t)+ _Q (t)] +cos[(ω ₀ −ω _Q )t+ ₀ (t)− _Q (t)]}
………(3) Now let ω ₀ = ω _Q , then υ _d (t)=A ₀ A _Q /2{cos[2ω ₀ t+ ₀ (t)+ _Q (
t)] + cos [ ₀ (t) − _Q (t)]} ………(4) When υ _d (t) is filtered by the low-pass filter 10 to remove the frequency component of 2ω ₀ , the output υ _C1 (t) is υ _C1 (t) = A ₀ A _Q /2cos [ ₀ (t) - _Q (t)]...
...(5) becomes. Therefore, the difference _e between ₀ (t) and _Q (t)
When (t) is _e (t) = ₀ (t) - _Q (t) = ±90° (6), υ _C1 (t) = 0. With this state as a reference, if _Q (t) leads or lags in phase with respect to ₀ (t), control is performed so that the difference between _Q (t) and ₀ (t) becomes 90°. That is, the outputs of the first voltage controlled oscillator 7 and the second voltage controlled oscillator 11 have a phase difference of 90 degrees.

The above explanation has been made assuming ω ₀ =ω _Q , but next we will explain the means for obtaining this state. A constant current Is is supplied from the current switch 12 to the connection point between the resistor R ₂ and the capacitor C of the low-pass filter 10 . Since the integrated voltage is obtained from the low-pass filter 10 by using the product of R ₂ and C as an integral constant, the second voltage controlled oscillator 11
The output frequency ω _Q (t) of is in a sweep state. Therefore, the difference ω _Q (t)−ω ₀ between this frequency and the output frequency ω ₀ of the first voltage controlled oscillator 7 is also in the sweep state.
If the speed of the sweep is Δω〓rad/sec ² , then the input phase of the PLL consisting of the phase comparator 9, low-pass filter 10, and voltage-controlled oscillator 11 becomes −1/2Δω〓t ² .

The aforementioned υ ₀ (t), ₀ (t), υ _Q (t), _Q (t)
, υ _d
(t), υ _C1 (t) and _e (t)−π/2 are expressed as V ₀ (s), Φ ₀ (s), V _Q (s), Φ _Q, respectively.
(s), V _d (s), V _C1 (s) and Φ _e (s), the sensitivity of the phase comparator 9 is K _d , the control sensitivity of the voltage controlled oscillator 11 is K _Q , and the sensitivity of the low-pass filter 10 is The transmission characteristics are F
(s), then V _d (s) = K _d [Φ ₀ (s) - Φ _Q (s)] ...... (7) V _C1 (s) = F (s) V _d (s) ... ...(8) Φ _Q (s) = K _Q V _C1 (s)/S ......(9) From equations (7) to (9), the loop transfer function H(s) is H(s) = Φ _Q (s) / Φ ₀ (s) = K _Q K _d F (s) /
s+K _Q K _d F (s) ......(10) And Φ _e (s) / Φ ₀ (s) = Φ ₀ (s) - Φ ₀ (s) - Φ _Q (
s)/Φ ₀ (s) = 1-H (s) = s/s + K _Q K _d F (s) ......(11) Therefore, the phase difference Φ _e (s) is Φ _e (s) = sΦ ₀ (s)/s+K _Q K _d F(s)
......(12) Φ ₀ (s) in equation (12) is the phase comparator 9 and the low-pass filter 1.
0, this is the input phase of the PLL consisting of the voltage controlled oscillator 11, so using -1/2∆ω〓t ² obtained earlier, Φ ₀ (s) = -∆ω〓/s ³ ...... (13) Also, If the configuration of the low-pass filter 10 is an active filter as shown in FIG. 2, F(s)=sτ ₂ +1/sτ ₁ ………(14) However, τ ₁ = R ₁ C, τ ₂ = _R2C .

Substituting equations (13) and (14) into equation (12), Φ _e (s)=−Δω〓/s/s ² +2ζω _o +ω _o ² …
...(15) However, Therefore, _e (t)-π/2 in steady state is lim t→∞ [ _e (t)-π/2] = lim s→ ₀ sΦ _e (s) = lim s→ ₀ −Δω〓/s ² +2ζω _o +ω _o ² = −Δω/ω〓 _o ² (rad) ......(16) It becomes a constant value. That is, it shows that ω _Q (t)=ω ₀ .

However, since the phase difference shown in equation (16) remains, in order to reduce this to 0, the current switch 12
After sufficient time has elapsed for the constant current Is to reach a steady state, the constant current Is is cut off. . At this time, equation (16) is applied to Φ ₀ (s) in equation (12).
If _we enter the _{value of} Output of first voltage controlled oscillator 7 and second voltage controlled oscillator 11
The phase difference of the output is 90°.

After all, the phase comparator 9 and the low-pass filter 1 in FIG.
0.0, a PLL consisting of a voltage controlled oscillator 11 could replace the 90° phase shifter in FIG. Moreover,
This is a 90° phase shifter over a wide frequency range.

Note that when the PLL mentioned above loses lock due to external noise, etc., the voltage controlled oscillator 11
In order to start sweeping, a voltage comparator 13 is provided to control the direction of the output current of the current switch 12 when the output voltage of the low-pass filter 10 reaches a certain value.

FIG. 3 shows a configuration for facilitating the retracting operation of the Costas loop shown in FIG. Blocks 1 to 8 in FIG. 3 each represent the same elements as 1 to 8 in FIG. 1, and their operations are also the same. Bandpass filter 14, differentiator 15, voltage multiplier 1
6. The low-pass filter 17 and the voltage adder 18 are PLL pull-in circuits.

If the input to the synchronous detector 1 is A _i cosω _i t, the output of the voltage controlled oscillator 7 is A ₀ cosω ₀ t, and the output of the 90° phase shifter 8 is A ₀ sinω ₀ t, then the output of the synchronous detector 1 is The output is υ _pc (t) = A ₁ A ₀ cosω _i tcosω ₀ t = A _i A ₀ /2{cos(ω _i +ω ₀ )t+cos(ω _i −ω ₀ )t
} ………(18) This signal is filtered by the low-pass filter 3 and the bandpass filter 14, and differentiated by the differentiator 15, so that υ _pc ′(t)=−A _i A ₀ /2(ω _i − ω ₀ ) sin(ω _i −ω ₀
) t ......(19) Also, the output of the synchronous detector 2 is υ _ps (t) = A _i A ₀ cosω _i tsinω ₀ t = A _i A ₀ /2 {sin (ω _i + ω ₀ ) t-sin (ω _i −ω ₀ )t
} ………(20) When this signal is low-pass filtered by low-pass filter 4, υ _ps ′(t)=−A _i A ₀ /2sin(ω _i −ω ₀ )t ………(21) These υ _pc ′(t) and υ _ps ′(t) by voltage multiplier 16
If you multiply by, υ _pcs (t)=1/2(A _i A ₀ /2) ² {ω _i
−ω ₀ ){1−cos2(ω _i −ω ₀ )t} (22) is obtained. This equation shows that the DC component of υ _pcs (t) is proportional to the difference between the input frequency to the synchronous detector 1 and the output frequency of the voltage controlled oscillator 7, and is a signal whose polarity is the sign of the difference. In other words, it shows a frequency discrimination function. The signal of equation (22) is passed through the low-pass filter 17
The output signal is low-pass filtered by the voltage adder 18, and the voltage is added to the output of the low-pass filter 6 of the Costas loop.

Incidentally, the filtering characteristics of the bandpass filter 14 are shown in FIG. The lower cutoff frequency f _L is determined from the noise bandwidth of the Costas loop, and f _U is determined from the pull-in range of this loop.

After all, in the configuration shown in FIG. 3, by frequency-discriminating the difference between the input frequency to the synchronous detector 1 and the output frequency of the voltage-controlled oscillator 7, it is possible to widen the range of the Costas loop.

Next, a specific example in which the present invention is applied to a television channel selection device will be described. In FIG. 4, a plurality of television broadcast waves are input to the high frequency input section 19. Let the video carrier wave of a certain channel among them be υ _V (t), and the audio carrier wave be υ _s (t). υ _v (t
)
is modulated by the residual sideband, so υ _v (t)=R _e {[I(t)+jQ(t)]expj[ω _v t
+ _v ]} =I(t)cos[ω _v t+ _v ]−Q(t)sin[ω _v
t+ _v ]......(23) Here, R _e is the real part of the expression in { },
I(t) includes a video signal with an amplitude of an in-phase component with respect to a carrier wave. Q(t) is the amplitude of a component orthogonal to the carrier wave, ω _v is the angular frequency of the video carrier wave, and _v is the phase of the video carrier wave.

Let the output of the first voltage controlled oscillator 20 be υ ₀ (t) = A ₀ cos (ω ₀ t+ ₀ ) ......(1)', and use this as the voltage along with the image carrier wave υ _v (t) in equation (23). When applied to the first synchronous detector 21 consisting of a multiplier, its output υ _pv (t) is υ _pv (t) = {I(t) cos [ω _v t+ _v ] −Q(t) sin [ω _v t+ _v ]}A ₀ cos(ω ₀ t+ ₀ ) =A ₀ I(t)cos[ω _v t+ _v ]cos(ω ₀ t+ ₀ ) −A ₀ Q(t) sin[ω _v t+ _v cos(ω ₀ t+ ₀ ) =A ₀ I(t)/2 {cos[ω _v +ω ₀ )t+ _v + ₀ ] +cos[(ω _v −ω ₀ )t+ _v − ₀ ]} −A ₀ Q(t)/2 {sin [(ω _v + ω ₀ ) t+ _v + ₀ ] + sin [(ω _v − _{ω 0} ) t+ _v − ₀ ]} ………(24) Now, when the voltage controlled oscillator output is synchronized with the video carrier wave, ω Since ₀ = ω _v , υ _pv (t) = A ₀ I(t)/2 {cos [2ω _v t+ _v + ₀ ]
+cos〔 _v − ₀ 〕} −A ₀ Q(t)／2{sin〔2ω _v t＋ _v ＋ ₀ 〕＋sin〔
_v − ₀ ]}……(25) When 2ω _v is removed by the low-pass filter 22, υ _pv (t)=A ₀ I(t)/2cos−A ₀ Q(t)/2sin……( 26) Here, is _v − ₀ , which is the phase difference between the video carrier and the voltage-controlled oscillation output. If = 0, υ _pv (t) = A ₀ I (t)/2 (27). That is, the amplitude of the synchronous component with respect to the video carrier wave is obtained as a detection output. However, the amplitude of the orthogonal component is not detected. The detection output A ₀ I(t)/2 is output as a video detection output through a low-pass filter 22, amplified by a signal amplifier 23, and outputted to a video output device 25 through a video signal amplifier 24. The filtering characteristics of the low-pass filter 22 are shown in FIG. In FIG. 6, 49 indicates a video signal baseband, and 50 indicates an audio carrier wave. The video signal is filtered at baseband as shown in this figure.

Since the audio carrier wave υ _s (t) of television broadcasting is frequency modulated, υ _s (t) = A _s cos [{ω _s + S (t)} t + _s ]
It can be expressed as (28). where A _s is the amplitude of the audio carrier, ω _s
is the angular frequency of the audio carrier wave, S(t) is the audio signal,
_s is the phase of the audio carrier.

This υ _s (t) and υ ₀ (t) of equation (1)′ are calculated by the synchronous detector 2.
1, its output is υ _ps = A _s cos [{ω _s + S(t)} t+ _s ] A ₀ cos (ω ₀
+ ₀ ) =A _s A ₀ /2cos [(ω _s + ω ₀ )t+S(t)t+ _s +
₀ ] +A _s A ₀ /2cos [(ω _s −ω ₀ )t+S(t)t+ _s −
₀ ]……(29) When the frequency component of ω _s + ω ₀ is removed by the low-pass filter 22, υ _ps (t)=A _s A ₀ /2cos [(ω _s −ω ₀ )t+S(t)
t+ _s − ₀ ] = A _s A ₀ /2cos [{ω _IF + S(t)} t+ _IF ]...
…(30) Here, ω _IF is the audio intermediate angular frequency at ω _s − ω ₀ , and _IF is
_s − ₀ is the phase of the audio intermediate carrier. Formula (30)
is nothing but the high frequency audio carrier wave shown in equation (28) converted into an intermediate frequency audio carrier wave.

The filtering characteristics of the low-pass filter 22 are designed to cover the intermediate frequency audio carrier frequency ω _IF as shown in FIG. Although 4.5MHz is used as the ω _IF , this figure is an example of the NTSC system, and the frequency is different for other systems. The intermediate frequency audio carrier wave passes through this low pass filter 22 and is amplified by a signal amplifier 23 and an audio intermediate frequency amplifier 26. The output is detected by a frequency discriminator 27 to obtain an audio signal S(t). S(t) is supplied to the audio output device 28.

In the above explanation, it is assumed that there is no difference between the phase of the video carrier wave υ _v (t) and the phase of the output υ ₀ (t) of the voltage controlled oscillator 20, that is, =0.
This state can be obtained as follows.

A PLL (9 in FIG. 2,
10 and 11), the second
Since the output υ _Q (t) of the voltage controlled oscillator 31 has a phase difference of 90° from the output of the first voltage controlled oscillator 20, υ _Q (t)=A ₀ sin(ω ₀ t+ ₀ ) ......( 31) When this is added to the second synchronous detector 32 consisting of a voltage multiplier along with the video carrier wave υ _v (t) in equation (23), its output υ _PQ (t) is υ _PQ (t)={I( t)cos[ω _v t+ _v ] −Q(t) sin[ω _v t+ _v ]}A ₀ sin(ω ₀ t+ ₀ ) =A ₀ I(t) cos[ω _v t+ _v ] sin(ω ₀ t+ ₀ ) −A ₀ Q(t) sin [ω _v t+ _v ] sin (ω ₀ t+ ₀ ) = A ₀ I(t)/2 {sin [(ω _v + ω ₀ ) t+ _v + ₀ ] −sin [( ω _v −ω ₀ )t+ _v − ₀ ]} −A ₀ Q(t)/2{−cos[(ω _v +ω ₀ )t+ _v + ₀
] +cos [(ω _v −ω ₀ )t+ _v − ₀ ]} ………(32) Since ω ₀ =ω _v , υ _PQ (t)=A ₀ I(t)/2{sin[2ω _v t+ _v + ₀ ]
−sin [ _v − ₀ ]} −A ₀ Q (t) / 2 {−cos [2ω _v t + _v + ₀ ] + cos
[ _v − ₀ ]}……(33) When the 2ω _v signal is removed by the low-pass filter 33, υ _PQ (t)=−A ₀ I(t)/2sin−A ₀ Q(t)/2cos
......(34) is obtained. This υ _PQ (t) is amplified by a signal amplifier 34 and applied to a phase comparator 35.

The control voltage υ _C2 (t) is determined by the phase comparator 35, and υ _PV (t) and υ _PQ obtained from equations (26) and (34)
(t) multiplied by voltage.

υ _C2 (t)=υ _PV (t)・υ _PQ (t) = {A ₀ I(t)/2cos−A ₀ Q(t)/2sin} {−A ₀ I(t)/2sin−A ₀ Q(t)/2cos} =-A ₀ ² /8 {I(t) ² -Q(t) ² }Θ −A ₀ ² /4{I(t)Q(t)} ......(35) Here Θ=2.

Since the video carrier wave υ _V (t) has vestigial sideband characteristics, the in-phase component I(t) is always larger than the quadrature component Q(t). Therefore, −A ₀ /8{I(t) ² −Q(t) ² }≠0. At this time, if the loop noise bandwidth is narrow enough to eliminate -A ₀ ² /4 {I(t)Q(t)}, the first voltage controlled oscillator 20
It is controlled so that That is, 20, 21,
22, 23, 29, 31, 32, 33, 34, 3
The PLL consisting of 5, 36, 37, and 38 is controlled so that =0. When = 0, formula (27)
As already shown in , the amplitude of the in-phase component of the video carrier wave is obtained as the detection output. This PLL
constitutes the Costas Loop. Note that the low-pass filter 36 is a low-pass filter that low-pass filters the signal expressed by equation 35, and the voltage adder 37 is the low-pass filter 36.
The voltage adder 38 is a voltage adder that adds the output of the Costas loop, that is, the control voltage of the Costas loop, and the output voltage of the frequency discrimination circuit for pulling in the Costas loop, which will be described next. This is a voltage adder that adds the output of the voltage selection circuit.

Bandpass filter 39, differentiator 40, voltage multiplier 4
1, the low-pass filter 42 and the voltage adder 37 are respectively 14, 15, 16, 17 and 1 in FIG.
This is a frequency discrimination circuit for drawing in a Costas loop corresponding to No. 8. The voltage selector 43 is the control input device 4
4, the circuit selects a channel selection voltage from the voltage storage device 45 in accordance with the channel desired to be selected. Voltage storage device 45 may be a variable resistor, digitized voltage storage device, or other storage device. Current switcher 46, voltage comparator 47
correspond to 12 and 13 in FIG. 2, respectively, and the voltage switching controller 48 and the current switching controller 46 correspond to 29, 30,
This is a controller for controlling cutting off of the constant current _Is after a sufficient time has elapsed for the PLL consisting of 31 to reach a steady state.

By adopting the above configuration, a tuning voltage is selected corresponding to the desired station inputted from the control input device 44, and this voltage is applied to the first voltage controlled oscillator 20 to first generate the required oscillation. A signal with a frequency close to the wave number is oscillated, and a PLL consisting of 29, 30, and 31 is used to oscillate the signal in the second voltage controlled oscillator 31 with a phase difference of 90°, and these two oscillators are The high frequency input section 1 is made up of a Costas loop as a voltage controlled oscillator and a wave number discriminator circuit for pulling in consisting of 39, 40, 41, 42 and 37.
It is possible to synchronously receive the desired station input from 9.

Note that the high frequency input section 19, the first voltage controlled oscillator 20, the first synchronous detector 21, and the phase comparator 2
9. High mobility devices, especially GaAs devices, are effective in operating the second voltage controlled oscillator 31 and the second synchronous detector 31 at a high frequency near 1 GHz.

As described above, in the present invention, in a synchronous receiver using a synchronous carrier regeneration method, the output of the first voltage controlled oscillator is input instead of the 90° phase shifter,
A PLL in which the output of the second voltage controlled oscillator is in phase with this input by 90° is used, and a constant current is supplied to the connection point of the integrating circuit element of the low-pass filter of this PLL to perform the second voltage control. Since it has a configuration that sweeps the oscillation frequency of the oscillator, a circuit that can replace the 90° phase shifter, which is easy to configure at a single frequency but difficult to configure over a wide range of frequencies, can be obtained. Moreover, it is configured to operate quickly.

For example, if the present invention is applied to a television receiver, the phase of the oscillation output of the first voltage controlled oscillator 20 is synchronized with the phase of the video carrier wave of the desired station to obtain a video signal and an audio intermediate frequency signal. Therefore, compared to the tuning method using an inductor and a variable capacitance diode as in the conventional television tuner, it is easier to integrate the circuit, and it is possible to eliminate adjustment in the tuner manufacturing process.

In addition, the video signal demodulated to baseband,
The audio intermediate frequency signal converted to an intermediate frequency is
If the configuration is such that the low-pass filter 22 with filtering characteristics as shown in the figure filters the signal, and the signal amplifier 23 amplifies both, the video signal and the audio intermediate frequency signal are amplified by one amplifier. Nevertheless, the video signal does not interfere with the audio signal as in the conventional intercarrier audio reception system. This is particularly effective as a countermeasure against buzz in stereo music broadcasts with overlapping audio.

[Brief explanation of drawings]

FIG. 1 is a block diagram of the main part of the conventional example, FIG. 2 is a block diagram of the main part of an embodiment of the present invention, FIG. 3 is a block diagram of the main part of the frequency discriminator circuit for Costas loop, and FIG. FIG. 5 is a block diagram of a main part of another embodiment of the invention, FIG. 5 is a diagram showing filtering characteristics of a bandpass filter of a frequency discrimination circuit for Costas loop, and FIG. 6 is a diagram showing an example of filtering characteristics of a signal amplifier. DESCRIPTION OF SYMBOLS 1...First synchronous detector, 2...Second synchronous detector, 3, 4...Low pass filter, 5...Second phase comparator, 6...Low pass filter, 7... ...First voltage controlled oscillator, 9... First phase comparator, 10... Low pass filter, 11... Second voltage controlled oscillator, 12
... Current switcher, 20 ... First voltage controlled oscillator, 29 ... First phase comparator, 30 ... Low pass filter, 31 ... Second voltage controlled oscillator.

Claims (1)

[Claims] 1. First and second voltage controlled oscillators, sweeping means for sweeping the output frequency of the second voltage controlled oscillator, and phases of the outputs of the first and second voltage controlled oscillators. a first phase comparator that compares the
The outputs of the first and second voltage controlled oscillators are respectively synchronous carrier waves, and the first and second voltage controlled oscillators detect in-phase and quadrature components of the modulated carrier wave input from the receiver input section using these two synchronous carrier waves. a synchronous detector, and a second phase comparator that detects a phase difference between the modulated carrier input and the first voltage controlled oscillator output from the outputs of the first and second synchronous detectors, a first low-pass filter for low-pass filtering the output of the first phase comparator;
The output of the first voltage controlled oscillator is input to a first phase lock loop including a voltage controlled oscillator, and the sweeping means is configured to input the constant current output from the current switch to the first low pass filter. The integrated voltage is applied to the second voltage controlled oscillator and the voltage comparator, and the output of the voltage comparator is used to control the current switch and the first phase lock loop. After a sufficient time to reach a steady state, the constant current is cut off, and the output of the second phase comparator passes through a second low-pass filter to the first voltage controlled oscillator. 1. A receiver comprising a second phase lock loop for controlling. 2. The receiver according to claim 1, wherein the modulated carrier wave is a television signal carrier wave.