JP7094685B2 - Switching power supply - Google Patents

Description

The present invention relates to a switching power supply device (for example, a resonant converter such as an LLC resonant converter).

FIG. 2 is a schematic configuration diagram of a conventional LLC resonance converter described in Patent Documents 1 to 3 and the like.

This LLC resonance converter has a pair of positive and negative input terminals 1a and 1b for inputting a DC voltage, and an input capacitor 2 and a switching circuit 3 are connected in parallel between the input terminals 1a and 1b. The switching circuit 3 is composed of a series circuit of two switching elements 3a and 3b that operate alternately on / off, and a resonance circuit 4 is connected to the output side thereof. The resonance circuit 4 is composed of a series circuit of a resonance capacitor 4a, a resonance inductor 4b, and an exciting inductor 4c, and a primary winding 5a of a transformer (hereinafter referred to as “transformer”) 5 is connected in parallel to the exciting inductor 4c. Has been done. A rectifier circuit 6 is connected to the secondary winding 5b of the transformer 5. The rectifier circuit 6 is configured by bridging four switching elements 6a, 6b, 6c, and 6d, and an output capacitor 7 is connected in parallel to the output side. A pair of positive and negative output terminals 8a and 8b are connected between both electrodes of the output capacitor 7.

In the LLC resonance converter having such a configuration, the DC voltage input to the input terminals 1a and 1b is converted into an AC voltage by the switching circuit 3, and the resonance circuit 4 resonates with the AC voltage. The resonated AC voltage is voltage-converted by the transformer 5, rectified by the rectifier circuit 6, smoothed by the output capacitor 7, and then output from the output terminals 8a and 8b.

In this type of LLC resonance converter, it is generally known that the excitation inductor 4c in the resonance circuit 4 uses the excitation inductance of the transformer 5 and is externally attached.
Further, when the output power of the LLC resonance converter becomes large, the circuits may be parallelized to disperse the current and heat.

FIG. 3 is a schematic configuration diagram of another conventional LLC resonance converter in the case where the primary side of the transformer is parallelized in two.

In the LLC resonance converter of FIG. 3, two switching circuits 3-1 and 3-2 are connected in parallel to the input terminals 1a and 1b. One switching circuit 3-1 is composed of a series circuit of two switching elements 3a1 and 3b1 that operate alternately on / off, and the other switching circuit 3-2 also operates two switching elements that operate alternately on / off. It is composed of a series circuit of 3a2 and 3b2. A resonance circuit 4-1 is connected in parallel to one switching element 3b1, and a resonance circuit 4-2 is also connected in parallel to the other switching element 3b2. One resonance circuit 4-1 is composed of a series circuit of a resonance capacitor 4a1, a resonance inductor 4b1 and an excitation inductor 4c1 (number of turns N1, impedance Lm1), and the other resonance circuit 4-2 also has a resonance capacitor 4a2 and a resonance inductor 4b2. It is composed of a series circuit of an exciting inductor 4c2 (number of turns N1, impedance Lm1).

The primary winding 5a1 (turn number n1) of the transformer 5A is connected in parallel to one excitation inductor 4c1, and the primary winding 5a2 (turn number n1) of the transformer 5A is also connected in parallel to the other excitation inductor 4c2. Has been done. Similar to FIG. 2, a rectifier circuit 6 is connected to the secondary winding 5b (number of turns n2) of the transformer 5A, and an output capacitor 7 and output terminals 8a and 8b are connected to the output side thereof.

In the LLC resonant converter of FIG. 3, the switching circuit 3-1 and the resonant circuit 4-1 and the switching circuit 3-2 and the resonant circuit 4-2 operate in parallel, so that when the output power becomes large. In addition, the current and heat can be distributed in half.

In Patent Documents 1 to 3, as shown in FIG. 2, a circuit configuration in which the exciting inductor 4c is separated from the transformer 5 is disclosed in Patent Document 1, but the exciting inductor 4c is separated from the transformer 5. There is no description of the advantages. Patent Document 2 only describes that the exciting inductance of the transformer 5 may be used as the exciting inductor 4c, or it may be externally attached. Further, Patent Document 3 discloses a circuit configuration in which the exciting inductor 4c is separated from the transformer 5, but does not describe the advantage that the exciting inductor 4c is separated from the transformer 5.

Japanese Unexamined Patent Publication No. 2011-259560 WO2015137069A1 Publication Japanese Unexamined Patent Publication No. 2016-63745

The conventional LLC resonance converters such as Patent Documents 1 to 3 have the following problems (A) to (C).

(A) In the conventional LLC resonance converter of FIG. 2, when the exciting inductance of the transformer 5 is used as the exciting inductor 4c, it is mainly to secure the desired inductance by adjusting the gap of the transformer core. However, if there is a gap in the transformer core, there is a problem that the magnetic flux swells from the cross section of the gap and interlinks with the windings 5a and 5b, so that the copper loss of the windings 5a and 5b increases.
(B) With respect to the above (A), if the exciting inductor 4c is made independent from the transformer 5, the transformer 5 can have a non-gap, so that there is an advantage that the copper loss due to the load current due to the influence of the gap can be reduced. As a harmful effect, the iron loss of the independent excitation inductor 4c increases, but it becomes effective when the ratio of the copper loss of the load current is large at high power.
(C) When the circuits are parallelized as shown in FIG. 3 in order to cope with a large amount of power, in order to connect the exciting inductors 4c1 and 4c2 independently from the transformer 5A, the current flowing through the exciting inductors 4c1 and 4c2 is also required. It needs to be evenly distributed. Therefore, it is necessary to connect inductors for the number of parallel circuits, which raises the problems of the number of parts used and the increase in the outer shape of the entire LLC resonance converter.

The switching power supply device of the present invention has a switching circuit that switches a DC input voltage and converts it into an AC voltage, a resonance capacitor and a resonance inductor connected in series, and a resonance circuit that resonates with the AC voltage. An exciting inductor comprising a transformer that converts a voltage output from a resonance circuit into a predetermined voltage, a rectifying and smoothing circuit that rectifies and smoothes the predetermined voltage to generate a DC output voltage, and constitutes the resonance circuit. It is connected in parallel between the output side of the transformer and the input side of the rectifying smoothing circuit.

Then, in the switching power supply device , the switching circuit has a plurality of connected switching circuits in which the input voltage is input, respectively, and the resonance circuit having the resonance capacitor and the resonance inductor The transformer has a plurality of resonance circuits connected to each other on the output side of the plurality of switching circuits, and the transformer has a plurality of primary windings connected to the output side of the plurality of resonance circuits and the plurality of 1s. The exciting inductor, which has a single secondary winding electromagnetically coupled to the next winding and further constitutes the resonant circuit , comprises the secondary winding and the input side of the rectifying smoothing circuit. It is characterized by being connected in parallel between them.

According to the switching power supply device of the present invention, it is possible to reduce the number of parts used and the outer shape of the entire switching power supply device without increasing the copper loss and iron loss of the transformer.

Schematic block diagram showing a switching power supply device (for example, an LLC resonant converter in which two circuits are parallelized) according to the first embodiment of the present invention. Schematic block diagram of a conventional LLC resonant converter Schematic block diagram of other conventional LLC resonant converters

The embodiments for carrying out the present invention will become clear when the following description of preferred embodiments is read in light of the accompanying drawings. However, the drawings are for illustration purposes only and do not limit the scope of the present invention.

(Structure of Example 1)
FIG. 1 is a schematic configuration diagram showing a switching power supply device (for example, an LLC resonant converter in which two circuits are parallelized) according to the first embodiment of the present invention.

This LLC resonance converter has a pair of positive and negative input terminals 11a and 11b to which a DC input voltage Vi is supplied, and an input capacitor 12 is connected between the input terminals 11a and 11b. Two half-bridge type switching circuits 20-1 and 20-2 are connected in parallel between both electrodes of the input capacitor 12. The switching circuit 20-1 has two switching elements (for example, MOS type FETs) 21-1,21-2 that operate on / off with each other, and they are connected in series. Similarly, the switching circuit 20-2 also has two switching elements (for example, MOS type FETs) 21-2 and 22-2 that operate on / off with each other, and they are connected in series.

A resonance circuit 30-1 is connected in parallel between the drain and source of the FET 22-1 in the switching circuit 20-1. A resonance circuit 30-2 is also connected in parallel between the drain and source of the FET 22-2 in the switching circuit 20-2. The resonant circuit 30-1 has a resonant capacitor 31-1 and a resonant inductor 32-1, which are connected in series. Similarly, the resonant circuit 30-2 also has a resonant capacitor 31-2 and a resonant inductor 32-2, which are connected in series. Each resonance inductor 32-1 and 32-2 is composed of, for example, a choke coil having a core, and the core of each choke coil is connected to each other.

In the resonant inductor 32-1, the winding start (black circle point side in FIG. 1) of the choke coil is connected to the resonant capacitor 31-1, and at the winding end, the primary winding 41a (turn number n1) in the transformer 40 The beginning of the winding is connected. The winding end of the primary winding 41a is connected to the source of the FET 22-1. Similarly, in the resonant inductor 32-2, the winding start of the choke coil is connected to the resonant capacitor 31-2, and the winding start of the primary winding 41b (turn number n1) in the transformer 40 is connected to the winding end thereof. There is. The end of the primary winding 41b is connected to the source of the FET 22-2.

The transformer 40 includes a plurality of (for example, two) primary windings 41a and 41b and a single secondary winding 42 (number of turns n2) magnetically coupled to these primary windings 41a and 41b. ,have. An inductor 50 is connected in parallel between the winding start and winding end of the secondary winding 42. The inductor 50 is composed of, for example, a choke coil having a core (number of turns N2, impedance Lm2), the winding start of the choke coil is connected to the winding start of the secondary winding 42, and the winding end of the choke coil is connected. It is connected to the end of the secondary winding 42.

A full bridge type rectifier circuit 60 is connected in parallel to the inductor 50. The rectifier circuit 60 is composed of four bridge-connected switching elements for rectification (for example, MOS type FET). An output capacitor 70 constituting a smoothing circuit is connected in parallel to the output side of the rectifier circuit 60, and a pair of positive and negative output terminals 71a and 71b are connected to both electrodes of the output capacitor 70.

In the LLC resonant converter of the first embodiment, the switching circuits 20-1 and 20-2 on the primary side of the transformer 40 and the resonant circuits 30-1 and 30-2 are double-paralleled, and the secondary winding of the transformer 40. The feature is that the inductor 50 is connected in parallel to 42.

(Operation of Example 1)
In the LLC resonance converter of FIG. 1, when a DC input voltage Vi is supplied to the input terminals 11a and 11b, the input voltage Vi is charged to the input capacitor 12 and then two switching circuits 20- connected in parallel. It is applied to 1, 20-2, respectively.

In one switching circuit 20-1, two FETs 21-1 and 22-1 are alternately turned on / off by a switching pulse supplied from a control unit (not shown). When the FET 21-1 is in the ON state, the DC current input to the input terminal 11a flows in the path of FET 21-1 → resonance capacitor 31-1 → resonance inductor 32-1 → primary winding 41a → input terminal 11b. .. Next, when the FET 21-1 is turned off and the FET 22-1 is turned on, the electric charge accumulated in the resonance capacitor 31-1 is charged from the FET 22-1 → the primary winding 41a → the resonance inductor 32-1 →. It flows in the path of the resonance capacitor 31-1. By such on / off operation of FETs 21-1 and 22-1, the input voltage Vi supplied to the input terminals 11a and 11b is converted into an AC voltage, and the AC voltage resonates in the resonance circuit 30-1.

The other switching circuit 20-2 and the resonant circuit 30-2 also perform the same operation in parallel with the one switching circuit 20-1 and the resonant circuit 30-1.

When an AC current flows through the primary windings 41a and 41b of the transformer 40, an AC voltage is induced in the secondary winding 42 of the transformer 40, charges are accumulated in the inductor 50, and the AC voltage is transferred to the rectifying circuit 60. Applied. In the rectifier circuit 60, the two FETs 61 and 62 are alternately turned on / off and the two FETs 63 and 64 are alternately turned off / on by a switching pulse supplied from a control unit (not shown).

When the two FETs 61 and 64 are turned on and the two FETs 62 and 63 are turned off, the winding of the secondary winding 42 starts → the FET 61 in the on state → the output capacitor 70 → the FET 64 in the on state → the secondary winding. A direct current flows through the path at the end of winding of 42. Next, when the two FETs 61 and 64 are turned off and the two FETs 62 and 63 are turned on, the winding end of the secondary winding 42 → the on state FET 63 → the output capacitor 70 → the on state FET 62 → 2 A direct current flows in the path of the start of winding of the next winding 42. In this way, the AC voltage induced in the secondary winding 42 is rectified by the rectifier circuit 60 and smoothed by the output capacitor 70, and then the desired DC output voltage Vo is output from the output terminals 71a and 71b.

(Structural comparison between FIG. 1 of Example 1 and conventional FIG. 3)
In FIG. 3, the number of parallel circuits on the primary side of the transformer is p (= 2), the inductance of the excitation inductors 4c1 and 4c2 inserted in each of the parallelized circuits is Lm1, and the number of turns of the excitation inductors 4c1 and 4c2 is N1. do. Further, the number of turns of the primary windings 5a1 and 5a2 of the transformer 5A is n1, and the number of turns of the secondary winding 5b is n2. On the other hand, in FIG. 1, the number of turns of the primary windings 41a and 41b of the transformer 40 is n1, the number of turns of the single secondary winding 42 of the transformer 40 is n2, and the inductor inserted in the secondary winding 42 thereof. The inductance of 50 is Lm2, and the number of turns of the inductor 50 is N2.

Assuming that the transformer primary winding voltage is Vn1, the voltage application time is Ton, the number of turns of the inductance Lm1 is N1, the magnetic flux density is Bm1, and the core cross-sectional area is Ae1.
Bm1 = (Vn1 * Ton) / (Ae1 * N1) ... (1)
Is true.
Similarly, assuming that the transformer secondary winding voltage is Vn2, the voltage application time is Ton, the number of turns of the inductance Lm2 is N2, the magnetic flux density is Bm2, and the core cross-sectional area is Ae2.
Bm2 = (Vn2 * Ton) / (Ae2 * N2) ... (2)
Is true.

Here, the upper limit values of the core magnetic flux density Bm1 in the exciting inductors 4c1 and 4c2 and the core magnetic flux density Bm2 in the inductor 50 are determined from the utilization rate in consideration of magnetic saturation. Further, the transformer primary winding voltage Vn1, the transformer secondary winding voltage Vn2, and the voltage application time Ton are determined from the conditions of the transformers 40 and 5A. Therefore, the parameters related to the outer shape of the transformers 40 and 5A are
Core cross-sectional area Ae1 * Number of turns N1 of excitation inductors 4c1 and 4c2
Core cross-sectional area Ae2 * Number of turns of inductor 50 N2
Will be.

Comparing the case of parallelizing p (= 2) on the primary side of the transformer 5A and the case of inserting the inductor 50 into the single secondary winding 42 of the transformer 40, as shown in the following equation (3). Become.
(P * Ae1 * N1): (Ae2 * N2)
= P (Vn1 * Ton) / (Bm1): (Vn2 * Ton) / (Bm2)
... (3)

Assuming that the upper limit values of the core magnetic flux density Bm1 in the exciting inductors 4c1 and 4c2 and the core magnetic flux density Bm2 in the inductor 50 are Bm, respectively.
Bm1 = Bm2 = Bm ... (4)
Will be.

From the relationship between the primary turns n1 and the secondary turns n2 of the transformer 40 and the voltages Vn1 and Vn2, the following equation (5) is obtained.
Vn1 / Vn2 = n1 / n2 ... (5)

From the above, equation (3) is
(P * Ae1 * N1): (Ae2 * N2)
= P (Vn1 * Ton) / (Bm1): (Vn2 * Ton) / (Bm2)
= P (Vn1 * Ton) / (Bm):
((N2 / n1) * Vn1 * Ton) / (Bm)
= P: (n2 / n1) ... (6)
Can be expressed as. Therefore, under the condition of p> (n2 / n1), when the inductor 50 is inserted into the single secondary winding 42 side of the transformer 40 as shown in FIG. 1 of the present embodiment 1, the conventional FIG. 3 shows. The size of the inductor 50 can be made smaller than that of the excitation inductors 4c1 and 4c2.

Further, if the excitation inductors 4c1 and 4c2 are suppressed to the outer shape of the inductor 50 corresponding to FIG. 1 in the conventional configuration of FIG. 3, it becomes necessary to reduce the cross-sectional area Ae1 or the number of turns N1 of the parameters Ae1 * N1 related to the outer shape. From the equation (1), the magnetic flux density Bm1 of the core becomes large, which leads to an increase in iron loss and a decrease in the saturation margin.

On the contrary, in the configuration of FIG. 1, if the outer shape equivalent to that of the conventional configuration of FIG. 3 is sufficient, the cross-sectional area Ae2 of the parameter Ae2 * N2 related to the outer shape can be increased by that amount, so that the magnetic flux of the core is obtained from the equation (2). The density Bm2 can be reduced, leading to suppression of iron loss and expansion of the magnetic saturation margin. Further, when the magnetic flux density Bm2 is not reduced, the number of turns N2 can be reduced by the amount that the cross-sectional area Ae2 can be increased, so that copper loss can be suppressed.

(Effect of Example 1)
Since the inductor 50 is connected to the single secondary winding 42 side of the transformer 40, it is possible to suppress the number of parts used and the increase in the outer shape of the entire LLC resonance converter as compared with the conventional case.

(Variation example of Example 1)
The present invention is not limited to the above-mentioned Example 1, and various usage forms and modifications are possible. Examples of this usage pattern and modification include the following (a) to (d).

(A) The resonance inductors 32-1 and 32-2 may be configured by the leakage inductance of the transformer 40, respectively. This makes it possible to make the outer shape of the entire switching power supply device smaller.
(B) The switching circuits 20-1 and 20-2 have a half-bridge type configuration, but even if they are transformed into a full-bridge type configuration, they may have substantially the same effects as those in the first embodiment. can.
(C) The rectifier circuit 60 has a bridge configuration using a plurality of FETs 61 to 64, but is not limited thereto. For example, as the plurality of FETs 61 to 64, other switching elements such as an insulated gate bipolar transistor (IGBT) may be used. Further, even if the rectifier circuit 60 is transformed into a configuration using a plurality of rectifying diodes or a configuration using a plurality of rectifying switch elements and rectifying diodes, the operation and effect substantially the same as those in the first embodiment can be obtained. Can play.
(D) In the first embodiment, the LLC resonance converter has been described as the switching power supply device, but the present invention can also be applied to the resonance type converter having other configurations.

20-1, 20-2 switching circuit
21-1,21-2,22-1,22-2 FET
30-1, 30-2 Resonant circuit
31-1 and 31-2 Resonant capacitors
32-1, 32-2 resonant inductor
40 transformer
41a, 41b primary winding
42 Secondary winding
50 inductor
60 Rectifier circuit
70 output capacitor

Claims (4)

A switching circuit that switches the DC input voltage and converts it to an AC voltage,
A resonant circuit having a resonant capacitor and a resonant inductor connected in series and resonating with the AC voltage,
A transformer that converts the voltage output from the resonance circuit into a predetermined voltage, and
A rectifying and smoothing circuit that rectifies and smoothes the predetermined voltage to generate a DC output voltage.
In a switching power supply device in which the exciting inductor constituting the resonant circuit is connected in parallel between the output side of the transformer and the input side of the rectifying smoothing circuit.
The switching circuit has a plurality of connected switching circuits in which the input voltage is input.
The resonance circuit having the resonance capacitor and the resonance inductor has a plurality of resonance circuits connected to each other on the output side of the plurality of switching circuits.
The transformer has a plurality of primary windings connected to each output side of the plurality of resonant circuits and a single secondary winding electromagnetically coupled to the plurality of primary windings. ,
A switching power supply device characterized in that the exciting inductor constituting the resonance circuit is connected in parallel between the secondary winding and the input side of the rectifying smoothing circuit. The resonant inductor is
The switching power supply device according to claim 1, wherein the switching power supply device is configured by the leakage inductance of the transformer. The switching circuit is
The switching power supply device according to claim 1 or 2, wherein the switching circuit is a half-bridge type or a full-bridge type switching circuit having a plurality of switching elements. The rectifying smoothing circuit
A rectifying circuit having a plurality of rectifying diodes and / or a plurality of rectifying switching elements,
A smoothing circuit that smoothes the voltage output from the rectifier circuit to generate the output voltage,
The switching power supply device according to any one of claims 1 to 3, wherein the switching power supply device comprises.

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