JP6999387B2 - Power converter - Google Patents

Description

The present invention relates to a power conversion device, and more particularly to a power conversion device that performs soft switching of a semiconductor switching element.

In recent years, with the increase in capacity and high frequency switching of power converters, the increase in switching loss and electromagnetic interference (EMI) noise of semiconductor switching elements in the main circuit has become a big problem. A resonance type power conversion device is used as a method for reducing switching loss and EMI noise, and various circuit configurations have been proposed.

As a typical example of the resonance type power conversion device, there is an auxiliary resonance commutation pole type power conversion device in which an auxiliary resonance circuit is provided in the power conversion device and the semiconductor switching element is soft-switched by the resonance operation of the auxiliary resonance circuit. For example, see Japanese Patent Application Laid-Open No. 2011-55580 (Patent Document 1).

Japanese Unexamined Patent Publication No. 2011-55580

The resonance type power conversion device described in Patent Document 1 is an inverter having a full bridge configuration in which a plurality of half-bridge circuits composed of two semiconductor switching elements connected in series are connected in parallel. In Patent Document 1, soft switching of the semiconductor switching element constituting each half-bridge circuit is realized by providing an auxiliary resonance circuit for each of the plurality of half-bridge circuits.

However, it is necessary to provide a plurality of auxiliary resonance circuits so as to correspond to each of the plurality of half-bridge circuits, and there is a concern that the configuration of the power conversion device becomes complicated and large. The present invention has been made to solve such a problem, and an object thereof is a switching loss of a semiconductor switching element with a simple configuration in a power conversion device that converts power between DC power and AC power. And to provide a new configuration that can reduce EMI noise.

The power conversion device according to the present invention is a power conversion device that performs power conversion between DC power and AC power, and is a high-potential side DC terminal, a low-potential side DC terminal, a high-potential side DC terminal, and a first. A first switching element connected between the nodes, a second switching element connected between the first node and the low potential side DC terminal, and between the high potential side DC terminal and the second node. A connected third switching element, a fourth switching element connected between the second node and the low potential side DC terminal, and a first AC terminal electrically connected to the first node. A second AC terminal electrically connected to the second node and a control circuit for controlling the on / off of the first to fourth switching elements are provided. The control circuit turns the third and fourth switching elements on and off in synchronization with the AC frequency, and turns the first and second switching elements on and off at a frequency higher than the AC frequency.

Preferably, the power converter further comprises a first capacitor connected in parallel to the first switching element, a second capacitor connected in parallel to the second switching element, and an auxiliary resonant circuit. The auxiliary resonant circuit includes a reactor and is configured to soft-switch the first and second switching elements by the resonant operation of the first and second capacitors and the reactor.

Preferably, the auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal. The reactor is electrically connected between the connection node of the first and second auxiliary switching elements and the first node.

Preferably, the auxiliary resonant circuit further comprises third and fourth capacitors electrically connected in series between the high potential side DC terminal and the low potential side DC terminal. The reactor is electrically connected between the connection node of the third and fourth capacitors and the first node. The auxiliary resonant circuit further includes a bidirectional switch electrically connected in series with the reactor between the connecting nodes of the third and fourth capacitors and the first node.

Preferably, the auxiliary resonance circuit is the primary winding electrically connected between the high potential side DC terminal and the low potential side DC terminal, and the reactor and the electrical between the first node and the low potential side DC terminal. A first transformer with a secondary winding connected in series with, a first auxiliary switching element electrically connected between the primary winding and the low potential side DC terminal, and the secondary winding and It includes a second auxiliary switching element electrically connected between the low potential side DC terminals.

Preferably, the winding ratio of the primary winding and the secondary winding of the first transformer is 2: 1.
Preferably, the first transformer has a reset winding electrically connected between the high potential side DC terminal and the low potential side DC terminal. The auxiliary resonant circuit was electrically connected in series with the reset winding between the high potential side DC terminal and the low potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. Includes more diodes.

Preferably, the auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal. The reactor is electrically connected between the connection node of the first and second auxiliary switching elements and the first node. The auxiliary resonance circuit is the first and first electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. The third and third electrical connections are electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. It further includes a diode of 4 and a second transformer. The second transformer is the primary winding electrically connected in series with the reactor between the connecting node of the first and second auxiliary switching elements and the first node, and the first and second diodes. It has a secondary winding electrically connected between the connection node and the connection node of the third and fourth diodes.

Preferably, the winding ratio of the primary winding and the secondary winding of the second transformer is 1: 2.
Preferably, the auxiliary resonant circuit further comprises a fifth diode and a sixth diode. The fifth diode is electrically connected in series with the first auxiliary switching element between the high potential side DC terminal and the connection node of the first and second auxiliary switching elements. The sixth diode is electrically connected in series with the second auxiliary switching element between the low potential side DC terminal and the connection node of the first and second auxiliary switching elements.

Preferably, each of the first and second switching elements has a first power semiconductor switching element and a first freewheeling diode connected in antiparallel to the first power semiconductor switching element. Each of the third and fourth switching elements has a second power semiconductor switching element and a second freewheeling diode connected in antiparallel to the second power semiconductor switching element. The first freewheeling diode has a shorter reverse recovery time than the second freewheeling diode.

Preferably, the first freewheeling diode is formed of a wide bandgap semiconductor.

Preferably, the control circuit generates a control signal for controlling the on / off of the first and second switching elements by pulse width modulation control.

The power conversion device according to the present invention includes a high potential side DC terminal and a low potential side DC terminal, a first switching element electrically connected between the high potential side DC terminal and the first node, and a first unit. A second switching element electrically connected between the node and the low potential side DC terminal, an AC terminal electrically connected to the first node, and a first switching element connected in parallel. The first capacitor, the second capacitor connected in parallel to the second switching element, and the reactor are included, and the first and second switching elements are made by the resonance operation between the first and second capacitors and the reactor. It includes an auxiliary resonance circuit configured for soft switching. The auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal. The reactor is electrically connected between the connection node of the first and second auxiliary switching elements and the first node.

The power conversion device according to the present invention includes a high potential side DC terminal and a low potential side DC terminal, a first switching element electrically connected between the high potential side DC terminal and the first node, and a first unit. A second switching element electrically connected between the node and the low potential side DC terminal, an AC terminal electrically connected to the first node, and a first switching element connected in parallel. The first capacitor, the second capacitor connected in parallel to the second switching element, and the reactor are included, and the first and second switching elements are made by the resonance operation between the first and second capacitors and the reactor. It includes an auxiliary resonance circuit configured for soft switching. The auxiliary resonance circuit is electrically connected to the primary winding between the high potential side DC terminal and the low potential side DC terminal, and electrically in series with the reactor between the first node and the low potential side DC terminal. A first transformer with a connected secondary winding, a first auxiliary switching element electrically connected between the primary winding and the low potential side DC terminal, and the secondary winding and low potential side. It includes a second auxiliary switching element electrically connected between the DC terminals.

Preferably, the winding ratio of the primary winding and the secondary winding of the first transformer is 2: 1.
Preferably, the first transformer has a reset winding electrically connected between the high potential side DC terminal and the low potential side DC terminal. The auxiliary resonant circuit was electrically connected in series with the reset winding between the high potential side DC terminal and the low potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. Includes more diodes.

The power conversion device according to the present invention includes a high potential side DC terminal and a low potential side DC terminal, a first switching element electrically connected between the high potential side DC terminal and the first node, and a first unit. A second switching element electrically connected between the node and the low potential side DC terminal, an AC terminal electrically connected to the first node, and a first switching element connected in parallel. The first capacitor, the second capacitor connected in parallel to the second switching element, and the reactor are included, and the first and second switching elements are made by the resonance operation between the first and second capacitors and the reactor. It includes an auxiliary resonance circuit configured for soft switching. The auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal. The reactor is electrically connected between the connection node of the first and second auxiliary switching elements and the first node. The auxiliary resonance circuit is the first and first electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. The third and third electrical connections are electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. Connection of the first and second diodes and the primary winding electrically connected in series with the reactor between the diode 4 and the connection nodes of the first and second auxiliary switching elements and the first node. Further included is a second transformer having a secondary winding electrically connected between the node and the connection node of the third and fourth diodes.

Preferably, the winding ratio of the primary winding and the secondary winding of the second transformer is 1: 2.
Preferably, the auxiliary resonance circuit is a fifth diode electrically connected in series with the first auxiliary switching element between the high potential side DC terminal and the connection node of the first and second auxiliary switching elements. And a sixth diode electrically connected in series with the second auxiliary switching element between the low potential side DC terminal and the connection node of the first and second auxiliary switching elements.

According to the present invention, in a power conversion device that converts power between DC power and AC power, switching loss and EMI noise of a semiconductor switching element can be reduced with a simple configuration.

It is a main circuit block diagram of the power conversion apparatus according to Embodiment 1 of this invention. It is a figure explaining the reverse conversion operation of the power conversion apparatus according to Embodiment 1. FIG. It is a figure explaining the reverse conversion operation of the power conversion apparatus according to Embodiment 1. FIG. It is a figure explaining the forward conversion operation of the power conversion apparatus according to Embodiment 1. FIG. It is a figure explaining the forward conversion operation of the power conversion apparatus according to Embodiment 1. FIG. It is an operation waveform diagram for demonstrating the on / off control of a switch in an asymmetric drive system. It is a figure explaining the reverse conversion operation of the power conversion apparatus by a symmetric drive system. It is a figure explaining the reverse conversion operation of the power conversion apparatus by a symmetric drive system. It is operation waveform diagram for demonstrating on / off control of a switch in a symmetric drive system. It is a main circuit block diagram of the power conversion apparatus according to Embodiment 2 of this invention. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing through the switch S2 to the switch S1. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing to the switch S2 to the switch S1. 11 is an operation waveform diagram when the operations shown in FIGS. 11 and 12 are executed. It is a figure which shows the frequency spectrum of the operation waveform of the drain-source voltage and the drain current of a switching element. It is a main circuit block diagram of the power conversion apparatus according to Embodiment 3 of this invention. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing through the switch S2 to the switch S1. It is an operation waveform diagram when the operation shown in FIG. 16 is executed. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing to the switch S1 to the switch S2. It is an operation waveform diagram when the operation shown in FIG. 18 is executed. It is a main circuit block diagram of the power conversion apparatus according to Embodiment 4 of this invention. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing through the switch S2 to the switch S1. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing through the switch S2 to the switch S1. 21 is an operation waveform diagram when the operations shown in FIGS. 21 and 22 are executed. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing to the switch S1 to the switch S2. It is a figure explaining the soft switching at the time of commutation from the state where the current is flowing to the switch S1 to the switch S2. It is operation waveform diagram when the operation shown in FIG. 24 and FIG. 25 is executed. It is a main circuit block diagram of the power conversion apparatus according to the modification of Embodiment 4. It is a main circuit block diagram of the power conversion apparatus according to Embodiment 5 of this invention. It is a figure explaining the operation of the power conversion apparatus according to the 5th Embodiment of this invention. It is a figure explaining the operation of the power conversion apparatus according to the 5th Embodiment of this invention. FIG. 3 is a main circuit configuration diagram showing another configuration example of the power conversion device to which the auxiliary resonance circuit shown in FIG. 15 is applied. FIG. 3 is a main circuit configuration diagram showing another configuration example of the power conversion device to which the auxiliary resonance circuit shown in FIG. 20 is applied. FIG. 3 is a main circuit configuration diagram showing another configuration example of the power conversion device to which the auxiliary resonance circuit shown in FIG. 27 is applied. FIG. 8 is a main circuit configuration diagram showing another configuration example of the power conversion device to which the auxiliary resonance circuit shown in FIG. 28 is applied.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the following, the same or corresponding parts in the figure are designated by the same reference numerals, and the explanations are not repeated in principle.

[Embodiment 1]
FIG. 1 is a main circuit configuration diagram of a power conversion device 1 according to the first embodiment of the present invention. The power conversion device 1 according to the first embodiment is an AC / DC converter that performs bidirectional power conversion between DC power and AC power.

With reference to FIG. 1, the power conversion device 1 according to the first embodiment includes DC terminals T1 and T2, AC terminals T3 and T4, and a power semiconductor switching element (hereinafter, also simply referred to as “switching element”) Q1. -Q4, capacitors C0 and C5, reactor L1 and a control circuit 5 are provided.

The DC terminal T1 (high potential side DC terminal) is electrically connected to the positive terminal of the DC power supply P1, and the DC terminal T2 (low potential side DC terminal) is electrically connected to the negative terminal of the DC power supply P1. The DC positive bus PL1 is connected to the DC terminal T1, and the DC negative bus NL1 is connected to the DC terminal T2. An AC system P2 is electrically connected between the AC terminals T3 and T4. As used herein, the term "electrically connected" refers to a connection state in which electrical energy can be transmitted by direct connection or connection via other elements.

The switching element Q1 is electrically connected between the DC positive bus PL1 (that is, the DC terminal T1) and the node a. The switching element Q2 is electrically connected between the node a and the DC negative bus NL1 (that is, the DC terminal T2). The node a is electrically connected to the AC terminal T3. The node a corresponds to the "first node", and the AC terminal T3 corresponds to the "first AC terminal".

The switching element Q3 is electrically connected between the DC positive bus PL1 and the node b. The switching element Q4 is electrically connected between the node b and the DC negative bus NL1. The node b is electrically connected to the AC terminal T4. The node b corresponds to the "second node", and the AC terminal T4 corresponds to the "second AC terminal".

Although MOSFET (Metal Oxide Semiconductor Field Effect Transistor) is used as the switching element in FIG. 1, any self-extinguishing type switching element such as an IGBT (Insulated Gate Bipolar Transistor) can be used. Diodes D1 to D4 are connected in antiparallel to each of the switching elements Q1 to Q4, respectively. Each of the diodes D1 to D4 is provided to carry a freewheel current when the corresponding switching element is off. When the switching element is a MOSFET, the freewheel diode may be composed of a parasitic diode (body diode). When the switching element is an IGBT without a built-in diode, the freewheel diode is composed of diodes connected in antiparallel to the IGBT.

The switching elements Q1 and Q2 constitute a "first half-bridge circuit A". The switching elements Q3 and Q4 form a "second half-bridge circuit B". The first half-bridge circuit A and the second half-bridge circuit B are connected in parallel to each other between the DC positive bus PL1 and the DC negative bus NL1. That is, the power conversion device 1 has a full bridge circuit.

The capacitor C0 is connected between the DC positive bus PL1 and the DC negative bus NL1 and smoothes the voltage between the DC terminals T1 and T2 (that is, the output voltage of the DC power supply P1).

The reactor L1 is connected between the node a (first node) and the AC terminal T3. The capacitor C5 is connected between the AC terminals T3 and T4. Smooth the voltage between the AC terminals T3 and T4. The reactor L1 and the capacitor C5 constitute a filter for removing a component of the switching frequency included in the AC power output from the full bridge circuit.

The control circuit 5 controls the on / off of the switching element in each of the first half-bridge circuit A and the second half-bridge circuit. The control circuit 5 outputs gate signals G1 to G4 for turning on / off the switching elements Q1 to Q4, respectively. Each of the switching elements Q1 to Q4 is turned on in response to the H (logical high) level gate signal G and turned off in response to the L (logical low) level gate signal G. As a result, the power conversion device 1 converts the DC power supplied from the DC power supply P1 into AC power and outputs it from the AC terminals T3 and T4, and the AC power supplied from the AC system P2 is converted to DC power. It is possible to execute a forward conversion operation of converting to DC and outputting from DC terminals T1 and T2.

Next, the inverse conversion operation of the power conversion device 1 will be described with reference to FIGS. 2 and 3. In the reverse conversion operation, the power conversion device 1 converts the DC power of the DC power supply P1 into AC power and supplies it to the load 2. In the following description, the output voltage (power supply voltage) of the DC power supply P1 is Vdc, the voltage applied to the load 2 (or load 3) is Vout, and the voltage applied to the reactor L1 is VL.

First, when the switching elements Q1 and Q4 are turned on, the circuit state shown in FIG. 2A is obtained. Since the potential Va = Vdc of the node a and the potential Vb = 0 of the node b, the voltage VL = Va-Vout-Vb = Vdc-Vout is added to the reactor L1, and the current is as shown by the arrow in the figure. It flows in a gradually increasing form. The direction of the voltage / current at this time is represented as positive (+), and the opposite case is represented as negative (-).

Next, when the switching element Q1 is turned off and the switching element Q2 is turned on, that is, when the switching elements Q2 and Q4 are turned on, the circuit state shown in FIG. 2B is obtained. The current passing through the reactor L1 cannot suddenly change its direction and magnitude due to the current maintaining action of the reactor, and continues to flow in the positive direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is applied to the reactor L1 and the current gradually decreases.

By alternately repeating the circuit state of FIG. 2A and the circuit state of FIG. 2B, a positive voltage is applied to the load 2. The magnitude of this positive voltage depends on the ratio of the on period of the switching element Q1 to the switching cycle in which the switching elements Q1 and Q2 are turned on and off, and can ideally be changed between 0 and Vdc. The switching element Q4 is fixed to ON in the switching cycle of the switching elements Q1 and Q2.

Next, when the switching elements Q2 and Q3 are turned on, the circuit state shown in FIG. 3A is obtained. Since the potential Vb = Vdc of the node b and the potential Va = 0 of the node a, the voltage VL = Va-Vout-Vb = -Vdc-Vout is added to the reactor L1, and as shown by the arrow in the figure, it is negative. The current flows in a form in which the absolute value gradually increases.

Next, when the switching elements Q1 and Q3 are turned on, the circuit state shown in FIG. 3B is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintenance action, and continues to flow in the negative direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is added to the reactor L1, and the absolute value of the current gradually decreases.

By alternately repeating the circuit state of FIG. 3A and the circuit state of FIG. 3B, a negative voltage is applied to the load 2. The magnitude of this negative voltage depends on the ratio of the on period of the switching element Q2 to the switching cycle in which the switching elements Q1 and Q2 are turned on and off, and can ideally be changed between 0 and −Vdc. The switching element Q3 is fixed to ON in the switching cycle of the switching elements Q1 and Q2.

In the power conversion device 1, the switching operation of FIG. 2 and the switching operation of FIG. 3 are alternately executed every 1/2 cycle in synchronization with the frequency (AC frequency) of the AC power output from the AC terminals T3 and T4. Thereby, the DC power is converted into the AC power of the frequency. The on / off control of the switching elements Q1 and Q2 is executed according to the pulse width modulation (PWM) control, as will be described later. On the other hand, the on / off control of the switching elements Q3 and Q4 is executed at the AC frequency.

Next, the forward conversion operation of the power conversion device 1 will be described with reference to FIGS. 4 and 5. In the forward conversion operation, the power conversion device 1 converts the AC power of the AC system P2 into DC power and supplies it to the load 3 (including the DC power supply P1). In the following description, the output voltage (power supply voltage) of the AC system P2 is V2. FIG. 4 shows a circuit state when the output voltage of the AC system P2 is positive, and FIG. 5 shows a circuit state when the output voltage of the AC system P2 is negative.

With reference to FIG. 4, when the switching elements Q2 and Q4 are turned on when the output voltage of the AC system P2 is positive, the circuit state shown in FIG. 4A is obtained. The series circuit of the reactor L1 and the AC system P2 is short-circuited, and a current flows as shown by an arrow in the figure. The direction of the current at this time is represented as positive (+), and the opposite case is represented as negative (-).

Next, when the switching element Q1 is turned on and the switching element Q2 is turned off, that is, when the switching elements Q1 and Q4 are turned on, the circuit state shown in FIG. 4B is obtained. The current passing through the reactor L1 cannot suddenly change its direction and magnitude due to the current maintaining action of the reactor, and continues to flow in the positive direction. The potential Va = Vout of the node a.

By alternately repeating the circuit state of FIG. 4A and the circuit state of FIG. 4B, a positive voltage is applied to the load 3. The magnitude of this positive voltage depends on the ratio of the on period of the switching element Q2 to the switching cycle in which the switching elements Q1 and Q2 are turned on and off, and can ideally be changed between V2 and ∞. The switching element Q4 is fixed to ON in the switching cycle of the switching elements Q1 and Q2.

With reference to FIG. 5, when the switching elements Q1 and Q3 are turned on when the output voltage of the AC system P2 is negative, the circuit state shown in FIG. 5A is obtained. The series circuit of the reactor L1 and the AC system P2 is short-circuited, and a current flows in the negative direction as shown by the arrow in the figure.

Next, when the switching element Q1 is turned off and the switching element Q2 is turned on, that is, when the switching elements Q2 and Q3 are turned on, the circuit state shown in FIG. 5B is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintenance action, and continues to flow in the negative direction. The potential Va = 0 of the node a, and the potential Vb = Vout of the node b.

By alternately repeating the circuit state of FIG. 5A and the circuit state of FIG. 5B, a positive voltage is applied to the load 3. The magnitude of this positive voltage depends on the ratio of the on period of the switching element Q1 to the switching cycle in which the switching elements Q1 and Q2 are turned on and off, and can ideally be changed between V2 and ∞. The switching element Q3 is fixed to ON in the switching cycle of the switching elements Q1 and Q2.

In the power conversion device 1, the switching operation of FIG. 4 and the switching operation of FIG. 5 are alternately executed every 1/2 cycle in synchronization with the frequency of the AC power input to the AC terminals T3 and T4. AC power is converted to DC power. The on / off control of the switching elements Q1 and Q2 is executed according to the PWM control as described later. On the other hand, the on / off control of the switching elements Q3 and Q4 is executed at the AC frequency.

FIG. 6 is an operation waveform diagram for explaining on / off control of the switching elements Q1 to Q4 at the time of the reverse conversion operation of the power conversion device 1. FIG. 6 shows the operation waveforms of the switching elements Q1 to Q4 and the operation waveform of the voltage Vab between the nodes ab.

With reference to FIG. 6, the control circuit 5 (FIG. 1) of the power converter 1 compares the voltage of the periodic carrier wave CW (triangle wave or sawtooth wave) with the voltage of the sinusoidal signal V * (modulated wave). Then, based on the comparison result, the gate signals G1 and G2 are output to the switching elements Q1 and Q2, respectively. Two types of carrier wave CW are prepared according to the polarity of the sinusoidal signal V *.

Specifically, when the potential of the sinusoidal signal V * is higher than the potential of the carrier wave CW, the switching element Q1 is turned on and the switching element Q2 is turned off. On the other hand, when the potential of the sinusoidal signal V * is lower than the potential of the carrier wave CW, the switching element Q1 is turned off and the switching element Q2 is turned on. That is, the switching elements Q1 and Q2 are controlled on and off by the frequency of the carrier wave CW (hereinafter, also referred to as carrier frequency).

On the other hand, the switching elements Q3 and Q4 are controlled on and off by the frequency of the sinusoidal signal V *. Specifically, during the period when the voltage of the sinusoidal signal V * is positive, the switching element Q3 is fixed off and the switching element Q4 is fixed on. During the period when the voltage of the sinusoidal signal V * is negative, the switching element Q3 is fixed on and the switching element Q4 is fixed off.

As described above, in the power conversion device according to the first embodiment, the first half-bridge circuit A (switching elements Q1 and Q2) is turned on and off at the carrier frequency, and the second half-bridge circuit B (switching elements Q3 and Q4) is turned on and off. ) Is turned on and off at the frequency of the AC power. Such a drive system is also called an "asymmetric drive system".

Here, as the drive method of the power conversion device, in addition to the asymmetric drive method, there is a method in which both the first half-bridge circuit A and the second half-bridge circuit B are turned on and off at the carrier frequency. Such a drive system is also called a "symmetric drive system". The inverse conversion operation in the symmetric drive system will be described with reference to FIGS. 7 and 8.

First, when the switching elements Q1 and Q4 are turned on, the circuit state shown in FIG. 7A is obtained. Since the potential Va = Vdc of the node a and the potential Vb = 0 of the node b, the voltage VL = Va-Vout-Vb = Vdc-Vout is added to the reactor L1, and as shown by the arrow in the figure, it is positive. The current flows in a gradually increasing form.

Next, when the switching element Q1 is turned off and the switching element Q2 is turned on, that is, when the switching elements Q2 and Q4 are turned on, the circuit state shown in FIG. 7B is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintaining action of the reactor, and continues to flow in the positive direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is applied to the reactor L1 and the current gradually decreases.

Next, when the switching element Q1 is turned on and the switching element Q2 is turned off, that is, when the switching elements Q1 and Q4 are turned on, the circuit state shown in FIG. 7C is obtained. Since the potential Va = Vdc of the node a and the potential Vb = 0 of the node b, the voltage VL = Va-Vout-Vb = Vdc-Vout is added to the reactor L1, and as shown by the arrow in the figure, it is positive. The current flows in a gradually increasing form.

Next, when the switching element Q3 is turned on and the switching element Q4 is turned off, that is, when the switching elements Q1 and Q3 are turned on, the circuit state shown in FIG. 7D is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintaining action of the reactor, and continues to flow in the positive direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is applied to the reactor L1 and the current gradually decreases.

By repeating the circuit states of FIGS. 7A to 7D, a positive voltage is applied to the load 2. The switching cycle for turning on / off the switching elements Q1 and Q2 and the switching cycle for turning on / off the switching elements Q3 and Q4 are equal to each other.

Next, when the switching elements Q2 and Q3 are turned on, the circuit state shown in FIG. 8A is obtained. Since the potential Vb = Vdc of the node b and the potential Va = 0 of the node a, the voltage VL = Va-Vout-Vb = -Vdc-Vout is added to the reactor L1, and as shown by the arrow in the figure, it is negative. The current flows in a form in which the absolute value gradually increases.

Next, when the switching element Q3 is turned off and the switching element Q4 is turned on, that is, when the switching elements Q2 and Q4 are turned on, the circuit state shown in FIG. 8B is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintenance action, and continues to flow in the negative direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is added to the reactor L1, and the absolute value of the current gradually decreases.

Next, when the switching element Q3 is turned on and the switching element Q4 is turned off, that is, when the switching elements Q2 and Q3 are turned on, the circuit state shown in FIG. 8C is obtained. Since the potential Vb = Vdc of the node b and the potential Va = 0 of the node a, the voltage VL = Va-Vout-Vb = -Vdc-Vout is added to the reactor L1, and as shown by the arrow in the figure, it is negative. The current flows in a form in which the absolute value gradually increases.

Next, when the switching element Q1 is turned on and the switching element Q2 is turned off, that is, when the switching elements Q1 and Q3 are turned on, the circuit state shown in FIG. 8D is obtained. The current passing through the reactor L1 cannot be immediately reduced to 0 due to the current maintenance action, and continues to flow in the negative direction. Since the series circuit of the reactor L1 and the load 2 is short-circuited and the voltage Vab = 0, the voltage VL = −Vout is added to the reactor L1, and the absolute value of the current gradually decreases.

The circuit states of FIGS. 8A to 8D are repeated, and a negative voltage is applied to the load 2. The switching cycle for turning on / off the switching elements Q1 and Q2 and the switching cycle for turning on / off the switching elements Q3 and Q4 are equal to each other.

FIG. 9 is an operation waveform diagram for explaining on / off control of the switching elements Q1 to Q4 in the symmetrical drive system. FIG. 9 shows the operation waveforms of the switching elements Q1 to Q4 and the operation waveform of the voltage Vab between the nodes ab.

With reference to FIG. 9, the voltage of the carrier wave CW and the voltage of the sine wave signal V * (modulated wave) are compared, and based on the comparison result, the gate signals G1 and G2 are assigned to the switching elements Q1 and Q2, respectively. It is output. Further, the voltage of the carrier wave CW and the voltage of the signal / V * in which the sine wave signal V * is inverted are compared, and the gate signals G3 and G4 are output to the switching elements Q3 and Q4 based on the comparison result, respectively. ..

Specifically, when the potential of the sinusoidal signal V * is higher than the potential of the carrier wave CW, the switching element Q1 is turned on and the switching element Q2 is turned off. On the other hand, when the potential of the sinusoidal signal V * is lower than the potential of the carrier wave CW, the switching element Q1 is turned off and the switching element Q2 is turned on. That is, the switching elements Q1 and Q2 are controlled on and off by the carrier frequency.

When the potential of the inverting signal / V * is higher than the potential of the carrier wave CW, the switching element Q3 is turned on and the switching element Q4 is turned off. On the other hand, when the potential of the inverting signal / V * is lower than the potential of the carrier wave CW, the switching element Q3 is turned off and the switching element Q4 is turned on. That is, the switching elements Q3 and Q4 are controlled on and off by the carrier frequency.

Comparing the operation waveform diagram of FIG. 6 with the operation waveform diagram of FIG. 9, in the asymmetric drive method, one of the two half-bridge circuits (second half-bridge circuit B in FIG. 6) has an AC frequency lower than the carrier frequency. Since it is turned on and off, the number of switchings of one half-bridge circuit can be reduced as compared with the symmetric drive system in which the two half-bridge circuits are both turned on and off at the carrier frequency. Since the reduction in the number of switching leads to the reduction of the switching loss generated in the half-bridge circuit, it can be expected that the power conversion efficiency of the entire power conversion device can be improved by using the optimum element.

In the asymmetric drive method, the switching elements constituting the half-bridge circuit can be used properly according to the switching frequency of the half-bridge circuit.

For example, for a half-bridge circuit (first half-bridge circuit A) in which the switching frequency is a carrier frequency, a wide bandgap semiconductor material capable of high-frequency operation (for example, SiC (silicon carbide), GaN (gallium nitride), etc.) is used. Switching elements formed in use (eg, SiC- MOSFETs, GaN-HEMTs, etc.) can be used.

In the power conversion operation, there is a mode in which a current flows not only through the channel of the switching element but also through the freewheel diode. When a current is passed through or cut off from a freewheel diode, a current flows instantaneously in the opposite direction of the diode due to the reverse recovery phenomenon. The longer the time for the current to flow in the opposite direction, the greater the loss that occurs during that time. Therefore, it is desirable that the switching element whose switching frequency is the carrier frequency has a freewheel diode (including a body diode) having a fast reverse recovery time. Preferably, the reverse recovery time is 200 ns or less. For example, a SiC Schottky barrier diode or an FRD (First Recovery Diode) can be connected to the IGBT in antiparallel. Alternatively, a high speed body diode type MOSFET can be used.

On the other hand, in the half-bridge circuit (second half-bridge circuit B) in which the switching frequency is the frequency of AC power, the conduction loss can be reduced while ensuring a high withstand voltage as compared with the planar type Si MOSFET. , Si Super Junction MOSFETs can be used.

Furthermore, by adopting the asymmetric drive method, the switching loss of the half-bridge circuit (second half-bridge circuit B) that switches at the frequency of AC power can be reduced, so that the half-bridge circuit that switches at the carrier frequency (first half bridge) By providing the auxiliary resonance circuit only in the circuit A), soft switching of the switching element can be realized. As a result, soft switching can be executed with a simple configuration, so that it is possible to realize high efficiency and low noise of the power conversion device.

[Embodiment 2]
In the following embodiments 2 to 5, the configuration for realizing soft switching in the asymmetrical drive system power conversion device 1 described in the first embodiment will be described.

FIG. 10 is a main circuit configuration diagram of the power conversion device 1 according to the second embodiment of the present invention. With reference to FIG. 10, the circuit configuration of the power conversion device 1 according to the second embodiment is basically the same as the circuit configuration of the power conversion device 1 shown in FIG. 1, and the auxiliary resonance circuit 4 and the auxiliary capacitors Cr1 and The difference is that Cr2 is provided.

The auxiliary capacitor Cr1 is connected in parallel to the switching element Q1. The auxiliary capacitor Cr2 is connected in parallel to the switching element Q2. As the auxiliary capacitors Cr1 and Cr2, an external capacitor may be used, or a parasitic capacitance of a switching element located at an equivalent position in the circuit configuration may be used.

The auxiliary resonance circuit 4 includes capacitors Cp and Cn, auxiliary switching elements Qr1 and Qr2, and an auxiliary reactor Lr. The capacitors Cp and Cn have the same capacitance and are connected in series between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The capacitors Cp and Cn are capacitors for dividing the voltage V1 of the DC power supply P1 into two. As the capacitors Cp and Cn, electrolytic capacitors having a large capacitance are generally used in order to stabilize the fluctuation of the voltage.

The auxiliary switching elements Qr1 and Qr2 are electrically connected in series between the connection point (node n) of the capacitors Cp and Cn and the node a. The on / off of the auxiliary switching elements Qr1 and Qr2 is controlled by the gate signals Gr1 and Gr2 output from the control circuit 5, respectively. Although MOSFETs are used as auxiliary switching elements Qr1 and Qr2 in FIG. 10, any self-extinguishing type switching element such as an IGBT can be used. Auxiliary diodes Dr1 and Dr2 are connected in anti-parallel to the auxiliary switching elements Qr1 and Qr2, respectively. The auxiliary switching elements Qr1 and Qr2 and the auxiliary diodes Dr1 and Dr2 correspond to one embodiment of the "bidirectional switch".

The auxiliary reactor Lr is electrically connected between the series circuit of the auxiliary switching elements Qr1 and Qr2 and the node a.

Next, the operation of the power conversion device 1 according to the second embodiment, particularly the soft switching by the auxiliary resonance circuit 4, will be described with reference to FIGS. 11 to 13. Here, soft switching when a positive voltage is applied to the load 2 (see FIG. 2) will be described.

In the following description, the series circuit of the auxiliary switching elements Qr1 and Qr2 in FIG. 10 will be replaced with the auxiliary switch Sr. The output voltage (power supply voltage) of the DC power supply P1 is Vdc. Io indicates the current flowing through the load 2, IL1 indicates the current flowing through the reactor L1, and ILr indicates the reactor current flowing through the auxiliary reactor Lr.

11 and 12 show a state in which the current IL1 is flowing through the diode D2 of the switching element Q2 (mode M0) and is transferred to the switching element Q1. FIG. 12 is an operation waveform diagram when the operation shown in FIG. 11 is executed.

With reference to FIG. 11, when the switching elements Q2 and Q4 are on (mode M0), the series circuit of the reactor L1 and the load 2 is short-circuited, and as shown by the arrow in the figure, the positive current IL1 gradually increases. It flows in a diminishing form.

When the auxiliary switch Sr (auxiliary switching element Qr1) is turned on in this state (mode M1), a positive current ILr begins to flow in the auxiliary reactor Lr. Assuming that the current ILr during the commutation period is constant, the reactor current ILr begins to increase linearly with a slope of Vdc / 2Lr as shown in FIG.

When the reactor current ILr becomes larger than the current IL1, the excess current flows from the auxiliary reactor Lr to the switching element Q2 and returns to the auxiliary reactor Lr via the capacitors Cp, Cn and the auxiliary switch Sr (mode M2).

When the switching element Q2 is turned off when the drain current Id of the switching element Q2 becomes positive (mode M3), the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 resonate with each other, so that the reactor current ILr has a peak value. After heading, it decreases toward 0. At this time, the reactor current ILr flows into the auxiliary capacitor Cr1 and also flows into the auxiliary capacitor Cr2. The auxiliary capacitor Cr2 is charged by this current and the auxiliary capacitor Cr1 is discharged, so that the potential of the connection point (node a) of the switching elements Q1 and Q2 rises. Therefore, as shown in FIG. 13, the drain-source voltage Vds of the switching element Q1 gradually decreases.

When the potential Va = Vdc of the node a is reached, the diode D1 of the switching element Q1 is turned on. If the switching element Q1 is turned on while the diode D1 is on, zero voltage switching is performed (mode M4).

By turning on the switching element Q1, a current flows from the positive electrode terminal of the DC power supply P1 to the switching element Q1 and returns to the negative electrode terminal of the DC power supply P1 via the reactor L1, the load 2 and the switching element Q4 (mode M5). The drain current Id flowing between the drain and the source of the switching element Q1 gradually increases, and the reactor current ILr decreases accordingly. After the reactor current ILr = 0, the auxiliary switch Sr (auxiliary switching element Qr1) is turned off (mode M6). Negative current does not flow due to the auxiliary diode Dr2 of the auxiliary switching element Qr2.

FIG. 14 shows the frequency spectra of the operation waveforms of the drain-source voltage Vds and the drain current Id of the switching element Q1. FIG. 14A shows the frequency spectrum (waveform k1) of the Vds waveform of the switching element Q1 of the power conversion device 1 (without auxiliary resonance circuit) according to the first embodiment, and the power conversion device 1 (auxiliary) according to the second embodiment. The frequency spectrum (waveform k2) of the Vds waveform of the switching element Q1 of the switching element Q1 (with resonance circuit) is shown.

FIG. 14B shows the frequency spectrum (waveform k3) of the Id waveform of the switching element Q1 of the power conversion device 1 according to the first embodiment and the Id waveform of the switching element Q1 of the power conversion device 1 according to the second embodiment. The frequency spectrum (waveform k4) is shown.

As shown in FIG. 14 (A), by soft-switching the switching element Q1 using the auxiliary resonance circuit 4, the intensity of Vds in the high frequency region is smaller than that in the case where the switching element Q1 is hard-switched. It has become. Also in FIG. 14B, by soft switching the switching element Q1, the intensity of the Id in the high frequency region is smaller than that in the case where the switching element Q1 is hard-switched. By soft-switching the switching element of the first half-bridge circuit A in this way, electromagnetic interference (EMI) noise can be reduced.

As described above, according to the power conversion device according to the second embodiment, in the full bridge circuit, by providing the auxiliary resonance circuit only in the half bridge circuit switching at the carrier frequency, the switching loss can be reduced and the switching loss can be reduced by a simple configuration. It is possible to reduce the EMI noise.

[Embodiment 3]
In the power conversion device 1 according to the second embodiment described above, in the auxiliary resonance circuit 4, the voltage Vdc of the DC power supply P1 is divided into two to generate the midpoint potential (Vdc / 2) of the capacitors Cp and Cr. A series circuit is used. Therefore, the midpoint potential may fluctuate depending on the magnitude of the load and the like. Further, when an electrolytic capacitor is used for the capacitors Cp and Cr, since the electrolytic capacitor is an electric component whose capacity decreases due to a temperature rise, the deterioration of the capacitors Cp and Cn progresses, so that the life of the entire power conversion device is shortened. There is concern that it will become.

Hereinafter, in the third to fifth embodiments, a new configuration of the auxiliary resonance circuit for realizing the soft switching of the switching elements Q1 and Q2 without using the series circuit of the capacitors Cp and Cr will be described.

FIG. 15 is a main circuit configuration diagram of the power conversion device 1 according to the third embodiment of the present invention. With reference to FIG. 15, the power conversion device 1 according to the third embodiment has a configuration of the auxiliary resonance circuit 4A different from the configuration of the auxiliary resonance circuit 4 of the power conversion device 1 according to the second embodiment shown in FIG. Since the other points are the same as those in the second embodiment, the same or corresponding parts are designated by the same reference numerals and the description is not repeated.

The auxiliary resonance circuit 4A includes auxiliary switching elements Qr1 and Qr2 and an auxiliary reactor Lr. The auxiliary switching elements Qr1 and Qr2 are electrically connected in series between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The auxiliary reactor Lr is electrically connected between the connection point (node c) of the auxiliary switching elements Qr1 and Qr2 and the node a.

Next, the operation of the power conversion device 1 according to the third embodiment, particularly the soft switching by the auxiliary resonance circuit 4A, will be described with reference to FIGS. 16 to 19. 16 and 17 describe soft switching when a positive voltage is applied to the load 2.

FIG. 16 shows a state when the current IL1 is flowing through the diode D2 of the switching element Q2 (mode M0) and is transferred to the switching element Q1. FIG. 17 is an operation waveform diagram when the operation shown in FIG. 16 is executed.

With reference to FIG. 16, when the switching elements Q2 and Q4 are on (mode M0), the series circuit of the reactor L1 and the load 2 is short-circuited, and as shown by the arrow in the figure, the positive current IL1 gradually increases. It flows in a diminishing form.

When the auxiliary switching element Qr1 is turned on in this state (mode M1), a positive reactor current ILr begins to flow in the auxiliary reactor Lr. The reactor current ILr is returned from the positive electrode terminal of the DC power supply P1 to the negative electrode terminal of the DC power supply P1 via the auxiliary switch Sr1, the auxiliary reactor Lr, the reactor L1, the load 2, and the switching element Q4. Assuming that the current IL1 during the commutation period is constant, the reactor current ILr begins to increase linearly with the slope of Vdc / Lr as shown in FIG.

When the reactor current ILr becomes larger than the current IL1, the excess flows from the auxiliary reactor Lr into the switching element Q2. When the switching element Q2 is turned off (mode M2) when the drain current Id of the switching element Q2 becomes positive, the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 resonate with each other. At this time, the reactor current ILr flows into the auxiliary capacitor Cr1 and also flows into the auxiliary capacitor Cr2. The auxiliary capacitor Cr2 is charged by this current and the auxiliary capacitor Cr1 is discharged, so that the potential of the connection point (node a) of the switching elements Q1 and Q2 rises. Therefore, as shown in FIG. 17, the drain-source voltage Vds of the switching element Q1 gradually decreases.

When the potential Va = Vdc of the node a is reached, the diode D1 of the switching element Q1 is turned on. If the switching element Q1 is turned on while the diode D1 is on, zero voltage switching is performed (mode M3). The auxiliary switching element Qr1 is turned off at the same timing as the switching element Q1 is turned on.

When the switching element Q1 is turned on (mode M3), the reactor current ILr flows from the auxiliary reactor Lr in this order from the auxiliary reactor Lr to the diode D1 of the switching element Q1, the positive electrode terminal of the DC power supply P1, and the auxiliary switching element Qr2 due to the current maintenance action of the auxiliary reactor Lr. At the same time, the current flows from the auxiliary reactor Lr to the reactor L1, the load 2, the switch S4, and the auxiliary diode Dr2 of the auxiliary switching element Qr2 in this order. As shown in FIG. 17, the reactor current ILr begins to decrease linearly.

When the reactor current ILr begins to decrease, a current flows from the positive electrode terminal of the DC power supply P1 to the reactor L1 in order to keep the current IL1 constant, and returns to the negative electrode terminal of the DC power supply P1 via the load 2 and the switching element Q4. (Mode M4). The drain current Id flowing between the drain and the source of the switching element Q1 gradually increases. When the reactor current ILr = 0 in this state (mode M5), the drain current Id becomes constant.

Next, with reference to FIGS. 18 and 19, soft switching when a negative voltage is applied to the load 2 will be described.

FIG. 18 shows a state when the current IL1 is flowing through the diode D1 of the switching element Q1 (mode M0) and is transferred to the switching element Q2. FIG. 19 is an operation waveform diagram when the operation shown in FIG. 18 is executed.

With reference to FIG. 18, when the switching elements Q1 and Q3 are on (mode M0), the series circuit of the reactor L1 and the load 2 is short-circuited, and as shown by the arrow in the figure, the absolute negative current IL1. The value gradually decreases.

When the auxiliary switching element Qr2 is turned on in this state (mode M1), a negative reactor current ILr begins to flow in the auxiliary reactor Lr. The reactor current ILr is returned from the positive electrode terminal of the DC power supply P1 to the negative electrode terminal of the DC power supply P1 via the switching element Q3, the load 2, the reactor L1, the auxiliary reactor Lr, and the auxiliary switching element Qr2. Assuming that the current IL1 during the commutation period is constant, the reactor current ILr begins to decrease linearly as shown in FIG.

When the switching element Q1 is turned off in this state (mode M2), the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 resonate with each other. At this time, the reactor current ILr flows into the auxiliary capacitor Cr1 and also flows into the auxiliary capacitor Cr2. The auxiliary capacitor Cr2 is discharged by this current and the auxiliary capacitor Cr1 is charged, so that the potential of the connection point (node a) of the switching elements Q1 and Q2 is lowered. Therefore, as shown in FIG. 19, the drain-source voltage Vds of the switching element Q2 gradually decreases.

When the potential Va = 0 of the node a is reached, the diode D2 of the switching element Q2 is turned on. If the switching element Q2 of the switch S2 is turned on while the diode D2 is on, zero voltage switching is performed (mode M3). The auxiliary switching element Qr2 is turned off at the same timing as the switching element Q2 is turned on.

When the auxiliary switching element Qr2 is turned off (mode M3), the reactor current ILr is changed from the auxiliary reactor Lr to the positive electrode terminal and the negative electrode terminal of the DC power supply P1 via the diode Dr1 of the auxiliary switching element Qr1 due to the current maintenance action of the auxiliary reactor Lr. The current flows in the order of the diode D2 of the switching element Q2, and also flows in the order of the auxiliary reactor Lr, the switching element Q3, the load 2, and the reactor L1. As shown in FIG. 19, the reactor current ILr begins to increase linearly.

When the reactor current ILr begins to increase, a current flows from the positive electrode terminal of the DC power supply P1 to the reactor L1 via the switching element Q3 and the load 2 in order to keep the current IL1 constant, and the DC power supply P1 passes through the switching element Q2. (Mode M4). The drain current Id flowing between the drain and the source of the switching element Q3 gradually increases. When the reactor current ILr becomes 0 in this state (mode M5), the drain current Id becomes constant.

As described above, according to the power conversion device 1 according to the third embodiment of the present invention, since the auxiliary resonance circuit does not include the series circuit of the capacitors Cp and Cr, the fluctuation of the midpoint potential of the series circuit is suppressed. can do. Further, it is possible to prevent the life of the power conversion device from being shortened due to the deterioration of the capacitors Cp and Cr.

[Embodiment 4]
In the auxiliary resonance circuit 4A according to the third embodiment described above, the auxiliary resonance circuit 4A is assisted in a state where a current is flowing through the auxiliary switching element Qr1 due to the resonance operation between the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 (mode M2 in FIG. 16). The switching element Qr1 is turned off (mode M3). Therefore, while the semiconductor switching element Q1 can be soft-switched, there is a concern that the auxiliary switching element Qr1 becomes hard switching.

In order to soft-switch the auxiliary switching elements Qr1 and Qr2, as shown in the second embodiment, it is necessary to create a voltage obtained by dividing the voltage Vdc of the DC power supply P1 into two. For the above reason, the capacitor Cp , It is not a good idea to use a series circuit of Cn.

In the following embodiments 4 and 5, the configuration of the auxiliary resonance circuit capable of generating the voltage obtained by dividing the voltage Vdc of the DC power supply P1 into two without using the series circuit of the capacitors Cp and Cn will be described.

FIG. 20 is a main circuit configuration diagram of the power conversion device 1 according to the fourth embodiment of the present invention. With reference to FIG. 20, in the power conversion device 1 according to the fourth embodiment, the configuration of the auxiliary resonance circuit 4B is different from the configuration of the auxiliary resonance circuit 4 of the power conversion device 1 according to the second embodiment shown in FIG. Since the other points are the same as those in the second embodiment, the same or corresponding parts are designated by the same reference numerals and the description is not repeated.

The auxiliary resonance circuit 4B includes auxiliary switching elements Qr1 and Qr2, an auxiliary reactor Lr, and a transformer Tr1. The transformer Tr1 includes a primary winding n1 and a secondary winding n2. The transformer Tr1 has an exciting inductance Lm and a leakage inductance in addition to the ideal transformer. By electromagnetically coupling the primary winding n1 and the secondary winding n2, an electromotive force having the same polarity is induced in the primary winding n1 and the secondary winding n2. The winding ratio between the primary winding n1 and the secondary winding n2 is 2: 1. The transformer Tr1 corresponds to one embodiment of the "first transformer".

The primary winding n1 of the transformer Tr1 is electrically connected between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The secondary winding n2 of the transformer Tr1 is electrically connected between the node a of the first half bridge circuit A and the DC negative bus NL1.

The auxiliary switching element Qr1 is electrically connected between the primary winding n1 of the transformer Tr1 and the DC negative bus NL1. The auxiliary switching element Qr2 is electrically connected between the secondary winding n2 of the transformer Tr1 and the DC negative bus NL1. Auxiliary diodes Dr1 and Dr2 are connected in anti-parallel to the auxiliary switching elements Qr1 and Qr2, respectively.

The auxiliary reactor Lr is connected between the secondary winding n2 and the node a. The auxiliary reactor Lr may utilize the leakage inductance of the transformer Tr1 or may be externally attached.

Next, the operation of the power conversion device 1 according to the fourth embodiment, particularly the soft switching by the auxiliary resonance circuit 4B, will be described with reference to FIGS. 21 to 26. 21 to 23 show soft switching when a positive voltage is applied to the load 2.

21 and 22 show a state in which the current IL1 is flowing through the diode D2 of the switching element Q2 (mode M0) and is transferred to the switching element Q1. FIG. 23 is an operation waveform diagram when the operations shown in FIGS. 21 and 22 are executed.

With reference to FIG. 21, when the switching elements Q2 and Q4 are on (mode M0), the series circuit of the reactor L1 and the load 2 is short-circuited, and as shown by the arrow in the figure, the positive current IL1 gradually increases. It flows in a diminishing form.

When the auxiliary switching element Qr1 is turned on in this state (mode M1), the voltage Vdc of the DC power supply P1 is applied to the primary winding n1 of the transformer Tr1, and the winding ratio between the primary winding n1 and the secondary winding n2 is increased. Since it is 2: 1, a voltage ½ of the voltage Vdc is applied to the connection point of the secondary winding n2 and the auxiliary reactor Lr. As a result, a positive reactor current ILr begins to flow in the auxiliary reactor Lr. The reactor current ILr is fed back from the secondary winding n2 of the transformer Tr1 to the secondary winding n2 via the auxiliary reactor Lr, the switching element Q4, and the auxiliary diode Dr2 of the auxiliary switching element Qr2. Assuming that the current IL1 during the commutation period is constant, as shown in FIG. 23, the reactor current ILr begins to increase linearly with a slope of Vdc / 2Lr. Further, the drain current Id of the switching element Q2 begins to increase linearly toward 0.

When the reactor current ILr becomes larger than the current IL1, the excess current flows from the auxiliary reactor Lr into the switching element Q2 (mode M2).

When the switching element Q2 is turned off (mode M3) when the drain current Id of the switching element Q2 becomes positive, the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 resonate with each other. At this time, the reactor current ILr flows into the auxiliary capacitor Cr2 and also flows into the auxiliary capacitor Cr1. The auxiliary capacitor Cr2 is charged by this current and the auxiliary capacitor Cr1 is discharged, so that the potential of the connection point (node a) of the switching elements Q1 and Q2 rises. Therefore, as shown in FIG. 23, the drain-source voltage Vds of the switching element Q1 gradually decreases.

When the potential Va = Vdc of the node a is reached, the diode D1 of the switching element Q1 is turned on. If the switching element Q1 is turned on while the diode D1 is on, zero voltage switching is performed (mode M4).

When the switching element Q1 is turned on (mode M5), a current flows from the positive electrode terminal of the DC power supply P1 to the switching element Q1 and returns to the negative electrode terminal of the DC power supply P1 via the reactor L1, the load 2, and the switching element Q4. The drain current Id flowing between the drain and the source of the switching element Q1 gradually increases. When the reactor current ILr = 0, the auxiliary switching element Qr1 is turned off (mode M6) to perform zero current switching of the auxiliary switching element Qr1.

Next, with reference to FIGS. 24 and 26, soft switching when a negative voltage is applied to the load 2 will be described.

24 and 25 show a state in which the current IL1 is flowing through the diode D1 of the switching element Q1 (mode M0) and is transferred to the switching element Q2. FIG. 26 is an operation waveform diagram when the operations shown in FIGS. 24 and 25 are executed.

With reference to FIG. 24, when the switching elements Q1 and Q3 are on (mode M0), the series circuit of the reactor L1 and the load 2 is short-circuited, and as shown by the arrow in the figure, the absolute negative current IL1. The value gradually decreases.

When the auxiliary switching element Qr2 is turned on in this state (mode M1), a negative reactor current ILr begins to flow in the auxiliary reactor Lr. The reactor current ILr is returned from the positive electrode terminal of the DC power supply P1 to the negative electrode terminal of the DC power supply P1 via the switching element Q3, the load 2, the reactor L1, the auxiliary reactor Lr, and the auxiliary switching element Qr2. At this time, a current flows through the path of the auxiliary diode Dr1, the primary winding n1, the DC power supply P1, and the auxiliary diode Dr1. A voltage Vdc is applied to the primary winding n1, and the winding ratio between the primary winding n and the secondary winding n2 is 2: 1. Therefore, a voltage is applied to the connection point between the secondary winding n2 and the auxiliary reactor Lr. A voltage of 1/2 of Vdc is applied. Assuming that the current IL1 during the commutation period is constant, the reactor current ILr begins to decrease linearly with a slope of −Vdc / 2Lr, as shown in FIG. Further, the drain current Id of the switching element Q1 begins to increase linearly toward 0.

When the reactor current ILr becomes smaller than the current IL1, the excess current flows into the switching element Q3 (mode M2).

When the switching element Q1 is turned off (mode M3) when the drain current Id of the switching element Q1 becomes positive, the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2 resonate with each other. At this time, the reactor current ILr flows into the auxiliary capacitor Cr2 and also flows into the auxiliary capacitor Cr1. The auxiliary capacitor Cr1 is charged by this current and the auxiliary capacitor Cr2 is discharged, so that the potential of the connection point (node a) of the switching elements Q1 and Q2 is lowered. Therefore, as shown in FIG. 26, the drain-source voltage Vds of the switching element Q2 gradually decreases.

When the potential Va = 0 of the node a is reached, the diode D2 of the switching element Q2 is turned on. If the switching element Q2 is turned on while the diode D2 is on, zero voltage switching is performed (mode M4).

When the switching element Q2 is turned on, a current flows from the positive electrode terminal of the DC power supply P1 to the switching element Q3 and returns to the negative electrode terminal of the DC power supply P1 via the load 2, the reactor L1 and the switching element Q2 (mode M5). The drain current Id flowing between the drain and the source of the switching element Q2 gradually increases. When the reactor current ILr = 0, the auxiliary switching element Qr2 is turned off (mode M6) to perform zero current switching of the auxiliary switching element Qr1.

As described above, according to the power conversion device 1 according to the fourth embodiment of the present invention, it is possible to generate a voltage obtained by dividing the voltage Vdc of the DC power supply P1 into two without using the series circuit of the capacitors Cp and Cn. Therefore, the auxiliary switching elements Qr1 and Qr2 can be soft-switched.

Further, in the power conversion device 1 according to the fourth embodiment, since both the auxiliary switching elements Qr1 and Qr2 are connected to the negative terminal (DC negative bus NL1) of the DC power supply P1, the gate signal Gr1 of the auxiliary switching element Qr1 is connected. And the gate signal Gr2 of the auxiliary switching element Qr2 can be used as a signal having the same potential. In the power conversion device 1 according to the third embodiment shown in FIG. 15, since the auxiliary switching elements Qr1 and Qr2 are connected in series between the DC positive bus PL1 and the DC negative bus NL1, the auxiliary switching element on the high potential side. It is necessary to make the potential of the gate signal Gr1 of Qr1 higher than the potential of the gate signal Gr2 of the auxiliary switching element Qr1 on the low potential side. Therefore, the gate drive circuit of the auxiliary switching element Qr1 has a built-in level shifter for raising the potential of the gate signal Gr1, and is electrically separated from the gate drive circuit of the auxiliary switching element Qr2.

On the other hand, in the fourth embodiment, since the auxiliary switching elements Qr1 and Qr2 are at the same potential, it is not necessary to electrically separate the gate drive circuits from each other. Therefore, the gate drive circuit can be simplified, and as a result, the power conversion device 1 can be miniaturized and reduced in cost.

[Modification example]
FIG. 27 is a main circuit configuration diagram of the power conversion device 1 according to the modification of the fourth embodiment. With reference to FIG. 27, in the power conversion device 1 according to the modification of the fourth embodiment, the configuration of the auxiliary resonance circuit 4C is the same as the configuration of the auxiliary resonance circuit 4B of the power conversion device 1 according to the fourth embodiment shown in FIG. different. Since the other points are the same as those in the second embodiment, the same or corresponding parts are designated by the same reference numerals and the description is not repeated.

In the power conversion device 1 according to this modification, the transformer Tr1 of the power conversion device 1 shown in FIG. 20 is replaced with the transformer Tr2, and the diode Da2 is added.

The transformer Tr2 includes a primary winding n1, a secondary winding n2 and a reset winding n3. One end of the reset winding n3 is connected to the DC positive bus PL, and the other end is connected to the cathode of the diode Da2. The anode of the diode Da2 is connected to the DC negative bus NL1. That is, the series circuit of the diode Da2 and the reset winding n3 is electrically connected between the DC positive bus PL1 and the DC negative bus NL1 so that the direction from the DC negative bus NL1 to the DC positive bus PL1 is the forward direction of the diode Da2. Is connected.

The reset winding n3 and the diode Da2 constitute a reset circuit for discharging the electromagnetic energy stored in the exciting inductance Lm of the transformer Tr2.

Specifically, when the auxiliary switching elements Qr1 and Qr2 are turned off, a counter electromotive force is generated by the exciting inductance Lm of the transformer Tr2. When the voltage (surge voltage) generated in the primary winding n1 and the secondary winding n2 is applied to the auxiliary switching elements Qr1 and Qr2 by this counter electromotive force, the auxiliary switching elements Qr1 and Qr2 may be damaged.

The series circuit of the reset winding n3 and the diode Da2 can regenerate this counter electromotive force to the DC power supply P1. As a result, it is possible to prevent an overvoltage from being applied to the auxiliary switching elements Qr1 and Qr2, so that damage to the auxiliary switching elements Qr1 and Qr2 can be prevented.

[Embodiment 5]
FIG. 28 is a main circuit configuration diagram of the power conversion device 1 according to the fifth embodiment of the present invention. With reference to FIG. 28, the power conversion device 1 according to the fifth embodiment has a different configuration of the auxiliary resonance circuit 4D from the configuration of the auxiliary resonance circuit 4A of the power conversion device 1 according to the third embodiment shown in FIG. Since the other points are the same as those in the second embodiment, the same or corresponding parts are designated by the same reference numerals and the description is not repeated.

The auxiliary resonant circuit 4D includes auxiliary switching elements Qr1, Qr2, auxiliary reactor Lr, transformer Tr3, and diodes Db1, Db2, Dc1, Dc2, Dr3, Dr4.

The auxiliary switching elements Qr1 and Qr2 are electrically connected in series between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The auxiliary reactor Lr is electrically connected between the connection point (node c) of the auxiliary switching elements Qr1 and Qr2 and the node a.

The diode Dr3 is electrically connected in series with the auxiliary switching element Qr1 between the DC positive bus PL1 and the node c. The diode Dr4 is electrically connected in series with the auxiliary switching element Qr2 between the DC negative bus NL1 and the node c. In the configuration example of FIG. 27, the anode of the diode Dr3 is connected to the auxiliary switching element Qr1, and the cathode is connected to the node c. The diode Dr3 may be connected between the DC negative bus PL1 and the auxiliary switching element Qr1. In the configuration example of FIG. 28, the anode of the diode Dr4 is connected to the node c, and the cathode is connected to the auxiliary switching element Qr2. The diode Dr4 may be connected between the auxiliary switching element Qr2 and the DC negative bus NL1. The diode Dr3 corresponds to one embodiment of the "fifth diode" and the diode Dr4 corresponds to one embodiment of the "sixth diode".

The diodes Db1 and Db2 are electrically connected in series between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The anode of the diode Db2 is electrically connected to the DC negative bus NL1, and the cathode of the diode Db1 is electrically connected to the DC positive bus PL1. The diode Db1 corresponds to one embodiment of the "first diode" and the diode Db2 corresponds to one embodiment of the "second diode".

The diodes Dc1 and Dc2 are electrically connected in series between the DC terminal T1 (DC positive bus PL1) and the DC terminal T2 (DC negative bus NL1). The anode of the diode Dc2 is electrically connected to the DC negative bus NL1, and the cathode of the diode Dc1 is electrically connected to the DC positive bus PL1. The diode Dc1 corresponds to one embodiment of the "third diode" and the diode Dc2 corresponds to one embodiment of the "fourth diode".

The transformer Tr3 includes a primary winding n1 and a secondary winding n2. The transformer Tr3 has an exciting inductance Lm and a leakage inductance in addition to the ideal transformer. By electromagnetically coupling the primary winding n1 and the secondary winding n2, an electromotive force having the same polarity is induced in the primary winding n1 and the secondary winding n2. The winding ratio between the primary winding n1 and the secondary winding n2 is 1: 2. The transformer Tr3 corresponds to one embodiment of the "second transformer".

The secondary winding n2 of the transformer Tr3 is connected between the connection point (node d) of the diodes Db1 and Db2 and the connection point (node e) of the diodes Dc1 and Dc2. The primary winding n1 of the transformer Tr3 is electrically connected in series with the auxiliary reactor Lr between the connection point (node c) of the auxiliary switching elements Qr1 and Qr2 and the node a. The auxiliary reactor Lr may utilize the leakage inductance of the transformer Tr3 or may be externally attached.

The diodes Db1, Db2, Dc1, Dc2 and the secondary winding n2 of the transformer Tr3 form a diode bridge circuit. The diode bridge circuit uses the connection point of the diodes Db1 and Dc1 (that is, the DC positive bus PL1) and the connection point of the diodes Db2 and Dc2 (that is, the DC negative bus NL1) as output terminals, and the connection point (node) of the diodes Db1 and Db2. d) and the connection point (node e) of the diodes Dc1 and Dc2 are used as input terminals.

When the auxiliary switching element Qr1 or Qr2 is turned on and a current is passed through the primary winding n1 of the transformer Tr3, a current is generated in the secondary winding n2 by electromagnetic induction. A voltage Vdc or −Vdc is applied to the secondary winding n2 by the current generated in the secondary winding n2 flowing through two diodes connected to both ends of the secondary winding n2. Since the winding ratio between the primary winding n1 and the secondary winding n2 is 2: 1, a voltage ½ of the voltage Vdc or −Vdc is applied to the primary winding n1.

The operation of the power conversion device 1 according to the fifth embodiment is the same as the operation of the power conversion device 1 according to the fourth embodiment shown in FIGS. 23 and 26. That is, the soft switching by the auxiliary resonance circuit 4D when a positive voltage is applied to the load 2 is the same as the soft switching by the auxiliary resonance circuit 4B shown in FIG. 23.

Specifically, as shown in the mode M1 of FIG. 23, in order to turn on the switching element Q1 by zero voltage switching, the auxiliary switching element Qr1 is turned on. Therefore, as shown in FIG. 29, the potential Vc = Vdc of the node c. On the other hand, since the switching element Q2 is on, the potential Va = 0 of the node a.

In this case, as shown by an arrow in the figure, a current flows from the node c toward the node a. In the transformer Tr3, the current flowing through the primary winding n1 induces the current flowing through the diode Dc2, the secondary winding n2, and the diode Db1. As a result, the voltage Vdc is applied to the secondary winding n2 so that the terminal on the node d side has a high potential and the terminal on the node e side has a low potential. Therefore, the primary winding n1 has a node c side. A voltage of 1/2 of the voltage Vdc is applied with the terminal of No. as a high potential. As a result, a voltage ½ of the voltage Vdc can be applied to the connection point of the auxiliary reactor Lr and the primary winding n1. Zero voltage switching can be realized by turning on the switching element Q1 when the potential Va = Vdc of the node a due to the resonance operation of the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2.

After the switching element Q1 is turned on (mode M4 in FIG. 23), when the reactor current ILr = 0, the auxiliary switching element Qr1 is turned off (mode M6) to perform zero current switching of the auxiliary switching element Qr1. Is done. The diode Dr3 is provided to prevent the reactor current ILr from starting to flow in the reverse direction (negative direction).

Similarly, the soft switching by the auxiliary resonance circuit 4D when a negative voltage is applied to the load 2 is the same as the soft switching by the auxiliary resonance circuit 4B shown in FIG. 26. As shown in the mode M1 of FIG. 26, in order to turn on the switching element Q2 by zero voltage switching, the auxiliary switching element Qr2 is turned on. Therefore, as shown in FIG. 30, the potential Vc of the node c becomes 0. On the other hand, since the switching element Q1 is on, the potential Va = Vdc of the node a.

In this case, as shown by the arrow in the figure, a current flows from the node a to the node c. In the transformer Tr3, the current flowing through the primary winding n1 induces the current flowing through the diode Db2, the secondary winding n2, and the diode Dc1. As a result, the voltage Vdc is applied to the secondary winding n2 so that the terminal on the node e side has a high potential and the terminal on the node d side has a low potential. Therefore, the primary winding n1 has the node a side. A voltage of 1/2 of the voltage Vdc is applied with the terminal of No. as a high potential. As a result, a voltage ½ of the voltage Vdc can be applied to the connection point of the auxiliary reactor Lr and the primary winding n1. Zero voltage switching can be realized by turning on the switching element Q2 when the potential Va = 0 of the node a due to the resonance operation of the auxiliary reactor Lr and the auxiliary capacitors Cr1 and Cr2.

After the switching element Q2 is turned on (mode M4 in FIG. 25), when the reactor current ILr = 0, the auxiliary switching element Qr2 is turned off to perform zero current switching of the auxiliary switching element Qr2. The diode Dr4 is provided to prevent the reactor current ILr from starting to flow in the reverse direction (positive direction).

As described above, according to the power conversion device 1 according to the fifth embodiment of the present invention, it is possible to generate a voltage obtained by dividing the voltage Vdc of the DC power supply P1 into two without using the series circuit of the capacitors Cp and Cn. Therefore, the auxiliary switching elements Qr1 and Qr2 can be soft-switched.

Further, in the power conversion device 1 according to the fifth embodiment, a diode bridge circuit can be used for the diodes Db1, Db2, Dc1 and Dc2. Since the diode bridge circuit is usually a small element, it is possible to create a voltage that is ½ of the voltage Vdc without increasing the size of the power conversion device 1.

[Application example of power converter]
In the above-described embodiments 2 to 5, the configuration for realizing soft switching in the power conversion device 1 including the asymmetrical drive type full bridge circuit described in the first embodiment has been described, but each embodiment has been described. The auxiliary resonance circuit shown in (1) can be applied to all power conversion devices having a half-bridge circuit as shown below.

31 to 34 show power conversion devices 1A to 1D each including a half-bridge circuit. With reference to FIG. 31, the power converter 1A includes a first half-bridge circuit A and an auxiliary resonant circuit 4A of FIG. With reference to FIG. 32, the power converter 1B has a first half-bridge circuit A and an auxiliary resonant circuit 4B of FIG. With reference to FIG. 33, the power converter 1C has a first half-bridge circuit A and an auxiliary resonant circuit 4C of FIG. 27. With reference to FIG. 34, the power converter 1D includes a first half-bridge circuit A and an auxiliary resonant circuit 4D of FIG. 28. The operation of the power conversion devices 1A to 1D is the same as the operation of the power conversion devices according to the second to fifth embodiments, except that the second half bridge circuit B is turned on and off.

The embodiments disclosed this time should be considered to be exemplary in all respects and not restrictive. The scope of the present invention is shown by the scope of claims rather than the above description, and is intended to include all modifications within the meaning and scope equivalent to the scope of claims.

1,1A to 1D power converter, 2,3 load, 4,4A, 4B, 4C, 4D auxiliary resonance circuit, 5 control circuit, A 1st half bridge circuit, B 2nd half bridge circuit, C0, C5, Cn , Cp capacitor, Cr1, Cr2 auxiliary capacitor, D1 to D4, Da1, Da2, Db1, Db2, Dc1, Dc2 diode, Dr1, Dr2 auxiliary diode, L1 reactor, Lr auxiliary reactor, PL1 DC positive bus, NL1 DC negative bus, P1 DC power supply, P2 AC system, Q1 to Q4 semiconductor switching element, Qr1, Qr2 auxiliary switching element, S1 to S4 switch, Sr, Sr1, Sr2 auxiliary switch, T1 DC terminal (high potential side DC terminal), T2 DC terminal ( Low potential side DC terminal), T3, T4 AC terminal, Tr1, Tr2, Tr3 transformer, n1 primary winding, n2 secondary winding, n3 reset winding.

Claims (10)

A power conversion device that converts power between DC power and AC power.
High potential side DC terminal and low potential side DC terminal,
A first switching element electrically connected between the high potential side DC terminal and the first node,
A second switching element electrically connected between the first node and the low potential side DC terminal, and
A third switching element electrically connected between the high potential side DC terminal and the second node,
A fourth switching element electrically connected between the second node and the low potential side DC terminal, and
The first AC terminal electrically connected to the first node,
A second AC terminal electrically connected to the second node,
A control circuit for controlling the on / off of the first to fourth switching elements is provided.
The control circuit is
The third and fourth switching elements are turned on and off in synchronization with the frequency of the AC power, and are also turned on and off.
The first and second switching elements are configured to be turned on and off at a frequency higher than the frequency of the AC power.
A first capacitor connected in parallel to the first switching element,
A second capacitor connected in parallel to the second switching element,
It further comprises an auxiliary resonant circuit including a reactor, configured to soft-switch the first and second switching elements by resonant operation of the first and second capacitors with the reactor.
The auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal.
The reactor is a power conversion device that is electrically connected between the connection node of the first and second auxiliary switching elements and the first node . A power conversion device that converts power between DC power and AC power.
High potential side DC terminal and low potential side DC terminal,
A first switching element electrically connected between the high potential side DC terminal and the first node,
A second switching element electrically connected between the first node and the low potential side DC terminal, and
A third switching element electrically connected between the high potential side DC terminal and the second node,
A fourth switching element electrically connected between the second node and the low potential side DC terminal, and
The first AC terminal electrically connected to the first node,
A second AC terminal electrically connected to the second node,
A control circuit for controlling the on / off of the first to fourth switching elements is provided.
The control circuit is
The third and fourth switching elements are turned on and off in synchronization with the frequency of the AC power, and are also turned on and off.
The first and second switching elements are configured to be turned on and off at a frequency higher than the frequency of the AC power.
A first capacitor connected in parallel to the first switching element,
A second capacitor connected in parallel to the second switching element,
It further comprises an auxiliary resonant circuit including a reactor, configured to soft-switch the first and second switching elements by resonant operation of the first and second capacitors with the reactor.
The auxiliary resonance circuit is
A primary winding electrically connected between the high-potential side DC terminal and the low-potential side DC terminal, and electrically in series with the reactor between the first node and the low-potential side DC terminal. A first transformer with a connected secondary winding, and
A first auxiliary switching element electrically connected between the primary winding and the low potential side DC terminal, and
A power conversion device including a second auxiliary switching element electrically connected between the secondary winding and the low potential side DC terminal . The power conversion device according to claim 2, wherein the winding ratio of the primary winding and the secondary winding of the first transformer is 2: 1. The first transformer has a reset winding electrically connected between the high potential side DC terminal and the low potential side DC terminal.
The auxiliary resonance circuit is electrically connected to the reset winding between the high potential side DC terminal and the low potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. The power conversion device according to claim 2 or 3 , further comprising a diode connected in series with. A power conversion device that converts power between DC power and AC power.
High potential side DC terminal and low potential side DC terminal,
A first switching element electrically connected between the high potential side DC terminal and the first node,
A second switching element electrically connected between the first node and the low potential side DC terminal, and
A third switching element electrically connected between the high potential side DC terminal and the second node,
A fourth switching element electrically connected between the second node and the low potential side DC terminal, and
The first AC terminal electrically connected to the first node,
A second AC terminal electrically connected to the second node,
A control circuit for controlling the on / off of the first to fourth switching elements is provided.
The control circuit is
The third and fourth switching elements are turned on and off in synchronization with the frequency of the AC power, and are also turned on and off.
The first and second switching elements are configured to be turned on and off at a frequency higher than the frequency of the AC power.
A first capacitor connected in parallel to the first switching element,
A second capacitor connected in parallel to the second switching element,
It further comprises an auxiliary resonant circuit including a reactor, configured to soft-switch the first and second switching elements by resonant operation of the first and second capacitors with the reactor.
The auxiliary resonant circuit further includes first and second auxiliary switching elements electrically connected in series between the high potential side DC terminal and the low potential side DC terminal.
The reactor is electrically connected between the connection node of the first and second auxiliary switching elements and the first node.
The auxiliary resonance circuit is
The first and second electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. Diode and
The third and fourth electrically connected in series between the low potential side DC terminal and the high potential side DC terminal with the direction from the low potential side DC terminal to the high potential side DC terminal as the forward direction. Diode and
A primary winding electrically connected in series with the reactor between the connection node of the first and second auxiliary switching elements and the first node, and a connection node of the first and second diodes. A power conversion device further comprising a second transformer having a secondary winding electrically connected between the third and fourth diode connection nodes . The power conversion device according to claim 5 , wherein the winding ratio of the primary winding and the secondary winding of the second transformer is 1: 2. The auxiliary resonance circuit is
A fifth diode electrically connected in series with the first auxiliary switching element between the high potential side DC terminal and the connection node of the first and second auxiliary switching elements.
A sixth diode electrically connected in series with the second auxiliary switching element is further included between the low potential side DC terminal and the connection node of the first and second auxiliary switching elements. The power conversion device according to claim 5 or 6 . A power conversion device that converts power between DC power and AC power.
High potential side DC terminal and low potential side DC terminal,
A first switching element electrically connected between the high potential side DC terminal and the first node,
A second switching element electrically connected between the first node and the low potential side DC terminal, and
A third switching element electrically connected between the high potential side DC terminal and the second node,
A fourth switching element electrically connected between the second node and the low potential side DC terminal, and
The first AC terminal electrically connected to the first node,
A second AC terminal electrically connected to the second node,
A control circuit for controlling the on / off of the first to fourth switching elements is provided.
The control circuit is
The third and fourth switching elements are turned on and off in synchronization with the frequency of the AC power, and are also turned on and off.
The first and second switching elements are configured to be turned on and off at a frequency higher than the frequency of the AC power.
Each of the first and second switching elements has a first power semiconductor switching element and a first freewheeling diode connected in antiparallel to the first power semiconductor switching element.
Each of the third and fourth switching elements has a second power semiconductor switching element and a second freewheeling diode connected in antiparallel to the second power semiconductor switching element.
The first freewheeling diode is a power conversion device having a shorter reverse recovery time than the second freewheeling diode . The power conversion device according to claim 8 , wherein the first freewheeling diode is formed of a wide bandgap semiconductor. The power conversion device according to any one of claims 1 to 9 , wherein the control circuit generates a control signal for controlling on / off of the first and second switching elements by pulse width modulation control.

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