JP6915133B2 - Motor control device, sheet transfer device, document reader and image forming device - Google Patents

Description

The present invention relates to the control of a motor in a motor control device, a sheet transfer device, a document reading device, and an image forming device.

Conventionally, as a method of controlling a motor, a control method called vector control is known in which the motor is controlled by controlling a current value in a rotating coordinate system based on the rotation phase of the rotor of the motor. Specifically, a control method for controlling a motor by performing phase feedback control for controlling a current value in a rotating coordinate system so that a deviation between a command phase of a rotor and a rotation phase becomes small is known. A control method for controlling the motor by performing speed feedback control for controlling the current value in the rotating coordinate system so that the deviation between the command speed of the rotor and the rotation speed becomes small is also known.

In vector control, the drive current flowing through the windings of the motor affects the q-axis component (torque current component), which is the current component that generates torque for the rotor to rotate, and the strength of the magnetic flux that penetrates the windings of the motor. It is represented by a d-axis component (exciting current component), which is a current component to be generated. By controlling the value of the torque current component according to the change in the load torque applied to the rotor, the torque required for rotation is efficiently generated. As a result, an increase in motor noise and an increase in power consumption due to excess torque are suppressed. In addition, the load torque applied to the rotor exceeds the output torque corresponding to the drive current supplied to the windings of the motor, causing the rotor to become out of sync with the input signal and the motor to be out of control (step-out). It is possible to prevent it from becoming a state).

Vector control requires a configuration that determines the rotation phase of the rotor. Patent Document 1 describes a configuration in which the rotation phase of the rotor is determined based on the induced voltage generated in the windings of each phase of the motor due to the rotation of the rotor.

The magnitude of the induced voltage generated in the winding decreases as the rotation speed of the rotor decreases. If the magnitude of the induced voltage generated in the winding is not large enough to determine the rotation phase of the rotor, the rotation phase may not be determined accurately. That is, the smaller the rotation speed of the rotor, the lower the accuracy of determining the rotation phase of the rotor.

Therefore, in Patent Document 2, when the command speed of the rotor is smaller than a predetermined rotation speed, a constant current control is used to control the motor by supplying a predetermined current to the winding of the motor. It is stated. In the constant current control, neither the phase feedback control nor the velocity feedback control is performed. Further, when the command speed of the rotor is equal to or higher than a predetermined rotation speed, a configuration in which vector control is used is described.

When the control of the motor is switched from the constant current control to the vector control, the rotation speed of the motor may decrease momentarily. This was given to the rotor by the first drive current after the motor control was switched, rather than the torque given to the rotor by the last drive current before the motor control was switched. This is because the torque may be smaller. That is, there is a difference between the torque given to the rotor immediately before the control of the motor is switched and the torque given to the rotor immediately after the control of the motor is switched.

In Patent Document 3, the load torque applied to the rotor during microstep driving is estimated (calculated) based on the signal output from the position detector. Then, a configuration is described in which the current to be supplied immediately after the motor control method is switched from the microstep drive control to the speed servo control is determined (calculated) based on the estimated load torque. Specifically, in Patent Document 3, the load torque is calculated by the load estimator in order to determine the current to be supplied immediately after the motor control method is switched. Then, the current value corresponding to the load torque is calculated by dividing the torque constant from the load torque, and the calculated current value is multiplied by the control gain to determine the current to be supplied.

Special Table 2012-509056 Japanese Unexamined Patent Publication No. 2005-39955 Japanese Unexamined Patent Publication No. 2010-289949

In Patent Document 3, in order to determine the current to be supplied immediately after the motor control method is switched, a process of dividing a preset torque constant from the load torque is performed. The torque constant has a different value depending on the temperature of the motor. Therefore, if a torque constant as a fixed value is used regardless of the temperature of the motor, an error will be included in the calculated current value. That is, the determined current also includes an error, and there is a possibility that the current to be supplied immediately after the motor control method is switched is not properly set. As a result, the control of the motor may become unstable.

In view of the above problems, it is an object of the present invention to suppress instability of control when the control mode for controlling the drive current is switched.

In order to solve the above problems, the motor control device according to the present invention is
In a motor control device that controls the motor based on a command phase representing the target phase of the rotor of the motor.
A supply means for supplying a drive current to the winding of the motor,
A detection means for detecting the drive current flowing through the winding and
A phase determining means for determining the rotational phase of the rotor based on the drive current detected by the detecting means, and
The winding is set so that the magnitude of the drive current detected by the detecting means becomes a target value set so that the deviation between the command phase and the rotation phase determined by the phase determining means becomes small. A control means including a first control mode for controlling the supplied drive current and a second control mode for controlling the drive current supplied to the winding based on a current having a predetermined magnitude.
Have,
In the first control mode, the control means performs integral control based on the deviation so that the deviation becomes small.
The initial value of the integral control in the first control mode when the control mode is switched from the second control mode to the first control mode was detected by the detection means during the execution of the second control mode. Set based on the drive current value,
The target value is set based on the integral control in which the initial value is set.
The control means changes the control mode from the second control mode in a state where the initial value is set based on the value of the drive current detected by the detection means during the execution of the second control mode. It is characterized by switching to the first control mode.

According to the present invention, it is possible to prevent the control from becoming unstable when the control mode for controlling the drive current is switched.

It is sectional drawing explaining the image forming apparatus which concerns on 1st Embodiment. It is a block diagram which shows the control structure of the image forming apparatus. It is a figure which shows the relationship between the two-phase motor which consists of A phase and B phase, and the rotating coordinate system represented by d-axis and q-axis. It is a block diagram which shows the structure of the motor control device which concerns on 1st Embodiment. It is a block diagram which shows the structure of the phase controller which concerns on 1st Embodiment. It is a block diagram which shows the structure of the current controller which concerns on 1st Embodiment. It is a block diagram which shows the example of the control structure of the conventional constant current control. It is a figure explaining the constant current control which concerns on 1st Embodiment. It is a block diagram which shows the example of the structure of the constant current controller which concerns on 1st Embodiment. It is a figure which shows the relationship between the rotation speed ω_ref', the threshold value ωth, and a switching signal. It is a flowchart which shows the control method of the motor which concerns on 1st Embodiment. It is a figure which shows the experimental result which shows the change of the drive current, the waveform of the drive voltage, and the rotation speed with switching of the control method of a motor. It is a block diagram which shows the example of the structure of the motor control device 157 which concerns on 2nd Embodiment. It is a block diagram which shows the structure of the current controller which concerns on 2nd Embodiment. It is a block diagram which shows the structure of the motor control device which performs speed feedback control.

A preferred embodiment of the present invention will be described below with reference to the drawings. However, the shapes of the components and their relative arrangements described in this embodiment should be appropriately changed depending on the configuration of the apparatus to which the present invention is applied and various conditions, and the scope of the present invention is limited. It is not intended to be limited to the following embodiments. In the following description, the case where the motor control device is provided in the image forming device will be described, but the case where the motor control device is provided is not limited to the image forming device. For example, it is also used in a sheet transfer device for transporting a sheet such as a recording medium or a document.

[First Embodiment]
[Image forming device]
FIG. 1 is a cross-sectional view showing the configuration of a monochrome electrophotographic copying machine (hereinafter, referred to as an image forming apparatus) 100 having a sheet transporting apparatus used in the present embodiment. The image forming apparatus is not limited to the copying machine, and may be, for example, a facsimile apparatus, a printing machine, a printer, or the like. Further, the recording method is not limited to the electrophotographic method, and may be, for example, an inkjet or the like. Further, the format of the image forming apparatus may be either monochrome or color.

Hereinafter, the configuration and function of the image forming apparatus 100 will be described with reference to FIG. As shown in FIG. 1, the image forming apparatus 100 includes a document feeding apparatus 201, a reading apparatus 202, and an image printing apparatus 301.

The documents loaded on the document loading section 203 of the document feeding device 201 are fed one by one by the paper feed roller 204, and are conveyed on the document glass base 214 of the reading device 202 along the transfer guide 206. Further, the original document is conveyed at a constant speed by the conveying belt 208, and is ejected by the paper ejection roller 205 to an output tray (not shown). The reflected light from the original image illuminated by the illumination 209 at the reading position of the reading device 202 is guided to the image reading unit 111 by an optical system including reflection mirrors 210, 211, and 212, and converted into an image signal by the image reading unit 111. Will be done. The image reading unit 111 includes a lens, a CCD that is a photoelectric conversion element, a driving circuit of the CCD, and the like. The image signal output from the image reading unit 111 is output to the image printing device 301 after various correction processes are performed by the image processing unit 112 configured by a hardware device such as an ASIC. As described above, the original is read. That is, the document feeding device 201 and the reading device 202 function as the document reading device.

Further, as the document scanning mode, there are a first scanning mode and a second scanning mode. The first scanning mode is a mode in which an image of a document conveyed at a constant speed is scanned by an illumination system 209 and an optical system fixed at a predetermined position. The second scanning mode is a mode in which the image of the document placed on the document glass 214 of the scanning device 202 is scanned by the illumination system 209 and the optical system that move at a constant speed. Normally, the image of the sheet-shaped original is read in the first reading mode, and the image of the bound original such as a book or booklet is read in the second reading mode.

Sheet storage trays 302 and 304 are provided inside the image printing apparatus 301. Different types of recording media can be stored in the sheet storage trays 302 and 304. For example, the sheet storage tray 302 stores A4 size plain paper, and the sheet storage tray 304 stores A4 size thick paper. The recording medium is one in which an image is formed by an image forming apparatus, and for example, paper, a resin sheet, a cloth, an OHP sheet, a label, and the like are included in the recording medium.

The recording medium stored in the sheet storage tray 302 is fed by the paper feed roller 303 and sent out to the registration roller 308 by the transport roller 306. Further, the recording medium stored in the sheet storage tray 304 is fed by the paper feed roller 305 and sent out to the registration roller 308 by the transport rollers 307 and 306.

The image signal output from the reading device 202 is input to the optical scanning device 311 including the semiconductor laser and the polygon mirror. Further, the outer peripheral surface of the photosensitive drum 309 is charged by the charger 310. After the outer peripheral surface of the photosensitive drum 309 is charged, the laser beam corresponding to the image signal input from the reading device 202 to the optical scanning device 311 is photosensitiveed from the optical scanning device 311 via the polygon mirror and the mirrors 312 and 313. The outer peripheral surface of the drum 309 is irradiated. As a result, an electrostatic latent image is formed on the outer peripheral surface of the photosensitive drum 309. For charging the photosensitive drum, for example, a charging method using a corona charger or a charging roller is used.

Subsequently, the electrostatic latent image is developed by the toner in the developer 314, and the toner image is formed on the outer peripheral surface of the photosensitive drum 309. The toner image formed on the photosensitive drum 309 is transferred to the recording medium by a transfer charger 315 provided at a position (transfer position) facing the photosensitive drum 309. At this transfer timing, the registration roller 308 feeds the recording medium to the transfer position.

As described above, the recording medium on which the toner image is transferred is sent to the fixing device 318 by the transport belt 317, heated and pressurized by the fixing device 318, and the toner image is fixed on the recording medium. In this way, the image forming apparatus 100 forms an image on the recording medium.

When image formation is performed in the single-sided printing mode, the recording medium that has passed through the fixing device 318 is ejected to an output tray (not shown) by the output rollers 319 and 324. When the image is formed in the double-sided printing mode, the recording medium is the paper ejection roller 319, the transport roller 320, and the reversing roller 321 after the fixing process is performed on the first surface of the recording medium by the fixing device 318. Is transported to the reverse path 325. After that, the recording medium is conveyed to the registration roller 308 again by the conveying rollers 322 and 323, and an image is formed on the second surface of the recording medium by the method described above. After that, the recording medium is ejected to an output tray (not shown) by the output rollers 319 and 324.

When the recording medium on which the image is formed on the first surface is discharged face-down to the outside of the image forming apparatus 100, the recording medium that has passed through the fixing device 318 passes through the paper ejection roller 319 and is conveyed to the roller 320. It is transported in the direction toward. After that, just before the rear end of the recording medium passes through the nip portion of the transfer roller 320, the rotation of the transfer roller 320 is reversed, so that the recording medium is a paper ejection roller with the first surface of the recording medium facing downward. It is discharged to the outside of the image forming apparatus 100 via 324.

The above is a description of the configuration and function of the image forming apparatus 100. The load in the present invention is an object driven by a motor. For example, various rollers (conveying rollers) such as paper feed rollers 204, 303, 305, registration rollers 308 and paper ejection rollers 319, photosensitive drums 309, transport belts 208, 317, lighting systems 209, optical systems and the like are in the present invention. Corresponds to the load. The motor control device of the present embodiment can be applied to a motor that drives these loads.

FIG. 2 is a block diagram showing an example of a control configuration of the image forming apparatus 100. As shown in FIG. 2, the system controller 151 includes a CPU 151a, a ROM 151b, and a RAM 151c. Further, the system controller 151 is connected to an image processing unit 112, an operation unit 152, an analog / digital (A / D) converter 153, a high voltage control unit 155, a motor control device 157, sensors 159, and an AC driver 160. .. The system controller 151 can send and receive data and commands to and from each connected unit.

The CPU 151a executes various sequences related to a predetermined image formation sequence by reading and executing various programs stored in the ROM 151b.

The RAM 151c is a storage device. The RAM 151c stores, for example, various data such as a set value for the high-voltage control unit 155, a command value for the motor control device 157, and information received from the operation unit 152.

The system controller 151 transmits to the image processing unit 112 the set value data of various devices provided inside the image forming device 100, which is necessary for the image processing in the image processing unit 112. Further, the system controller 151 receives the signals from the sensors 159 and sets the set value of the high voltage control unit 155 based on the received signals. The high-voltage control unit 155 supplies the voltage required for the high-voltage unit 156 (charger 310, developer 314, transfer charger 315, etc.) according to the set value set by the system controller 151. The sensors 159 include sensors and the like that detect the recording medium conveyed by the transfer roller.

The motor control device 157 controls the motor 509 that drives the load in response to a command output from the CPU 151a. Although only the motor 509 is described as the motor of the image forming apparatus in FIG. 2, it is assumed that a plurality of motors are actually provided in the image forming apparatus. Further, one motor control device may be configured to control a plurality of motors. Further, in FIG. 2, although only one motor control device is provided, it is assumed that a plurality of motor control devices are actually provided in the image forming device.

The A / D converter 153 receives the detection signal detected by the thermistor 154 for detecting the temperature of the fixing heater 161, converts the detection signal from an analog signal to a digital signal, and transmits the detection signal to the system controller 151. The system controller 151 controls the AC driver 160 based on the digital signal received from the A / D converter 153. The AC driver 160 controls the fixing heater 161 so that the temperature of the fixing heater 161 becomes a temperature required for performing the fixing process. The fixing heater 161 is a heater used for the fixing process, and is included in the fixing device 318.

The system controller 151 displays the operation screen for the user to set the type of recording medium to be used (hereinafter referred to as paper type) on the display unit provided in the operation unit 152. To control. The system controller 151 receives the information set by the user from the operation unit 152, and controls the operation sequence of the image forming apparatus 100 based on the information set by the user. Further, the system controller 151 transmits information indicating the state of the image forming apparatus to the operation unit 152. The information indicating the state of the image forming apparatus is, for example, information on the number of images formed, the progress of the image forming operation, jams and double feeds of sheet materials in the document reading apparatus 201 and the image printing apparatus 301. The operation unit 152 displays the information received from the system controller 151 on the display unit.

As described above, the system controller 151 controls the operation sequence of the image forming apparatus 100.

[Motor control device]
Next, the motor control device in this embodiment will be described. The motor control device in the present embodiment can control the motor by any control method of vector control as the first control mode and constant current control as the second control mode.

<Vector control>
First, a method in which the motor control device 157 in the present embodiment performs vector control will be described with reference to FIGS. 3 and 4. It is assumed that the motor in the following description is not provided with a sensor such as a rotary encoder for detecting the rotation phase of the rotor of the motor.

FIG. 3 shows a stepping motor (hereinafter referred to as a motor) 509 composed of two phases, A phase (first phase) and B phase (second phase), and a rotating coordinate system represented by the d-axis and the q-axis. It is a figure which shows the relationship of. In FIG. 3, in the rest coordinate system, the α-axis, which is the axis corresponding to the A-phase winding, and the β-axis, which is the axis corresponding to the B-phase winding, are defined. Further, in FIG. 3, the d-axis is defined along the direction of the magnetic flux generated by the magnetic poles of the permanent magnet used in the rotor 402, and the direction is 90 degrees counterclockwise from the d-axis (perpendicular to the d-axis). The q-axis is defined along the direction of the magnet. The angle formed by the α-axis and the d-axis is defined as θ, and the rotation phase of the rotor 402 is represented by the angle θ. In vector control, a rotating coordinate system based on the rotation phase θ of the rotor 402 is used. Specifically, in vector control, the q-axis component (torque current component) and the winding, which are the current components in the rotation coordinate system of the current vector corresponding to the drive current flowing in the winding, and generate torque in the rotor. A d-axis component (exciting current component) that affects the strength of the magnetic flux penetrating is used.

Vector control is a motor by performing phase feedback control that controls the value of the torque current component and the value of the exciting current component so that the deviation between the command phase representing the target phase of the rotor and the actual rotation phase becomes small. It is a control method to control. In addition, the motor is controlled by performing speed feedback control that controls the value of the torque current component and the value of the exciting current component so that the deviation between the command speed representing the target speed of the rotor and the actual rotation speed becomes small. There is also a method.

FIG. 4 is a block diagram showing an example of the configuration of the motor control device 157 that controls the motor 509. The motor control device 157 is composed of at least one ASIC, and executes each function described below.

As shown in FIG. 4, the motor control device 157 includes a constant current controller 700 that performs constant current control and a vector controller 701 that performs vector control.

The motor control device 157 includes a phase controller 502, a current controller 503, 504, a coordinate inverse converter 505, a coordinate converter 511, a PWM inverter 506 that supplies a drive current to the windings of the motor, and the like as circuits that perform vector control. Has. The coordinate converter 511 represents the current vector corresponding to the drive current flowing through the windings of the A phase and the B phase of the motor 509 from the stationary coordinate system represented by the α axis and the β axis by the q axis and the d axis. Convert coordinates to a rotating coordinate system. As a result, the drive current flowing through the winding is represented by the current value of the q-axis component (q-axis current) and the current value of the d-axis component (d-axis current), which are current values in the rotating coordinate system. The q-axis current corresponds to the torque current that generates torque in the rotor 402 of the motor 509. Further, the d-axis current corresponds to an exciting current that affects the strength of the magnetic flux penetrating the winding of the motor 509, and does not contribute to the generation of torque of the rotor 402. The motor control device 157 can independently control the q-axis current and the d-axis current. As a result, the motor control device 157 can efficiently generate the torque required for the rotor 402 to rotate by controlling the q-axis current according to the load torque applied to the rotor 402. That is, in vector control, the magnitude of the current vector shown in FIG. 3 changes according to the load torque applied to the rotor 402.

The motor control device 157 determines the rotation phase θ of the rotor 402 of the motor 509 by a method described later, and performs vector control based on the determination result. The CPU 151a generates a command phase θ_ref representing the target phase of the rotor 402 of the motor 509, and outputs the command phase θ_ref to the motor control device 157 in a predetermined time period T.

The subtractor 101 calculates the deviation between the rotation phase θ of the rotor 402 of the motor 509 and the command phase θ_ref, and outputs the deviation to the phase controller 502.

FIG. 5 is a block diagram showing the configuration of the phase controller 502. The configuration of the phase controller 502 shown in FIG. 5 is an example of the present embodiment, and the configuration of the phase controller 502 is not limited to this.

As shown in FIG. 5, the phase controller 502 includes a proportional control unit 502a that performs proportional control (P), an integral control unit 502b that performs integral control (I), and a differential control unit 502c that performs differential control (D). .. Further, the phase controller 502 includes an adder 502d that adds signals output from the proportional control unit 502a, the integral control unit 502b, and the differential control unit 502c. Further, the phase controller 502 includes a q-axis current generator 502e that generates a q-axis current command value (target value) iq_ref based on the signal output from the adder 502d, and a d-axis current command value (target value) id_ref. It has a d-axis current generation unit 502f for generating the above.

The phase controller 502 generates a q-axis current command value iq_ref based on the proportional control (P), the integral control (I), and the differential control (D) so that the deviation output from the subtractor 101 becomes small. And output. Specifically, the phase controller 502 generates and outputs the q-axis current command value iq_ref so that the deviation output from the subtractor 101 becomes 0 based on the P control, the I control, and the D control.

More specifically, the proportional control unit 502a outputs a value proportional to the deviation so that the deviation output from the subtractor 101 becomes zero. Further, the integration control unit 502b outputs a value proportional to the time integration of the deviation so that the deviation output from the subtractor 101 becomes zero. Further, the differential control unit 502c outputs a value proportional to the time change of the deviation so that the deviation output from the subtractor 101 becomes zero. The current value iq'input to the integration control unit 502b will be described later.

Then, the adder 502d adds the values output from the proportional control unit 502a, the integral control unit 502b, and the differential control unit 502c, and the added values are output to the q-axis current generation unit 502e. The q-axis current generation unit 502e generates and outputs the q-axis current command value iq_ref based on the value output from the adder 502d. Specifically, for example, the q-axis current command value iq_ref is generated and output by multiplying the value output from the adder 502d by a preset proportional coefficient.

Further, the d-axis current generation unit 502f sets the d-axis current command value id_ref to 0 and outputs the current. In the present embodiment, the d-axis current generation unit 502f sets the d-axis current command value id_ref, which affects the strength of the magnetic flux penetrating the winding, to 0, but the present invention is not limited to this. For example, the d-axis current generation unit 502f may set the d-axis current command value id_ref to a value other than 0 and output it based on the command from the CPU 151a.

The phase controller 502 in the present embodiment generates the q-axis current command value iq_ref based on the PID control, but the present invention is not limited to this. For example, the phase controller 502 may generate the q-axis current command value iq_ref based on PI control.

The drive currents flowing through the A-phase and B-phase windings of the motor 509 are detected by the current detectors 507 and 508, and then converted from analog values to digital values by the A / D converter 510. The cycle in which the current detectors 507 and 508 detect the current may be, for example, the same cycle as the cycle T in which the CPU 151a outputs the command phase θ_ref to the motor control device 157, or may be shorter than the cycle T.

The current value of the drive current converted from the analog value to the digital value by the A / D converter 510 is determined by the following equation using the phase θe of the current vector shown in FIG. 3 as the current values iα and iβ in the stationary coordinate system. expressed. The phase θe of the current vector is defined as the angle formed by the α axis and the current vector. Further, I indicates the magnitude of the current vector.
iα = I * cosθe (1)
iβ = I * sinθe (2)
These current values iα and iβ are input to the coordinate converters 511 and 517 and the induced voltage determinant 512.

The coordinate converter 511 converts the current values iα and iβ in the stationary coordinate system into the current value iq of the q-axis current and the current value id of the d-axis current in the rotating coordinate system by the following equation.
id = cosθ * iα + sinθ * iβ (3)
iq = −sinθ * iα + cosθ * iβ (4)
In vector control, the q-axis current command value iq_ref output from the phase controller 502 is input to the subtractor 102 via the changeover switch 516a. Further, the current value iq output from the coordinate converter 511 is input to the subtractor 102. The subtractor 102 calculates the deviation between the q-axis current command value iq_ref and the current value iq, and outputs the deviation to the current controller 503.

Further, in the vector control, the d-axis current command value id_ref output from the phase controller 502 is input to the subtractor 103 via the changeover switch 516a. Further, the current value id output from the coordinate converter 511 is input to the subtractor 103. The subtractor 103 calculates the deviation between the d-axis current command value id_ref and the current value id, and outputs the deviation to the current controller 504. The changeover switch 516a will be described later.

FIG. 6 is a block diagram showing the configuration of the current controller 503. The configuration of the current controller 503 shown in FIG. 6 is an example in the present embodiment, and the configuration of the current controller 503 is not limited to this.

As shown in FIG. 6, the current controller 503 includes a proportional control unit 503a that performs P control, an integral control unit 503b that performs I control, and a differential control unit 503c that performs D control. Further, the current controller 503 includes an adder 503d that adds signals output from the proportional control unit 502a, the integral control unit 502b, and the differential control unit 502c. Further, the current controller 503 generates a drive voltage Vq as the value of the q-axis component of the voltage vector corresponding to the drive voltage to be applied to the winding of the motor 509 based on the signal output from the adder 503d. It has a generation unit 503e.

The current controller 503 generates the drive voltage Vq based on the PID control so that the deviation output from the subtractor 102 becomes small. Specifically, the current controller 503 generates a drive voltage Vq so that the deviation output from the subtractor 102 becomes 0, and outputs the drive voltage Vq to the coordinate inverse converter 505.

More specifically, the proportional control unit 503a outputs a value proportional to the deviation so that the deviation output from the subtractor 102 becomes zero. Further, the integration control unit 503b outputs a value proportional to the time integration of the deviation so that the deviation output from the subtractor 102 becomes zero. Further, the differential control unit 503c outputs a value proportional to the time change of the deviation so that the deviation output from the subtractor 102 becomes zero.

Then, the adder 503d adds the values output from the proportional control unit 503a, the integral control unit 503b, and the differential control unit 503c, and the added values are output to the drive voltage generation unit 503e. The drive voltage generation unit 503e generates and outputs the drive voltage Vq based on the value output from the adder 503d. Specifically, for example, the drive voltage Vq is generated and output by multiplying the value output from the adder 503d by a preset proportional coefficient.

In this way, the current controller 503 functions as a generation means for generating the drive voltage. The current controller 504 has the same configuration as the current controller 503, and the d-axis component of the voltage vector corresponding to the drive voltage to be applied to the winding of the motor 509 by the same method as the current controller 503. The drive voltage Vd is generated as the value of.

The current controllers 503 and 504 in the present embodiment generate drive voltages Vq and Vd based on PID control, but the present invention is not limited to this. For example, the current controller 503 may generate drive voltages Vq and Vd based on PI control.

The coordinate inverse converter 505 reversely converts the drive voltages Vq and Vd in the rotating coordinate system output from the current controllers 503 and 504 into the drive voltages Vα and Vβ in the rest coordinate system by the following equation.
Vα = cosθ * Vd-sinθ * Vq (5)
Vβ = sinθ * Vd + cosθ * Vq (6)
The coordinate inverse converter 505 outputs the inversely transformed drive voltages Vα and Vβ to the induced voltage determinant 512 and the PWM inverter 506.

The PWM inverter 506 has a full bridge circuit. The full bridge circuit is driven by a PWM signal based on the drive voltages Vα and Vβ input from the coordinate inverse converter 505. As a result, the PWM inverter 506 generates drive currents iα and iβ corresponding to the drive voltages Vα and Vβ, and supplies the drive currents iα and iβ to the windings of each phase of the motor 509 to drive the motor 509. .. That is, the PWM inverter 506 functions as a supply means for supplying a current to the windings of each phase of the motor 509. In the present embodiment, the PWM inverter has a full bridge circuit, but the PWM inverter may be a half bridge circuit or the like.

Next, a configuration for determining the rotation phase θ will be described. In determining the rotation phase θ of the rotor 402, the values of the induced voltages Eα and Eβ induced in the windings of the A phase and the B phase of the motor 509 by the rotation of the rotor 402 are used. The value of the induced voltage is determined (calculated) by the induced voltage determinant 512. Specifically, the induced voltages Eα and Eβ were input to the induced voltage determinant 512 from the coordinate inverse converter 505 and the current values iα and iβ input from the A / D converter 510 to the induced voltage determinant 512. It is determined by the following equation from the drive voltages Vα and Vβ.
Eα = Vα-R * iα-L * diα / dt (7)
Eβ = Vβ-R * iβ-L * diβ / dt (8)
Here, R is the winding resistance and L is the winding inductance. The values of the winding resistance R and the winding inductance L are values peculiar to the motor 509 used, and are stored in advance in a memory (not shown) provided in the ROM 151b or the motor control device 600 or the like.

The induced voltages Eα and Eβ determined by the induced voltage determinant 512 are output to the phase determinant 513.

The phase determinant 513 determines the rotation phase θ of the rotor 402 of the motor 509 by the following equation based on the ratio of the induced voltage Eα and the induced voltage Eβ output from the induced voltage determinant 512.
θ = tan ^ -1 (-Eβ / Eα) (9)

In the present embodiment, the phase determinant 513 determines the rotation phase θ by performing an operation based on the equation (9), but this is not the case. For example, the phase determinant 513 rotates by referring to a table stored in ROM 151b or the like showing the relationship between the induced voltage Eα and the induced voltage Eβ and the rotation phase θ corresponding to the induced voltage Eα and the induced voltage Eβ. The phase θ may be determined.

The rotation phase θ of the rotor 402 obtained as described above is input to the subtractor 101, the changeover switch 516b, and the coordinate converter 517. When vector control is performed, the rotation phase θ is input to the coordinate inverse converter 505 and the coordinate converter 511 via the changeover switch 516b. The changeover switch 516b and the coordinate converter 517 will be described later.

When the motor control device 157 performs vector control, the motor control device 157 repeats the above-mentioned control.

As described above, the motor control device 157 in the present embodiment performs vector control using phase feedback control that controls the current value in the rotating coordinate system so that the deviation between the command phase θ_ref and the rotation phase θ becomes small. By performing vector control, it is possible to suppress the motor from being out of step, the increase in motor noise due to excess torque, and the increase in power consumption. Further, by performing the phase feedback control, the rotation phase of the rotor can be controlled so that the rotation phase of the rotor becomes a desired phase. Therefore, in the image forming apparatus, the vector control by the phase feedback control is applied to the motor that drives the load (registration roller, etc.) that needs to accurately control the rotation phase of the rotor, so that the image on the recording medium is obtained. The formation can be performed properly.

<Constant current control>
Next, the constant current control in the present embodiment will be described in comparison with the conventional constant current control.

In constant current control, a drive current flowing through the windings is controlled by supplying a predetermined current to the windings of the motor based on the operation sequence of the motor. In constant current control, it corresponds to the torque that a predetermined margin is added to the torque that is assumed to be required for the rotation of the rotor so that the motor does not step out even if the load torque applied to the rotor fluctuates. A drive current with an amplitude is supplied. This is because in constant current control, a configuration in which the amplitude of the drive current is controlled based on the determined (estimated) rotation phase and rotation speed of the rotor is not used (feedback control is not performed). This is because the drive current cannot be adjusted according to the load torque applied to the child. The larger the amplitude of the current, the larger the torque applied to the rotor. Also, the amplitude corresponds to the magnitude of the current vector.

In the following description, during constant current control, the motor is controlled by supplying a current having a constant amplitude to the windings of the motor, but this is not the case. For example, during constant current control, the motor may be controlled by supplying a current having a predetermined amplitude to the windings of the motor according to each of acceleration and deceleration of the motor.

FIG. 7 is a block diagram showing an example of a conventional constant current control control configuration. First, the conventional constant current control will be described.

The CPU 151a outputs the command phase θ_ref to the constant current controller 801. The constant current controller 801 generates and outputs the command values iα_ref and iβ_ref of the current in the rest coordinate system corresponding to the command phase θ_ref output from the CPU 151a. In the present embodiment, the magnitude of the current vector corresponding to the command values iα_ref and iβ_ref of the current in the stationary coordinate system is always constant.

The drive currents flowing through the A-phase and B-phase windings of the motor 509 are detected by the current detectors 806 and 807. The detected drive current is converted from an analog value to a digital value by the A / D converter 809, and is expressed as current values iα and iβ as in equations (1) and (2).

The current value iα output from the A / D converter 809 and the current command value iα_ref output from the constant current controller 801 are input to the subtractor 802. The subtractor 102 calculates the deviation between the current command value iα_ref and the current value iα, and outputs the deviation to the current controller 804.

Further, the current value iβ output from the A / D converter 809 and the current command value iβ_ref output from the constant current controller 801 are input to the subtractor 803. The subtractor 803 calculates the deviation between the current command value iβ_ref and the current value iβ, and outputs the deviation to the current controller 804.

The current controller 804 outputs the drive voltages Vα and Vβ based on the PID control so that the input deviation becomes small. Specifically, the current controller 804 outputs the drive voltages Vα and Vβ so that the input deviation approaches zero.

The PWM inverter 506 supplies a drive current to the windings of each phase of the motor 509 to drive the motor 509 based on the input drive voltages Vα and Vβ by the method described above.

As described above, in the conventional constant current control, the current values iα and iβ in the rest coordinate system are used.

Next, the constant current control in this embodiment will be described.

FIG. 8 is a diagram illustrating constant current control in the present embodiment. The dc axis shown in FIG. 8 indicates a direction in which the command phase θ_ref advances counterclockwise from the α axis, and the qc axis indicates a direction 90 degrees forward in the counterclockwise direction from the dc axis (direction orthogonal to the dc axis). In the constant current control in this embodiment, a rotating coordinate system represented by the dc axis and the qc axis based on the command phase θ_ref is used. Specifically, in the constant current control in the present embodiment, as shown in FIG. 8, the phase θe of the current vector corresponding to the drive current supplied to the winding is set to θ_ref. That is, the drive current supplied to the winding is generated so that the direction of the current vector corresponding to the drive current supplied to the winding coincides with the dc axis. In FIG. 8, the configuration of the winding of the motor and the like as shown in FIG. 3 is omitted.

FIG. 9 is a block diagram showing an example of the configuration of the constant current controller 700 shown in FIG. As shown in FIG. 9, the constant current controller 700 includes a current generator 700a and a coordinate converter 700b.

Hereinafter, a method in which the motor control device 157 in the present embodiment performs constant current control will be described with reference to FIGS. 4, 8 and 9.

The CPU 151a outputs the command phase θ_ref to the current generator 700a and the coordinate converter 700b provided in the constant current controller 700. The current generator 700a generates command values iα_ref and iβ_ref of the current in the stationary coordinate system corresponding to the command phase θ_ref output from the CPU 151a and outputs the command values iα_ref and iβ_ref to the coordinate converter 700b.

The coordinate converter 700b sets the command values iα_ref and iβ_ref of the current in the stationary coordinate system to the command values iq_ref and d-axis of the q-axis current in the rotating coordinate system based on the command phase θ_ref according to the equations (3) and (4). It is converted to the current command value id_ref and output. In the present embodiment, the magnitude of the current vector corresponding to the current command values iα_ref and iβ_ref (the magnitude of the current vector corresponding to the q-axis current command value iq_ref and the d-axis current command value id_ref) is always set. It is constant.

The drive currents flowing through the A-phase and B-phase windings of the motor 509 are detected by the current detectors 507 and 508. The detected drive current is converted from an analog value to a digital value by the A / D converter 510, and is represented as current values iα and iβ as in equations (1) and (2).

In the constant current control, the command phase θ_ref is input to the coordinate converter 511 via the switch 516b. The coordinate converter 511 uses the equations (3) and (4) to convert the current values iα and iβ in the stationary coordinate system output from the A / D converter 510 into the q-axis in the rotational coordinate system based on the command phase θ_ref. It is converted into the current value iq of the current and the current value id of the d-axis current.

In the constant current control, the q-axis current command value iq_ref output from the constant current controller 700 is input to the subtractor 102 via the changeover switch 516a. Further, the current value iq output from the coordinate converter 511 is input to the subtractor 102. The subtractor 102 calculates the deviation between the q-axis current command value iq_ref and the current value iq, and outputs the deviation to the current controller 503.

Further, in the constant current control, the d-axis current command value id_ref output from the constant current controller 700 is input to the subtractor 103 via the changeover switch 516a. Further, the current value id output from the coordinate converter 511 is input to the subtractor 103. The subtractor 103 calculates the deviation between the d-axis current command value id_ref and the current value id, and outputs the deviation to the current controller 504. The changeover switch 516a will be described later.

The current controllers 503 and 504 output the drive voltages Vq and Vd in the rotating coordinate system based on the command phase θ_ref so that the input deviation becomes small. Specifically, the current controller 503 outputs the drive voltages Vq and Vd so that the input deviation approaches 0.

In the constant current control, the command phase θ_ref is input to the coordinate inverse converter 505 via the switch 516b. The coordinate inverse transformor 505 sets the drive voltages Vq and Vd in the rotating coordinate system based on the command phase θ_ref output from the current controllers 503 and 504 in the stationary coordinate system according to the equations (5) and (6). It is converted back to the drive voltage Vα and Vβ.

The coordinate inverse converter 505 outputs the inversely transformed drive voltages Vα and Vβ to the PWM inverter 506. The PWM inverter 506 supplies a drive current to the windings of each phase of the motor 509 to drive the motor 509 by the method described above.

As described above, in the constant current control in the present embodiment, the rotating coordinate system represented by the dc axis and the qc axis based on the command phase θ_ref is used.

Further, in the constant current control in the present embodiment, neither the phase feedback control nor the speed feedback control is performed. That is, in the constant current control in this embodiment, the drive current supplied to the winding is not adjusted according to the rotation state of the rotor. Therefore, in the constant current control, a current obtained by adding a predetermined margin to the current required to rotate the rotor is supplied to the winding so that the motor does not step out. Specifically, the command values iα_ref and iβ_ref of the current in the stationary coordinate system include a current value required for rotating the rotor and a current value corresponding to a predetermined margin.

<Switching between vector control and constant current control>
Next, a method of switching between vector control and constant current control will be described. As shown in FIG. 4, the motor control device 157 in the present embodiment has a configuration for switching between constant current control and vector control. Specifically, the motor control device 157 includes a control switch 515, changeover switches 516a and 516b, and a delay circuit 518. A circuit that performs vector control is also in operation during the period of constant current control. That is, the circuit for determining the rotation phase θ of the rotor is operating during the period during which the constant current control is performed. On the other hand, during the period during which the vector control is performed, the circuit that performs the constant current control may be in operation or may be stopped.

As shown in FIG. 4, a rotation speed ω_ref'instead of the command speed of the rotor determined by the CPU 151a based on the command phase θ_ref is input to the control switch 515. The control switch 515 switches between constant current control and vector control by comparing the rotation speed ω_ref'and the threshold value ωth, and further outputs a switching signal indicating the control switching. The CPU 151a determines the rotation speed ω_ref'based on the amount of change in the command phase θ_ref in a predetermined period. That is, the rotation speed ω_ref'changes in a predetermined time period T.

FIG. 10 is a diagram showing the relationship between the rotation speed ω_ref'and the threshold value ωth and the switching signal. The threshold value ωth in the present embodiment is set to the smallest rotation speed among the rotation speeds at which the rotation phase θ is accurately determined, but the present invention is not limited to this. For example, the threshold value ωth may be set to a value equal to or higher than the smallest rotation speed among the rotation speeds at which the rotation phase θ is accurately determined.

As shown in FIG. 10, the control switch 515 sets the switching signal to ‘H’ when constant current control is performed, and sets the switching signal to ‘L’ when vector control is performed. The switching signal output from the control switch 515 is input to the phase controller 502 and the delay circuit 518, as shown in FIG. The control switch 515 outputs a switching signal in the same cycle as the cycle T in which the CPU 151a outputs the rotation speed ω_ref', for example.

The delay circuit 518 outputs the input switching signal after a predetermined delay time after the switching signal is output from the control switch 515. The predetermined delay time is longer than the time from when the switching signal is output from the control switching device 515 until the phase controller 502 outputs iq_ref and id_ref in response to the switching signal. The configuration in which the phase controller 502 outputs iq_ref and id_ref according to the switching signal will be described later.

During control by the constant current controller 700, when the rotation speed ω_ref'is equal to or higher than the threshold value ωth (ω_ref'≥ωth), the control switch 515 switches the controller that controls the motor 509. That is, the control switch 515 switches the switching signal from ‘H’ to ‘L’ and outputs it so that the controller that controls the motor 509 is switched from the constant current controller 700 to the vector controller 701. The delay circuit 518 outputs the input changeover signal to the changeover switches 516a and 516b after a predetermined delay time after the changeover signal is output from the control switch 515. As a result, the states of the changeover switches 516a and 516b are switched according to the changeover signal, and vector control is performed by the vector controller 701. The threshold value ωth is stored in advance in, for example, ROM 151b.

Further, when the rotation speed ω_ref'is smaller than the threshold value ωth (ω_ref'<ωth) during the control by the constant current controller 700, the control switch 515 does not switch the controller that controls the motor 509. That is, the control switch 515 outputs the switching signal ‘H’ so that the motor 509 maintains the state controlled by the constant current controller 700. The delay circuit 518 outputs the input changeover signal to the changeover switches 516a and 516b after a predetermined delay time after the changeover signal is output from the control switch 515. As a result, the state of the changeover switches 516a and 516b is maintained, and the constant current control by the constant current controller 700 is continued.

During control by the vector controller 701, when the rotation speed ω_ref'is smaller than the threshold value ωth (ω_ref'<ωth), the control switch 515 switches the controller that controls the motor 509. That is, the control switch 515 switches the switching signal from ‘L’ to ‘H’ and outputs it so as to switch the controller that controls the motor 509 from the vector controller 701 to the constant current controller 700. The delay circuit 518 outputs the input changeover signal to the changeover switches 516a and 516b after a predetermined delay time after the changeover signal is output from the control switch 515. As a result, the states of the changeover switches 516a and 516b are switched, and the constant current controller 700 performs constant current control.

Further, when the rotation speed ω_ref'is equal to or higher than the threshold value ωth (ω_ref'≥ωth) during the control by the vector controller 701, the control switch 515 does not switch the controller that controls the motor 509. That is, the control switch 515 outputs the switching signal ‘L’ so that the motor 509 maintains the state controlled by the vector controller 701. The delay circuit 518 outputs the input changeover signal to the changeover switches 516a and 516b after a predetermined delay time after the changeover signal is output from the control switch 515. As a result, the state of the changeover switches 516a and 516b is maintained, and the vector control by the vector controller 701 is continued.

<Processing when switching control>
Next, the process performed by the motor control device 157 when the motor control method is switched from the constant current control to the vector control will be described.

As described above, when the control of the motor is switched from the constant current control to the vector control, the rotation speed of the motor may decrease (or increase) momentarily. This is because there is a difference between the torque generated in the rotor immediately before the control of the motor is switched and the torque generated in the rotor immediately after the control of the motor is switched.

Therefore, in the present embodiment, the following configuration is applied to the motor control device 157 to prevent the motor control from becoming unstable.

As shown in FIG. 4, the motor control device 157 in this embodiment is provided with a coordinate converter 517. In the following description, it is assumed that the coordinate converter 517 is in operation during the period during which the constant current control is performed. Further, the coordinate converter 517 may be in operation or may be stopped during the period in which the vector control is being performed. Further, in the following description, it is assumed that the phase controller 502 is operating during the period during which the constant current control is performed.

As shown in FIG. 4, the current values iα and iβ output from the A / D converter 510 and the rotation phase θ output from the phase determinant 513 are input to the coordinate converter 517. The coordinate converter 517 uses the equations (3) and (4) to set the current values iα and iβ based on the rotation phase θ output from the phase determinant 513, and uses the rotation phase θ as a reference for the rotation coordinate system. Convert to current values iq'and id'. The current value iq'converted by the coordinate converter 517 is input to the phase controller 502. Specifically, as shown in FIG. 5, the current value iq'is input to the integration control unit 502b provided inside the phase controller 502. The coordinate converter 517 outputs iq'in the same cycle as the current detectors 507 and 508 detect the current.

When the switching signal is switched from ‘H’ to ‘L’, the integration control unit 502b outputs the control result of the integration control unit 502b based on the current value iq ′ acquired immediately before the switching signal is switched. Specifically, when the switching signal is switched from'H'to'L', the integration control unit 502b calculates the value obtained by multiplying the current value iq'acquired immediately before the switching signal is switched by the proportional coefficient Kq. It is set as the initial value of the integral of 502b. That is, when the switching signal is switched from'H'to'L', the integration control unit 502b deletes the integration result up to immediately before the switching signal is switched, and sets the current value iq'acquired immediately before the switching signal is switched. The value obtained by multiplying the proportional coefficient Kq is set as the initial value of integration. The proportional coefficient Kq is a coefficient for preventing the value corresponding to the margin added to the current from being included in the current value iq'so that the motor does not step out in the constant current control. Therefore, by multiplying the proportional coefficient Kq by the current value iq', the integration control unit 502b can perform integration control based on a value that does not include a value corresponding to the margin. As a result, the torque given to the rotor immediately after the motor control is switched from constant current control to vector control is more appropriate than when the current value iq', which is not multiplied by the proportional coefficient Kq, is set as the initial value. It becomes torque. After the switching signal is output from the control switching device 515, the predetermined delay time for the delay circuit 518 to delay the switching signal is longer than the time for the phase controller 502 to perform the above-mentioned processing, and the switching signal switches the control. The time is shorter than the cycle output from the device 515.

FIG. 11 is a flowchart showing a method of controlling the motor by the motor control device 157. The control of the motor 509 according to the present embodiment will be described below with reference to FIG. The processing of this flowchart is executed by the motor control device 157 that receives the instruction from the CPU 151a.

First, when the enable signal ‘H’ is output from the CPU 151a to the motor control device 157, the motor control device 157 starts driving the motor 509 based on the command output from the CPU 151a. The enable signal is a signal that permits or prohibits the operation of the motor control device 157. When the enable signal is'L (low level)', the CPU 151a prohibits the operation of the motor control device 157. That is, the control of the motor 509 by the motor control device 157 is terminated. When the enable signal is'H (high level)', the CPU 151a permits the operation of the motor control device 157, and the motor control device 157 controls the motor 509 based on the command output from the CPU 151a. ..

Next, in S1001, the control switch 515 outputs a switching signal ‘H’ so that the drive of the motor 509 is controlled by the constant current controller 517. As a result, constant current control is performed by the constant current controller 700.

After that, in S1002, when the CPU 151a outputs the enable signal ‘L’ to the motor control device 157, the motor control device 157 ends the driving of the motor 509.

Further, in S1002, when the CPU 151a outputs the enable signal ‘H’ to the motor control device 157, the motor control device 157 advances the process to S1003.

Next, in S1003, when the rotation speed ω_ref'is less than the threshold value ωth, the process returns to S1001 again. That is, the constant current control by the constant current controller 700 is maintained.

Further, in S1003, when the rotation speed ω_ref'is equal to or higher than the threshold value ωth, in S1004, the control switch 515 switches the switching signal from ‘H’ to ‘L’ and outputs it.

After that, in S1005, the integration control unit 502b deletes the integration result until just before the switching signal is switched to'L', sets the current value iq'acquired immediately before the switching signal is switched as the initial value of integration, and outputs the result. do.

Then, when the predetermined delay time elapses in S1006, the changeover signal ‘L’ is output from the delay circuit 518 to the changeover switches 516a and 516b in S1007. As a result, vector control is performed by the vector controller 701.

In S1008, when the rotation speed ω_ref'is equal to or higher than the threshold value ωth, the process returns to S1007 again, and the vector control by the vector controller 701 is continued.

Further, in S1008, when the rotation speed ω_ref'is smaller than the threshold value ωth, the process returns to S1001 again, and the control switch 515 switches the controller that controls the drive of the motor 509. That is, the control switch 515 switches the switching signal from ‘L’ to ‘H’ and outputs it so as to switch the controller that controls the motor 509 from the vector controller 701 to the constant current controller 700. The delay circuit 518 outputs the input changeover signal to the changeover switches 516a and 516b after a predetermined delay time after the changeover signal is output from the control switch 515. As a result, the states of the changeover switches 516a and 516b are switched, and the constant current controller 700 performs constant current control.

After that, the motor control device 157 repeats the above-mentioned control until the CPU 151a outputs the enable signal ‘L’ to the motor control device 157. Even during vector control, if the CPU 151a outputs the enable signal ‘L’ to the motor control device 157, the motor control device 157 stops the control of the motor.

As described above, in the present embodiment, the current value iq ′ corresponding to the load torque applied to the rotor is determined based on the current value detected during the constant current control. Then, the q-axis current command value iq_ref immediately after the switching signal is switched from ‘H’ to ‘L’ is generated based on the current value iq ′ immediately before the switching signal is switched from ‘H’ to ‘L’. Specifically, a value based on the current value iq'as a current detection result is set as an initial value of integration in the phase controller 502. As a result, the torque generated in the rotor corresponding to the current supplied immediately before the control of the motor is switched and the torque generated in the rotor corresponding to the current supplied immediately after the control of the motor is switched are changed. It is possible to suppress the occurrence of a difference. As a result, it is possible to prevent the rotation speed of the motor from fluctuating when the control of the motor is switched from the constant current control to the vector control.

In addition, by setting the detected current value as the initial value of integration, the load torque is calculated, and the drive current to be supplied to the winding is set based on the calculated load torque in a shorter time. , The drive current to be supplied to the winding can be set. As a result, it is possible to suppress the difference between the torque corresponding to the current to be supplied determined based on the load torque immediately before the motor control is switched and the load torque immediately after the motor control is switched as much as possible. be able to. That is, even if the load torque fluctuates when the motor control is switched from the constant current control to the vector control, it is possible to suppress the fluctuation of the rotation speed of the motor, and the motor control becomes unstable. Can be suppressed.

FIG. 12 is a diagram showing experimental results showing changes in the drive current, the waveform of the drive voltage, and the rotation speed due to the switching of the control method of the motor. The rotation speed in FIG. 12 was measured by attaching a rotary encoder to the motor for an experiment.

FIG. 12A shows the current values iα and iβ of the stationary coordinate system in a state where the present embodiment is not applied, that is, the initial value of the integration control unit 502b is not set based on the current value iq ′ during constant current control. , The figure which shows the rotation speed ω. As shown in FIG. 12A, when this embodiment is not applied, the initial value of the integration control unit 502b is 0, so that the current value immediately after switching from constant current control to vector control becomes approximately 0. There is. This means that no torque is generated in the rotor immediately after switching from constant current control to vector control. That is, there is a difference between the torque generated in the rotor corresponding to the current supplied immediately before the control of the motor is switched and the torque generated in the rotor corresponding to the current supplied immediately after the control of the motor is switched. Means that is occurring. As a result, the rotation speed ω fluctuates when the motor control is switched from the constant current control to the vector control.

FIG. 12B is a diagram showing the current values iα and iβ of the rest coordinate system and the rotation speed ω based on the signal of the rotary encoder in the state where the present embodiment is applied. As shown in FIG. 12B, when the present embodiment is applied, the initial value of the integration control unit 502b is set based on the current value iq', so that immediately after switching from the constant current control to the vector control. Even if there is, the current waveform changes in a sinusoidal shape. This means that torque is generated in the rotor immediately after switching from constant current control to vector control. Further, in the present embodiment, since the initial value of the integral control unit 502b is set based on the current value iq'detected during the constant current control, the torque applied to the rotor during the constant current control and the control switching are performed. The difference between the torque applied to the rotor immediately afterwards can be made as small as possible. That is, there is a difference between the torque generated in the rotor corresponding to the current supplied immediately before the control of the motor is switched and the torque generated in the rotor corresponding to the current supplied immediately after the control of the motor is switched. Can be suppressed from occurring. As a result, it is possible to prevent the rotation speed of the motor from fluctuating when the control of the motor is switched from the constant current control to the vector control.

FIG. 12C is a diagram showing driving electric summer Vd and Vq of the rotating coordinate system in the state where the present embodiment is applied. In the present embodiment, the current controllers 503 and 504 generate drive voltages Vq and Vd based on the deviation of the current values in the rotating coordinate system in both the constant current control and the vector control control methods. That is, the integral control unit in the current controller was used in both constant current control and vector control. Therefore, the control of the integral control unit in the current controller immediately after the motor control is switched from the constant current control to the vector control is based on the integral value of the integral control unit immediately before the motor control is switched from the constant current control to the vector control. Is done. With such a configuration, as shown in FIG. 12 (c), it is possible to suppress abrupt fluctuations in the values of the drive voltages Vq and Vd at the time of control switching. As a result, it is possible to prevent the control of the motor from becoming unstable.

[Second Embodiment]
Since the configuration of the image forming apparatus is the same as that of the first embodiment, the description thereof will be omitted.

In the first embodiment, it is based on the deviation between the current command value iq_ref and the current value iq in the rotating coordinate system based on the command phase θ_ref and the deviation between the current command value id_ref and the current value id in the rotating coordinate system. Constant current control was performed. In the present embodiment, the constant current control is performed based on the deviation between the current command value iα_ref and the current value iα in the stationary coordinate system and the deviation between the current command value iβ_ref and the current value iβ in the stationary coordinate system. explain. In the following description, the description of the portion where the configuration of the vector control and the configuration of switching the control of the motor are the same as those of the first embodiment will be omitted.

<Constant current control>
FIG. 13 is a block diagram showing an example of the configuration of the motor control device 157 according to the present embodiment. The motor control device 157 is composed of at least one ASIC, and executes each function described below.

The CPU 151a outputs the command phase θ_ref to the constant current controller 700. The constant current controller 700 generates and outputs the command values iα_ref and iβ_ref of the current in the rest coordinate system corresponding to the command phase θ_ref output from the CPU 151a.

Next, the current detectors 507 and 508 detect the drive current flowing through the A-phase and B-phase windings of the motor 509. After that, the detected drive current is converted from an analog value to a digital value by the A / D converter 510, and is expressed as current values iα and iβ as in equations (1) and (2).

The command value iα_ref output from the constant current controller 700 is input to the subtractor 102 via the changeover switch 516a. Further, the current value iα output from the A / D converter 510 is input to the subtractor 102 via the changeover switch 516c. The subtractor 102 outputs the deviation between the current value iα and the command value iα_ref to the current controller 503.

Further, the command value iβ_ref output from the constant current controller 700 is input to the subtractor 103 via the changeover switch 516a. Further, the current value iβ output from the A / D converter 510 is input to the subtractor 103 via the changeover switch 516c. The subtractor 103 outputs the deviation between the current value iβ and the command value iβ_ref to the current controller 504.

FIG. 14 is a block diagram showing the configuration of the current controller 503 in the present embodiment. The configuration of the current controller 503 shown in FIG. 14 is an example in this embodiment, and the configuration of the current controller 503 is not limited to this.

As shown in FIG. 14, the current controller 503 has a proportional control unit 503aq that performs P control when vector control is performed, an integral control unit 503bq that performs I control, and a differential control unit 503cq that performs D control. Further, the current controller 503 has an adder 503dq that adds signals output from the proportional control unit 502aq, the integration control unit 502bq, and the differential control unit 502cq. Further, the current controller 503 generates a drive voltage Vq as the value of the q-axis component of the voltage vector corresponding to the drive voltage to be applied to the winding of the motor 509 based on the signal output from the adder 503dq. It has a generation unit 503eq. Since the processing method of each configuration is the same as the method described in the first embodiment, the description thereof will be omitted.

Further, the current controller 503 includes a proportional control unit 503aα that performs P control when constant current control is performed, an integral control unit 503bα that performs I control, and a differential control unit 503cα that performs D control. Further, the current controller 503 has an adder 503dα that adds signals output from the proportional control unit 502aα, the integration control unit 502bα, and the differential control unit 502cα. Further, the current controller 503 generates a drive voltage Vα as the value of the α-axis component of the voltage vector corresponding to the drive voltage to be applied to the winding of the motor 509 based on the signal output from the adder 503dα. It has a generation unit 503eα.

In the constant current control, the current controller 503 generates a drive voltage Vα based on the PID control so that the deviation input via the changeover switch 516d becomes small. Specifically, the current controller 503 generates and outputs a drive voltage Vα so that the input deviation becomes zero.

More specifically, the proportional control unit 503aα outputs a value proportional to the deviation so that the deviation output from the subtractor 102 becomes 0. Further, the integration control unit 503bα outputs a value proportional to the time integration of the deviation so that the deviation output from the subtractor 102 becomes zero. Further, the differential control unit 503cα outputs a value proportional to the time change of the deviation so that the deviation output from the subtractor 102 becomes zero.

Then, the adder 503dα adds the values output from the proportional control unit 503aα, the integral control unit 503bα, and the differential control unit 503cα, and the added values are output to the drive voltage generation unit 503eα. The drive voltage generation unit 503eα generates and outputs a drive voltage Vα based on the value output from the adder 503dα. Specifically, for example, the drive voltage Vα is generated by multiplying the value output from the adder 503d by a preset proportional coefficient, and is output via the changeover switch 516e.

In this way, the current controller 503 functions as a generation means for generating the drive voltage. The current controller 504 has the same configuration as the current controller 503, and the β-axis component of the voltage vector corresponding to the drive voltage to be applied to the winding of the motor 509 by the same method as the current controller 503. The drive voltage Vβ is generated as the value of.

The current controllers 503 and 504 in the present embodiment generate drive voltages Vα and Vβ based on PID control, but the present invention is not limited to this. For example, the current controllers 503 and 504 may generate drive voltages Vα and Vβ based on PI control.

The drive voltages Vα and Vβ output from the current controllers 503 and 504 are input to the PWM inverter 506 via the switch 516b, and the PWM inverter 506 winds each phase of the motor 509 in the same manner as in the first embodiment. A drive current is supplied to the wire to drive the motor 509.

As described above, in the constant current control, neither the phase feedback control nor the velocity feedback control is performed. That is, in the constant current control, the drive current supplied to the winding is not adjusted according to the rotation state of the rotor. Therefore, in the constant current control, a current obtained by adding a predetermined margin to the current required to rotate the rotor is supplied to the winding so that the motor does not step out. Specifically, the command values iα_ref and iβ_ref of the current in the stationary coordinate system include a current value required for rotating the rotor and a current value corresponding to a predetermined margin.

The above is the description of the constant current control in this embodiment. In the present embodiment, as shown in FIG. 13, when the control of the motor is switched, the control switch 515 outputs a change signal to the changeover switches 516a to 516e.

<Processing of current controller when switching control>
When the current controllers 503 and 504 have a PID control configuration (for a stationary coordinate system) used in constant current control and a PID control configuration (for a rotating coordinate system) used in vector control, they are as follows. Problems can occur. Specifically, when the motor control is switched from the constant current control to the vector control, the initial values of the drive voltages Vq and Vd immediately after the motor control is switched from the constant current control to the vector control are not appropriate (at 0). It may be generated based on the integral control part of the state. If the initial value is not appropriate, the drive voltage Vq and Vd may fluctuate rapidly (discontinuously change) when the motor control is switched from constant current control to vector control, and the motor control may become unstable. There is sex.

Therefore, in the present embodiment, by applying the following configuration, it is possible to prevent the drive voltages Vq and Vd from suddenly fluctuating when the motor control is switched from the constant current control to the vector control. Specifically, as shown in FIG. 14, when the motor control is switched from the constant current control to the vector control, the CPU 151a determines the integral control unit based on the integral value of the integral control unit 503bα immediately before the switching signal is input. Set the initial value of the integral of 503bq.

By using such a configuration, the values of the drive voltages Vq and Vd can be gradually changed at the time of control switching without suddenly changing. As a result, it is possible to prevent the control of the motor from becoming unstable.

In the present embodiment, the CPU 151a sets (determines) the initial value of the integration of the integration control unit 503bq based on the integration value of the integration control unit 503bα immediately before the switching signal is input, but this is not the case. For example, when the control of the motor is switched from the constant current control to the vector control, the CPU 151a may set the initial value of the integration of the integration control unit 503bq to a predetermined value.

Further, in the present embodiment, the CPU 151a sets (determines) the initial value of the integration of the integration control unit 503bq based on the integration value of the integration control unit 503bα immediately before the switching signal is switched, but this is not the case. For example, the CPU 151a may set the initial value of the integration in the integration control unit 502b based on the integration value of the integration control unit 503bα two times before, not immediately before.

Further, the motor control device in the present embodiment has a part (current controllers 503, 504, PWM inverter 506, etc.) that is partially shared between the circuit that performs vector control and the circuit that performs constant current control. This is not the case. For example, a circuit that performs vector control and a circuit that performs constant current control may be provided independently.

In the constant current control in the first embodiment and the second embodiment, a current obtained by adding a predetermined margin to the current required to rotate the rotor is supplied to the winding so that the motor does not step out. NS. Therefore, the current value iq'also includes a value corresponding to the margin.

In the first embodiment and the second embodiment, the phase controller 502 sets a value obtained by multiplying the current value iq'by the proportional coefficient Kq as the initial value of integration in the integration control unit 502b, but this is not the case. For example, the phase controller 502 may set a value obtained by subtracting a predetermined value from the current value iq'as the initial value of integration in the integration control unit 502b. Further, the current value iq'may be set as it is as the initial value of integration in the integration control unit 502b.

Further, in the first embodiment and the second embodiment, the initial value of integration in the integration control unit 502b is set based on the current value iq'acquired immediately before the switching signal is switched, but this is not the case. For example, the initial value of integration in the integration control unit 502b may be set based on the current value iq'two times before, not immediately before.

Further, in the first embodiment and the second embodiment, the phase controller 502 sets a value corresponding to the current value iq'as the initial value of integration in the integration control unit 502b, and PID control based on the set initial value. The q-axis current command value iq_ref was generated by, but this is not the case. For example, the phase controller 502 may use the current value iq'as it is as the q-axis current command value iq_ref. When the current value iq'is output as the q-axis current command value iq_ref as it is, the initial value of integration in the integration control unit 502b is 0, so the integration is performed with the initial value being 0 at the next PID control. Control is done. Therefore, in this case, the current value iq'is output as the q-axis current command value iq_ref as it is until the feedback of the rotation phase θ (integral control by the q-axis current command value iq_ref based on the current value iq') is performed at least once. Will be done. That is, the q-axis current command value iq_ref based on PID control is generated by the feedback of the rotation phase θ from the second time onward. Further, even if the current value iq'is output as the q-axis current command value iq_ref as it is, the configuration as described above is provided as long as the current value iq'is set as the initial value of the integration control unit. Is not needed. That is, it is not necessary to have a configuration in which the current value iq'is output as the q-axis current command value iq_ref as it is until the feedback of the rotation phase θ is performed at least once.

Further, in the first embodiment and the second embodiment, the phase controller 502 is in operation during the constant current control, but the phase controller 502 may be stopped. Specifically, for example, among the configurations provided in the phase controller 502, the configuration may be such that the integration control unit is operating.

Further, in the first embodiment and the second embodiment, the circuit that controls the drive of the motor 509 by using the vector controller 518 corresponds to the first control circuit in the present invention. Further, in the first embodiment and the second embodiment, the circuit that controls the drive of the motor 509 by using the constant current controller 517 corresponds to the second control circuit in the present invention.

Further, in the vector control in the first embodiment and the second embodiment, the motor 509 is controlled by performing the phase feedback control, but the present invention is not limited to this. For example, the motor 509 may be controlled by feeding back the rotation speed ω of the rotor 402. Specifically, as shown in FIG. 15, a speed determinant 514 is provided inside the motor control device, and the speed determinant 514 determines the rotation speed ω based on the time change of the rotation phase θ output from the phase determinant 513. decide. The following equation (10) shall be used to determine the speed.
ω = dθ / dt (10)

Then, the CPU 151a outputs a command speed ω_ref indicating the target speed of the rotor. Further, a speed controller 500 is provided inside the motor control device, and the speed controller 500 generates and outputs a q-axis current command value iq_ref so that the deviation between the rotation speed ω and the command speed ω_ref becomes small. .. The motor 509 may be controlled by performing such speed feedback control. In such a configuration, since the rotation speed is fed back, the rotation speed of the rotor can be controlled to be a predetermined speed. Therefore, in the image forming apparatus, speed feedback control is applied to a motor that drives a load (for example, a photosensitive drum, a transport belt, etc.) in which the rotation speed needs to be controlled to a constant speed in order to properly form an image on a recording medium. Apply the vector control used. As a result, the image can be appropriately formed on the recording medium. In this case, the command speed ω_ref is also used when performing constant current control. Further, the control switching may be performed based on the command speed ω_ref, or may be performed based on the rotation speed ω determined by the speed determinant 514.

Further, in the first embodiment and the second embodiment, the rotation speed ω_ref'is determined based on the amount of change in the command phase θ_ref in a predetermined period, but the present invention is not limited to this. For example, the rotation speed ω_ref'based on the period in which the magnitude of the periodic signal correlated with the rotation period of the rotor 402 becomes 0, such as the drive current iα or iβ, the drive voltage Vα or Vβ, and the induced voltage Eα or Eβ. May be determined.

Further, in the first embodiment and the second embodiment, the stepping motor is used as the motor for driving the load, but other motors such as a DC motor may be used. Further, the present embodiment can be applied not only to the case where the motor is a two-phase motor but also to another motor such as a three-phase motor.

Further, in the first embodiment and the second embodiment, a permanent magnet is used as the rotor, but the present invention is not limited to this.

157 Motor controller 402 Rotor 502 Phase controller 503,504 Current controller 506 PWM inverter 507,508 Current detector 509 Motor 513 Phase controller 700 Constant current controller 701 Vector controller

Claims (8)

In a motor control device that controls the motor based on a command phase representing the target phase of the rotor of the motor.
A detection means for detecting the drive current flowing through the winding of the motor, and
A phase determining means for determining the rotational phase of the rotor based on the drive current detected by the detecting means, and
A conversion means for converting the current value of the stationary coordinate system detected by the detection means into a current value of the rotating coordinate system based on the rotation phase determined by the phase determining means.
The value of the torque current component as a converted current value of the rotating coordinate system by transformation means a current value for generating a torque on the rotor, the command phase and the rotational phase determined by the phase determining means the first to be the target value deviation between is set to be smaller, a first control mode for controlling the driving current supplied to the winding, the driving current detected by said detecting means A control means including a second control mode for controlling the drive current supplied to the winding so that the current value in the stationary coordinate system becomes a predetermined second target value.
The control means has
In a period during which the first control mode is executed, the first of the so that the first deviation which is a deviation between the value of the torque current component of the detected driving current by the target value and the detection means is reduced A first generation means that performs a first integral control based on the first deviation and generates a drive voltage to be applied to the winding based on the first integral control .
During the period in which the second control mode is executed, the second deviation, which is the deviation between the second target value and the current value of the stationary coordinate system of the drive current detected by the detection means, becomes smaller. A second generation means that performs a second integral control based on the second deviation and generates a drive voltage to be applied to the winding based on the second integral control.
A supply means for supplying a drive current to the winding based on a drive voltage to be supplied to the winding.
With
When the control mode for controlling the drive current is switched from the second control mode to the first control mode, the initial value of the first integral control is the second integral during execution of the second control mode. A motor control device characterized in that it is set based on an integral value of control. In a motor control device that controls the motor based on a command speed representing a target speed of the rotor of the motor.
A detection means for detecting the drive current flowing through the winding of the motor, and
A speed determining means for determining the rotational speed of the rotor based on the drive current detected by the detecting means, and
A phase determining means for determining the rotational phase of the rotor and
A conversion means for converting the current value of the stationary coordinate system detected by the detection means into a current value of the rotating coordinate system based on the rotation phase determined by the phase determining means.
Rotational speed value of the torque current component, which is determined by the command speed and the speed determining means as a current value for generating a torque to the rotor a current value of the rotating coordinate system converted by said converting means the first to be the target value deviation between is set to be smaller, a first control mode for controlling the driving current supplied to the winding, the driving current detected by said detecting means A control means including a second control mode for controlling the drive current supplied to the winding so that the current value in the stationary coordinate system becomes a predetermined second target value.
Have,
The control means
In a period during which the first control mode is executed, the first of the so that the first deviation which is a deviation between the value of the torque current component of the detected driving current by the target value and the detection means is reduced A first generation means that performs a first integral control based on the first deviation and generates a drive voltage to be applied to the winding based on the first integral control .
During the period in which the second control mode is executed, the second deviation, which is the deviation between the second target value and the current value of the stationary coordinate system of the drive current detected by the detection means, becomes smaller. A second generation means that performs a second integral control based on the second deviation and generates a drive voltage to be applied to the winding based on the second integral control.
A supply means for supplying a drive current to the winding based on a drive voltage to be supplied to the winding.
With
When the control mode for controlling the drive current is switched from the second control mode to the first control mode, the initial value of the first integral control is the second integral during execution of the second control mode. A motor control device characterized in that it is set based on an integral value of control. When the value corresponding to the rotation speed of the rotor becomes a value larger than a predetermined value during the execution of the second control mode, the control means controls the control mode from the second control mode to the first control. The motor control device according to claim 1 or 2, wherein the mode is switched. The motor control device according to claim 3 , wherein the value corresponding to the rotation speed of the rotor is a value indicating the target speed of the rotor. The initial value of the first integral control when the control mode is switched from the second control mode to the first control mode is immediately before the control mode is switched from the second control mode to the first control mode. The motor control device according to any one of claims 1 to 4, wherein the motor control device is set based on the integral value in the second integral control. A transport roller that transports the sheet and
The motor that drives the transfer roller and
The motor control device according to any one of claims 1 to 5.
Have,
The motor control device is a sheet transfer device characterized in that it controls the drive of a motor that drives the transfer roller. The loading section on which the recording medium is loaded and
A feeding unit that feeds the recording medium loaded on the loading unit, and
An image forming means for forming an image on a recording medium fed by the feeding unit, and
The motor that drives the load and
The motor control device according to any one of claims 1 to 5.
Have,
The motor control device is an image forming device characterized by controlling a motor that drives the load. The image forming apparatus according to claim 7 , wherein the load is a transfer roller that conveys the recording medium.

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