JP6899720B2 - Linear motor system and compressor with it - Google Patents

Description

The present invention relates to a linear motor system and a compressor having the same.

A linear motor is known that drives a mover to which an elastic body is connected at a mechanical resonance frequency in the elastic body and the mover system.
For example, Patent Document 1 discloses a motor drive device that controls a linear vibration motor, the motor drive device has a motor driver, and the motor driver performs arithmetic processing for calculating a thrust constant. Described is a configuration having a control unit that switches between a thrust constant calculation mode (non-operation mode) in which a DC voltage is applied to the motor and an operation mode in which an AC voltage is applied to the linear vibration motor so that the linear vibration motor is normally operated. Has been done.
Further, in Patent Document 1, the motor drive device includes a thrust detection unit for obtaining a thrust detection value when a DC voltage is applied, which is used for calculating a thrust constant, and the mover or the mover can detect the thrust of the mover. It is disclosed that it is performed by a sensor such as a pressure sensor or a strain gauge attached to a contact portion.

Japanese Unexamined Patent Publication No. 2004-274997

However, in Patent Document 1, the wiring is configured because the thrust required for calculating the thrust constant of the linear vibration motor is detected by a sensor such as a pressure sensor or a strain gauge provided in the linear vibration motor. Not only is it complicated, but it can also lead to an increase in the size of the device. Further, no consideration is given to the point that the reactive current is suppressed by applying a voltage corresponding to the induced voltage of the linear vibration motor and the linear motor is driven with high efficiency.
Therefore, the present invention has a linear motor system capable of suppressing an invalid current flowing in a winding and driving the linear motor with high efficiency without arranging a sensor for detecting the thrust of the linear motor in the linear motor. Provide a compressor.

In order to solve the above problems, at least an armor having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an elastic body connected to the armature or the field magnet are provided. and, that the relative position of the armature and the field element according to the deformation of the elastic body is varied, the linear motor for relatively reciprocating the field element and the armature in the axial direction, the A control unit that applies a voltage to the winding is provided, and the control unit is based on the rate of change of the magnetic flux generated by the fluctuation of the relative positions of the armature and the field of the voltage applied to the winding. , The absolute value of the applied voltage component is reduced at the relative position where the rate of change of the magnetic flux is small, and when the relative position of the armature and the field magnet exceeds a predetermined value, the voltage applied to the winding Among them, the absolute value of the voltage component to be applied is reduced according to the rate of change of the magnetic flux, and the control unit is at least the position of the field magnet detected by the position detector, the reference phase, and the frequency command value. A voltage command value maker for generating a first voltage command value based on the above is provided, and the voltage command value maker has a characteristic curve showing the relationship between the position of the field and the gain in advance, and the position detector It has an end-induced voltage regulator that outputs a gain according to the position of the field detected by the above as an end-induced voltage adjustment gain, and the voltage command value creator obtains the end-induced voltage adjustment gain. It is characterized in that it is multiplied by the first voltage command value and output to the linear motor.
Further, the linear motor system according to the present invention has at least an armature having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an armature connected to the armature or the field magnet. and a body, that the relative position of the armature and the field element varies according to the deformation of the elastic body, a linear for relatively reciprocating the armature and the field element in the axial direction The control unit includes a motor and a control unit that applies a voltage to the winding, and the control unit is a harmonic of the voltage applied to the winding in response to an increase in the relative position between the field magnet and the armature. It is characterized by increasing the component content.
Further, the compressor according to the present invention has at least an armature having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an elastic body connected to the armature or the field magnet. has the door, that the relative position of the armature and the field element varies according to the deformation of the elastic body, a linear motor for relatively reciprocating the field element and the armature in the axial direction A compressor that includes a control unit that applies a voltage to the winding and compresses the working fluid by the piston connected to the field reciprocating in the cylinder. The control unit is Of the voltage applied to the winding, the absolute value of the applied voltage component is set relative to the relative position where the rate of change of the field is small, based on the rate of change of the magnetic flux generated by the fluctuation of the relative positions of the armature and the field. When the relative position between the armature and the field magnet exceeds a predetermined value, the absolute value of the voltage component to be applied according to the rate of change of the magnetic flux among the voltages applied to the winding is changed. the small and, being et al., the control unit, based on the current detection value flowing through the windings is detected by at least the current detector, a position estimator for estimating the position of the field element, by the position estimator The first voltage command value is generated based on the position estimated value of the field magnet, the reference phase and the frequency command value, and corresponds to the load current obtained by the current detection value and the voltage drop based on the reference phase. A voltage command value maker for generating a second voltage command value is provided, and the voltage command value maker has a characteristic curve showing the relationship between the position of the field and the gain in advance, and the position estimator is used. The voltage command value maker has an end-induced voltage regulator that outputs a gain corresponding to the estimated value of the position of the field magnet as an end-induced voltage adjustment gain, and the voltage command value maker outputs the end-induced voltage adjustment gain to the first. Multiply the voltage command value of and add it to the second voltage command value, or multiply the result of adding the first voltage command value and the second voltage command value by the end-induced voltage adjustment gain. , It is characterized by outputting to the linear motor.

According to the present invention, there is a linear motor system capable of suppressing an invalid current flowing in a winding and driving the linear motor with high efficiency without arranging a sensor for detecting the thrust of the linear motor in the linear motor. It becomes possible to provide a compressor.
Issues, configurations and effects other than those described above will be clarified by the description of the following embodiments.

It is an overall schematic block diagram of the linear motor system of Example 1 which concerns on one Example of this invention. It is a perspective view of the structural example of an armature. It is a figure which shows the example of the change of the armature interlinkage magnetic flux with respect to the position of a mover (field magnet). It is a schematic diagram which shows the vertical cross section of a magnetic pole and the flow of a magnetic flux. It is explanatory drawing of the polarity generated in the magnetic pole tooth. It is explanatory drawing of the external mechanism connected to a mover. It is explanatory drawing of the relationship between a drive frequency and a stroke. It is explanatory drawing of the phase relationship between a mover position and a mover speed, and the phase relationship between an applied voltage and a motor current. It is a vector diagram of applied voltage and current. It is explanatory drawing which shows the structural example of the phase difference detector which constitutes the control part shown in FIG. It is explanatory drawing of the relationship between a drive frequency and a phase difference detector output. It is explanatory drawing which shows the structural example of the drive frequency adjuster which comprises the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage command value creator which comprises the control part shown in FIG. It is explanatory drawing which shows the 1st structural example of the end-induced voltage regulator which constitutes the voltage command value creator shown in FIG. It is explanatory drawing which shows the 2nd structural example of the end-induced voltage regulator which constitutes the voltage command value creator shown in FIG. It is explanatory drawing which shows the 3rd structural example of the end-induced voltage regulator which constitutes the voltage command value creator shown in FIG. It is explanatory drawing of the applied voltage waveform. It is explanatory drawing which shows the structural example of the stroke controller which constitutes the voltage command value creator shown in FIG. It is a figure which shows the structural example of the power conversion circuit which comprises the linear motor drive apparatus shown in FIG. It is a vertical sectional view of the closed type compressor of Example 2 which concerns on another Example of this invention. It is an overall schematic block diagram of the linear motor system of Example 2. FIG. It is explanatory drawing which shows the structural example of the phase difference detector which constitutes the control part shown in FIG. It is explanatory drawing which shows the structural example of the load current detector which constitutes the control part shown in FIG. It is explanatory drawing which shows the other configuration example of the load current detector which constitutes the control part shown in FIG. It is explanatory drawing which shows the structural example of the voltage command value maker which constitutes the control part shown in FIG. It is a vector figure which shows the vector sum in the voltage command value creator at the time of a light load and the time of a heavy load. It is explanatory drawing which shows the other configuration example of the voltage command value creator which constitutes the control part shown in FIG. It is explanatory drawing which shows the structural example of the position estimator constituting the control part shown in FIG. It is a figure which shows the structural example of the power conversion circuit which comprises the linear motor drive apparatus shown in FIG. It is explanatory drawing which shows the structural example of the verification system of Example 3 which concerns on another Example of this invention. It is a circuit diagram of the air suspension system of Example 4 which concerns on another Example of this invention. It is the schematic of the vehicle equipped with the air suspension system shown in FIG. 31. It is explanatory drawing which shows the structural example of the end-induced voltage regulator which comprises the voltage command value creator of Example 4.

Hereinafter, examples of the present invention will be described in detail with reference to the accompanying drawings. Similar components are designated by the same reference numerals, and duplicate description will be omitted.
The various components of the present invention do not necessarily have to be independent of each other, and a plurality of components are formed as one member, and one component is formed of a plurality of members. It is allowed that one component is a part of another component, that a part of one component overlaps with a part of another component, and so on.
Hereinafter, examples of the present invention will be described with reference to the drawings.

In this embodiment, for convenience of explanation, the terms front-back direction, left-right direction, and up-down direction are used, but the gravity directions do not necessarily have to be parallel to the downward direction, and the front-back direction, left-right direction, and up-down direction are used. It can be parallel to the direction or any other direction.

<Linear motor drive device 101>
FIG. 1 is an overall schematic configuration diagram of the linear motor system 100 of the first embodiment according to the first embodiment of the present invention. The linear motor system 100 includes a linear motor drive device 101 and a linear motor 104. As will be described later, the linear motor 104 has an armature 9 and a mover 6 that move relative to each other.
The linear motor drive device 101 includes a position detector 106, a control unit 102, and a power conversion circuit 105.
The position detector 106 detects the relative position (movable element position) of the mover 6 with respect to the armature 9. In this embodiment, the mover (field magnet) 6 moves in the vertical direction, but the armature 9 and the mover (field magnet) 6 need only move relative to each other, and the armature 9 moves in the vertical direction. But it's okay. Further, the armature 9 and the mover (field magnet) 6 may reciprocate along the vertical direction (axial direction) at different speeds. In any case, it is preferable to connect one end of the resonance spring (assist spring) 23 to an object that moves along the vertical direction (axial direction). In the following, a case where the mover (field magnet) 6 reciprocates in the vertical direction will be described as an example, but the direction of the reciprocating motion is not limited to the vertical direction. For example, the mover (field magnet) 6 may be configured to reciprocate in the horizontal direction, or the mover (field magnet) 6 may reciprocate in a direction having an arbitrary angle with respect to the vertical direction. May be. The same applies to the armature 9.
The control unit 102 outputs an output voltage command value to the power conversion circuit 105 or a drive signal (pulse signal) for driving the power conversion circuit 105, depending on the detection result of the position detector 106. Although details will be described later, the power conversion circuit 105 is an example of a power conversion unit that converts the voltage of the DC voltage source 120 (FIG. 19) and outputs an AC voltage. A DC current source may be used instead of the DC voltage source 120.

<Linear motor 104>
FIG. 2 is a perspective view of the linear motor 104 (a perspective view of an armature configuration example). The linear motor 104 of this embodiment has a mover 6 that can move relative to the armature 9 in the direction (front-back direction) in which the permanent magnets 2 (2a, 2b) are arranged. The armature 9 has two magnetic poles 7 facing each other in the vertical direction via a gap, and a winding 8 wound around the magnetic pole 7. The mover 6 is arranged in this gap. The magnetic pole 7 has magnetic pole teeth 70 (also referred to as teeth) as end faces facing the mover 6.
The armature 9 can apply a force in the front-rear direction (hereinafter, referred to as a thrust) to the mover 6. For example, as will be described later, the thrust can be controlled so that the mover 6 reciprocates in the front-rear direction.
The mover 6 has two flat plate-shaped permanent magnets 2 (2a, 2b) magnetized in the vertical direction. The rear permanent magnet 2a and the front permanent magnet 2b are magnetized in opposite directions. In this embodiment, the rear permanent magnet 2a has an N pole on the upper side, and the front permanent magnet 2b has an S pole on the upper side. In FIG. 2, the permanent magnets 2a and 2b are shown, but the mover 6 is not shown. As the mover 6, for example, a flat plate-shaped one to which a flat plate-shaped permanent magnet 2 is fixed can be adopted.
The control unit 102 outputs a drive signal so that the mover 6 is reciprocated within a range in which the permanent magnets 2a and 2b face the armature 9.

FIG. 3 is a diagram showing an example of a change in the armature interlinkage magnetic flux with respect to the position of the mover (field magnet). The horizontal axis is the mover (field magnet) position (m), the vertical axis is the magnetic flux (Wb), and the magnetic flux (Wb) is the interlinkage magnetic flux when no current is flowing through the winding, and the magnetic flux due to the permanent magnet 2. Shows a change in.
A voltage corresponding to the time derivative of the magnetic flux is induced in the winding 8, and this voltage is generally referred to as an induced voltage, a counter electromotive voltage, a velocity electromotive voltage, or the like. Even with the same motor, it changes according to the speed of the mover. For example, when the speed of the mover is high, the induced voltage becomes high.

Assuming that the slope of the change in magnetic flux (rate of change in magnetic flux) with respect to the position of the mover (field magnet) is the induced voltage constant Ke, the induced voltage constant Ke can be regarded as constant in the section (1) shown in FIG. In this case, the induced voltage can be calculated by multiplying the induced voltage constant Ke by the velocity of the mover.
On the other hand, in the section (2) shown in FIG. 3, since the slope of the change in the magnetic flux (rate of change in the magnetic flux) is not constant, the induced voltage constant Ke has a characteristic that depends on the position of the mover. That is, in the linear motor having the characteristics as shown in FIG. 3, there is a problem that it becomes difficult to calculate the induced voltage when the mover (field element) position reciprocates away from the center.

In the case of a free piston structure in which the mover is not mechanically constrained, the maximum or minimum position of the mover position is not always constant, so even if the instantaneous speed of the mover is the same, the induced voltage changes depending on the position of the mover. There is a problem. In addition, when a mechanism for converting a rotary motion into a linear motion is provided, such a problem does not occur.
The current flowing through the linear motor is determined by the relationship between the voltage applied to the linear motor and the induced voltage. Therefore, if the induced voltage cannot be calculated correctly, the current increases unnecessarily due to the excessive applied voltage, and the efficiency of the linear motor decreases. This causes a problem that the efficiency of the circuit for driving the linear motor is also lowered. When the efficiency of the drive circuit decreases, the amount of heat generated increases accordingly, so that it is necessary to increase the area of the heat radiation fins, which increases the size of the system.

FIG. 4 is a cross-sectional view taken along the line AA'of FIG. 2 in a plane (A-A'cross-sectional arrow view). As shown in FIG. 3, the magnetic pole 7 and the yoke 7e are integrally formed of a magnetic material such as iron to form a magnetic circuit. The arrow line in FIG. 4 shows an example of a magnetic flux line when a current is passed through the two windings 8. The direction of the magnetic flux flow is not limited to the one shown in the figure because it can be in the opposite direction depending on the direction of the current flowing through the winding 8. The magnetic flux lines magnetize the magnetic pole teeth 70.

[Thrust given to the mover 6]
FIG. 5 is a diagram for explaining the thrust received by the mover 6 due to the magnetization of the magnetic pole teeth 70. The polarities of the magnetic pole teeth 70 generated by the current flowing through the winding 8 are represented by "N" and "S" attached in the vicinity of the magnetic pole teeth 70 in the drawing. Further, in FIG. 5, the white arrow indicates the direction of the current flowing through the winding 8. In the left figure of FIG. 5, the upper magnetic pole tooth 70a is magnetized to "S" and the lower magnetic pole tooth 70b is magnetized to "N" by the current flowing through the winding 8, so that the mover 6 exerts a force in the forward direction. An example is shown in which the mover 6 has moved forward. In the right figure of FIG. 5, the upper magnetic pole tooth 70a is magnetized to "N" and the lower magnetic pole tooth 70b is magnetized to "S" by the current flowing through the winding 8, so that the mover 6 exerts a force in the backward direction. An example is shown in which the mover 6 is moved backward.

By applying a voltage or current to the winding 8 in this way, magnetic flux can be supplied to the magnetic circuit including the two magnetic poles 7 to magnetize the two opposing magnetic pole teeth 70 (magnetic pole tooth assembly). By applying an alternating voltage or current such as a sine wave or a square wave (square wave) as a voltage or current, a thrust for reciprocating the mover 6 can be given. Thereby, the movement of the mover 6 can be controlled.

The thrust applied to the mover 6 can be changed by changing the amplitude of the applied AC current or AC voltage. Further, the displacement of the mover 6 can be changed as desired by appropriately changing the thrust applied to the mover 6 by using a known method. Here, when the mover 6 reciprocates (for example, the movement generated in the mover 6 by sequentially repeating the magnetization of the magnetic pole teeth 70 as shown in the left and right figures of FIG. 5), it changes in an AC waveform. The amount of change in the displacement of the mover 6 is called a stroke.
Since the magnetic pole teeth 70 are magnetic materials, a magnetic attraction force that attracts the permanent magnet 2 acts on them. In this embodiment, since the two magnetic pole teeth 70 are arranged so as to sandwich the mover 6 with the magnetic attraction teeth 70 facing each other, the resultant force of the magnetic attraction force acting on the mover 6 can be reduced.

[Movement 6 external mechanism]
FIG. 6 is an explanatory view of an external mechanism connected to the mover 6. For example, an external mechanism composed of a resonance spring 23 (assist spring) which is a coil spring is connected to one end of the mover 6, and the spring force thereof. It is a figure explaining the mechanism that the mover 6 is returned by this. One end of the resonance spring 23 is connected to the mover 6 via the intermediate portion 24, and the other end is fixed to the base portion 25. Further, a side portion 26 that extends substantially parallel to the extending direction of the resonance spring 23 and guides or supports the resonance spring 23 is provided. When the linear motor 104 is reciprocated, acceleration and deceleration are repeated every time the movement direction of the mover 6 changes. During deceleration, the velocity energy of the mover 6 is converted into electrical energy (regenerative operation), but a loss occurs due to the resistance of the wiring to the linear motor 104. On the other hand, as shown in FIG. 6, when a resonance spring 23 (assist spring) is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, the mover 6 is movable. The speed energy of the child 6 can be effectively used, and a highly efficient linear motor drive system can be configured. Instead of the resonance spring 23, for example, a leaf spring or an elastic body such as rubber which has an appropriate Young's modulus and expands and contracts in the same manner as when a coil spring is used may be used. With this configuration, the mover (field magnet) 6 is configured as a movable element (field magnet) that moves in the vertical direction, but instead of the mover 6, an elastic body is connected to the armature 9. The armature 9 may be configured as an armature moving type that moves the armature 9 in the vertical direction.

FIG. 7 is an explanatory diagram of the relationship between the drive frequency and the stroke, in which the drive frequency of the AC voltage is on the horizontal axis and the stroke of the mover 6 is on the vertical axis, and these relationships are shown. The amplitude of the AC voltage at each drive frequency is the same. As can be seen from FIG. 7, the stroke of the mover 6 sharply increases near the resonance frequency, and the stroke decreases as the distance from the resonance frequency increases. The resonance frequency is given by the square root of the value obtained by dividing the spring constant k of the resonance spring 23 by the mass m of the mover 6, but this value is an approximate value depending on the system of the linear motor 104.
In this way, by applying an AC voltage having a resonance frequency or a drive frequency in the vicinity thereof, it is possible to vibrate with a large stroke (large energy). That is, when controlling the linear motor 104 in which an elastic body such as a resonance spring 23 is added to the mover 6, it is important to detect or estimate the resonance frequency of the mover 6. It is important to detect or estimate the resonance frequency of the mover 6 even when the stroke of the mover 6 is controlled as desired.

[Phase relationship during driving]
FIG. 8 is an explanatory diagram of the phase relationship between the position of the mover and the speed of the mover, and the phase relationship between the applied voltage and the motor current. When the linear motor 104 is driven, the upper figure of FIG. 8 shows the relationship between the position and speed of the mover 6 over time, and the lower figure of FIG. 8 shows the relationship between the applied voltage waveform and the time change of the current flowing through the linear motor 104. .. The upper and lower figures of FIG. 8 are waveforms having the same timing. FIG. 9 is a diagram showing the AC waveform of FIG. 8 as a vector. When the mover 6 connected to the external mechanism having the resonance spring 23 is reciprocated, the displacement of the mover 6 changes in a sinusoidal shape. Since the velocity of the mover 6 is the time derivative of the displacement, it changes in a cosine wave shape. Therefore, these can be shown as vectors on the two orthogonal axes. From FIGS. 8 and 9, it can be seen that the speed, applied voltage, and motor current of the mover 6 are substantially in phase.
Further, when the resonance spring 23 is added to the mover 6 and the mover 6 is reciprocated at a mechanical resonance frequency determined by the mass of the mover 6 and the spring constant, the phase of the position of the mover 6 is the winding 8 It is known that the phase difference is 90 degrees with respect to the phases of the voltage Vm applied to, the motor current Im, and the speed of the mover 6. That is, when any of these relationships is established, it can be estimated that the vehicle is driven at the resonance frequency.

Depending on the value of the winding resistance and winding inductance of the linear motor 104, or the load element applied to the mover 6, the phase relationship between the voltage Vm applied to the winding 8 and the motor current Im, and the speed of the mover 6 may change. Since the phase difference is not always 90 degrees, it is desirable to provide a control unit that changes depending on the conditions. At this time, it is particularly desirable to consider the change in the motor current Im due to the fluctuation of the load.
If the mass of the mover 6 deviates from the assumption due to manufacturing variation, or if the mass connected to the resonance spring 23 changes due to the load element added to the mover 6, the resonance frequency changes. Further, when the load element added to the mover 6 has position dependence, the resonance frequency changes during driving. Even in such a case, in order to obtain a desired stroke, it is preferable to detect or estimate the resonance frequency that changes depending on the conditions with high accuracy.
<Control unit 102>
As shown in FIG. 1, the control unit 102 adjusts the drive ^{frequency command value ω *} so that the phase difference dlt θ, which is the output of the phase difference detector 130 and the phase difference detector 130, follows the phase difference command value. frequency regulator 131, an integrator 140, the voltage command value generator 103 outputs a voltage command value Vm ^*, and, by comparing the voltage command value Vm ^* and the triangular wave carrier signal, driving the power conversion circuit 105 which outputs a voltage It is composed of a PWM signal maker 134 that outputs a drive signal to be output.

The control unit 102 inputs the mover (field element) position detection value xm by the position detector 106. Here, as the position detector 106, for example, a laser displacement meter or the like is used. The mover (field) position detection value xm input to the control unit 102 is input to the phase difference detector 130 constituting the control unit 102. ^{The phase difference detector 130 outputs the phase difference dlt θ of the reference phase θ *} , which is a phase command value described in detail later, and the mover (field magnet) position detection value xm. The phase difference dltθ output from the phase difference detector 130 is input to the drive frequency regulator 131. The drive frequency regulator 131 outputs the ^{frequency command value ω *} based on the input phase difference dltθ. The applied voltage Vm based on the frequency command value ω ^* is output to the linear motor 104.

When the mover (field magnet) 6 is reciprocating, the mover (field magnet) position xm is an AC waveform, and the difference between the maximum displacement and the minimum displacement is the stroke. In the case of a free piston structure, the absolute values of the maximum displacement and the minimum displacement may differ. In this case, the center where the elastic forces of the elastic body are balanced is used as a reference (zero), and the positive maximum displacement is the positive stroke. The absolute value of the maximum negative displacement is called the negative stroke. That is, the addition of the positive stroke and the negative stroke is the stroke. The positive stroke and the negative stroke may differ depending on the driving conditions.

If the mass of the mover 6 or the spring constant of the resonance spring 23 deviates from the assumption, the resonance frequency will change from the assumed value, but the effects are the voltage Vm applied to the winding 8, the motor current Im, and the movable. Since it can be seen from the phase relationship of the speed of the child 6, a highly efficient linear motor system can be configured by controlling based on these phase relationships. Even when the load element added to the mover 6 has position dependence, it can be increased by controlling it based on the phase relationship between the voltage Vm applied to the winding 8 and the motor current Im, and the speed of the mover 6. An efficient linear motor system can be configured.
Hereinafter, the configuration or operation of each of the above-described units constituting the control unit 102 will be described.

<Reference phase creator>
^{The reference phase θ *,} which is the phase command value of this embodiment, is ^{obtained by integrating the drive frequency command value ω *} , which is the output of the drive frequency regulator 131 shown in FIG. 1, with the integrator 140 as the reference phase creator. .. That is, the reference phase theta ^* applied voltage Vm (θ ^*) is the phase theta ^* at each time of the wave having a driving frequency command value omega ^* corresponding to the target frequency. Thus, reference the phase theta ^* as is used driving frequency command value of the drive frequency regulator 131 omega ^*, for example, may be fixed to the mechanical resonance frequency of the vibration body including the movable member 6 in the present embodiment.
The reference phase θ ^* is a sawtooth wave whose range is, for example, [−π, π], [0,2π], or a wider range for each time while the drive frequency command value ω ^{* is constant.} Or it may be increased linearly with respect to the time. As will be described later, when the drive frequency command value ω ^* fluctuates, the shape of the sawtooth wave or linear increase fluctuates (the slope changes) accordingly.

<Phase difference detector 130>
When the mover 6 is reciprocating, the position xm and speed of the mover 6 and the motor current Im are periodic functions. Since the periodic function can be expressed by a Fourier series, if the position xm of the mover 6 is expressed by using the Fourier transform equation, it can be defined as the following equation (1).

Here, _{x 0} is the DC offset value, _{a n} and _{b n} are the Fourier coefficients of the order n is obtained by the following formula (2) and (3).

Here, T ₀ is the period of the fundamental wave (the period of reciprocating motion of the mover 6), that is, the reciprocal of the primary frequency (driving frequency). In a so-called spring mass system in which a resonance spring 23 is added to the mover 6, the mechanical resonance frequency determined by the mass of the mover 6 and the spring constant is the dominant component. Therefore, you should pay attention to the fundamental wave.

When controlling the mover 6 to be driven at the resonance frequency, the higher-order component is not important, and the primary component, that is, the drive frequency component may be focused on. In particular, the phase of the primary frequency component (drive frequency component) at the position xm of the mover 6 is important. The position xm of the mover 6 with respect to the sinusoidal applied voltage Vm can be obtained by the following equation (4) by the inverse tangent of the first-order Fourier coefficient.

In the equation (4), the integration range is -2π to 0. This is because when the phase difference detector 130 is realized by a semiconductor integrated circuit such as a microcomputer or a DSP (Digital Signal Processor), only past information can be acquired.

FIG. 10 is an explanatory diagram showing a configuration example of the phase difference detector 130 constituting the control unit 102 shown in FIG. 1, and is an explanatory diagram when the equation (4) is shown in a block diagram. ^{A reference phase θ *} , which is a phase command value, is input to each of the sine calculator 81 that outputs the sine of the input value and the cosine calculator 82 that outputs the cosine of the input value, and the reference phase θ ^* (phase command value) is relative to the reference phase θ * (phase command value). Obtain sine and cosine. The value obtained by multiplying each of the sine and the cosine by the mover position xm is output from the multiplier 92. Integrating the outputs with the integrators 94a and 94b gives the first-order Fourier coefficients of the sine and cosine, respectively. That is, since the frequency component higher than the drive frequency ω of the Fourier expansion can be eliminated, it can be robustly configured for higher-order noise.
The outputs of the integrators 94a and 94b are input to the inverse tangent 86. The inverse tangent device 86 outputs an inverse tangent value based on the input sine and cosine components. The inverse tangent 86 of this embodiment outputs the inverse tangent value of the phase in which the numerator is the output of the integrator 94a and the denominator is the output of the integrator 94b, but even if the numerator and the denominator are reversed, the value is output. good.

FIG. 11 is an explanatory diagram of the relationship between the drive frequency and the output from the phase difference detector 130. That is, it is a figure which shows the relationship between the drive frequency (horizontal axis) of an AC voltage, and the output value (phase difference dltθ (θpos ^)) (vertical axis) of an inverse tangent device 86. As can be seen from FIG. 11, in this embodiment, when the drive frequency is the resonance frequency, 0 ° is output from the inverse tangent device 86. The value output from the inverse tangent 86 is greater than 0 ° when the drive frequency is higher than the resonance frequency and less than 0 ° when the drive frequency is lower than the resonance frequency. As a result, the phase difference dltθ with respect to ^{the reference phase θ *} of the primary frequency component of the input AC signal to the phase difference detector 130 with respect ^{to the reference phase θ * (position xm of the mover 6 in this embodiment) can be obtained.} , The resonance frequency can be estimated. Then, for example, when the phase difference dltθ is more than 0 °, the phase difference dltθ may be controlled to decrease, and when the phase difference dltθ is less than 0 °, it may be controlled to increase. It is preferable to control the phase difference dltθ when the reference phase θ ^* and the fundamental frequency ω are equivalent as the target value dltθ ^*.

An incomplete integrator can be used instead of the integrators 94a and 94b. The incomplete integrator is a kind of low-pass filter (low-pass filter) and can have the same configuration as the first-order lag filter. Of course, the low-pass filter may be configured not only with the first-order lag filter but also with other known configurations such as a second-order lag filter. In addition, a high-pass filter (not shown) can be provided in front of the integrators 94a and 94b (or the incomplete integrator) in place of or in addition to the incomplete integrator. The cutoff frequency of the high-pass filter can be, for example, 10 Hz or 5 Hz or less.

In this way, the phase difference detector 130 uses the inverse tangent of the ratio of the first-order Fourier coefficients of the drive frequency component to obtain the phase difference at the position xm of the mover 6 with respect to the ^{AC voltage command value V *.} It has great sensitivity only to the primary frequency component at the position xm of the mover 6, which is the input AC signal to the detector 130. That is, for example, even when a DC offset or higher-order noise is superimposed on the position xm of the mover 6, the position xm of the mover 6 which is an input AC signal to the phase difference detector 130 with respect to ^{the reference phase θ *.} The phase dlt θ of the primary frequency component of is obtained more accurately. Further, when the high-pass filter is provided as described above, it can be robustly configured even for a frequency smaller than the drive frequency ω.
Therefore, as a method for detecting the position xm of the mover 6, it is particularly effective when a system in which noise is easily superimposed, for example, a system in which the inductance is highly dependent on the mover position or a system in which another device exists in the vicinity is adopted. Control can be realized. In this way, it is possible to detect or estimate the resonance frequency with high accuracy and realize highly efficient linear motor drive.

<Drive frequency regulator 131>
FIG. 12 is an explanatory diagram showing a configuration example of the drive frequency regulator 131 constituting the control unit 102 shown in FIG. The drive frequency regulator 131 obtains the difference between the phase difference command value dltθ ^* (for example, 0 °) and the phase difference dltθ obtained by the phase difference detector 130 with the subtractor 91, and obtains the proportional gain Kp_attr with the multiplier 92b. The calculation result proportionally controlled by multiplication and the calculation result of integration control by multiplying the integration gain Ki_attr by the multiplier 92c and integrating the result by the adder 94c are added by the adder 90, and the frequency command initial stage is further added to the addition result. The drive frequency command value ω ^* is output by adding the value (ω _0).
The initial value of the frequency command (ω ₀ ) may be obtained from a higher-level controller (not shown), or may be set to, for example, 0 ° in advance. Further, although the drive frequency regulator 131 of this embodiment has a configuration of proportional integration control, other control configurations such as proportional control and integral control can also be applied.

[Realization of high efficiency drive]
The operation of the phase difference detector 130 and the drive frequency adjuster 131 when the linear motor 104 is driven at a mechanical resonance frequency determined by the mass of the mover 6 and the spring constant will be described.
For example, if the mass of the mover 6 is heavier than the design value, the actual resonance frequency will be lower than the design value. That is, when the initial value of the drive frequency is determined using the mass design value of the mover 6 (when the initial value of the drive frequency command value ω ^* is determined using the design value), the actual resonance frequency is used. Will be driven at a high frequency. At this time, the phase difference dltθ obtained by the phase difference detector 130 becomes a value larger than the ^{phase difference command value dltθ *.} Therefore, the drive frequency regulator 131 ^{executes control to reduce the drive frequency command value ω *,} and as a result, the drive frequency command value ω ^* matches the actual resonance frequency. Therefore, the speed energy of the mover 6 can be effectively utilized, and the linear motor 104 can be driven with high efficiency.

<Voltage command value creator>
FIG. 13 is an explanatory diagram showing a configuration example of the voltage command value creator 103 constituting the control unit 102 shown in FIG. _{The voltage command value creator 103 generates a voltage command value V m2} ^* corresponding to the induced voltage generated according to the speed of the mover 6.
The upper figure of FIG. 13 is a so-called open command type configuration example without a stroke controller. The voltage command value generator 103 includes a cosine arithmetic unit 82b, multipliers 92d to 92f, and an end induced voltage regulator 133.
^{The voltage command value creator 103 has a stroke command value l *} obtained from a preset or upper controller (not shown) or the like ^{, and a drive frequency command value ω *} which is an output of the drive frequency regulator 131. ^{The reference phase θ *} , which is the output of the integrator 140 as the reference phase creator, and the mover position xm, which is the output of the position detector 106, are input.

^{The reference phase θ *} is input to the cosine arithmetic unit 82b that outputs the cosine of the input value, and the cosine output, the stroke command value l ^*, and the drive frequency command value ω ^* are multiplied by the multiplier 92d to obtain the speed. Obtain the command value vm ^*. The speed command value vm ^{* can also be obtained from the derivative of the command value xm *} at the position of the mover 6, and the arithmetic expression that is the basis of the configuration of FIG. 13 is shown in the following equation (5).

The velocity command value vm ^* can be represented by a sine wave having the stroke command value l ^{* as the amplitude.} Since the sine wave is time-differentiated to become a cosine wave, the differential operation can be omitted if the configuration shown in the upper figure of FIG.

A speed command value vm ^*, multiplied by the induced voltage constant Ke ^* at multiplier 92e, the induced voltage constant Ke ^* to generate a voltage command value _{V m1} ^* corresponding to the induced voltage when a constant.
Here, as a characteristic of the linear motor 104 to be driven, in the case where the change in the interlinkage magnetic flux of the armature 9 with respect to the position of the mover is as shown in FIG. 3, the induced voltage constant Ke ^* is constant when the stroke is large. Cannot be regarded as.
If the voltage command value V _m1 ^* is created with the induced voltage constant Ke ^* constant, the applied voltage becomes excessive (applied voltage> induced voltage) near the end of the stroke (maximum positive displacement and maximum negative displacement). The efficiency of the linear motor decreases due to the flow of useless current.
Further, not only the efficiency of the linear motor but also the loss in the power conversion circuit 105 described later increases. Further, in order to cope with a large current, it is necessary to use a switching element having a large current capacity (generally a large area). As a result, the power conversion circuit itself and the heat radiation fins become large, which makes it difficult to reduce the size of the linear motor drive device 101.
Therefore, in this embodiment, _{the voltage command value V m1} ^* corresponding to the induced voltage when the ^{induced voltage constant Ke *} is constant is subjected to the end induction which is the output of the end induction voltage regulator 133 by the multiplier 92f. By multiplying the voltage adjustment gain, the voltage command value V _m2 ^* applied to the linear motor 104 is generated. The end-induced voltage regulator 133, which will be described in detail later, inputs the mover position xm from the position detector 106 and outputs the end-induced voltage adjustment gain corresponding to the mover position xm.

The lower figure of FIG. 13 is a configuration example of a so-called close command type voltage command value creator 103b having a stroke controller 153. The voltage command value generator 103b includes a cosine arithmetic unit 82b, multipliers 92d to 92f, an end induced voltage regulator 133, and a stroke controller 153.
^{The voltage command value creator 103b has a stroke command value l *} obtained from a preset or upper controller (not shown) or the like ^{, and a drive frequency command value ω *} which is an output of the drive frequency regulator 131. ^{The reference phase θ *} , which is the output of the integrator 140 as the reference phase creator, and the mover position xm, which is the output of the position detector 106, are input.

^{The reference phase θ *} is input to the cosine arithmetic unit 82b that outputs the cosine of the input value, and the stroke command output from the stroke controller 153 based on the cosine output, the input stroke command value l ^{*, and the mover position xm.} The value l ^** and the drive frequency command value ω ^* are multiplied by the multiplier 92d to obtain the ^{speed command value vm *.}
A speed command value vm ^*, multiplied by the induced voltage constant Ke ^* at multiplier 92e, the induced voltage constant Ke ^* to generate a voltage command value _{V m1} ^* corresponding to the induced voltage when a constant. _{By multiplying the voltage command value V m1} ^* corresponding to the induced voltage when the induced voltage constant Ke ^* is constant by the end induced voltage adjustment gain which is the output of the end induced voltage regulator 133 with the multiplier 92f. , Generates a _{voltage command value V m2} ^* to be applied to the linear motor 104.
That is, the voltage command value creator 103b feeds back the stroke detection value or the stroke estimation value corresponding to the mover position xm, which is the output of the position detector 106, and the stroke controller 153 controls the stroke command value.

<End induced voltage regulator>
FIG. 14 is an explanatory diagram showing a first configuration example of the end-induced voltage regulator 133 constituting the voltage command value creators 103 and 103a shown in FIG. The end-induced voltage regulator 133 receives the mover position xm, which is the output of the position detector 106, as an input, and outputs the end-induced voltage adjustment gain. Various forms can be considered for determining the end-induced voltage adjustment gain. In either form, at the mover position xm where the rate of change of the magnetic flux is small, the voltage component applied according to the change of the magnetic flux is reduced in the applied voltage.

First, in the example shown in FIG. 14, the absolute value (| cosθ |) of the cosine function is output as the end induced voltage adjustment gain according to the input mover position xm. As shown in FIG. 3, the interlinkage magnetic flux of the armature 9 with respect to the position of the mover (field magnet) 6 of the linear motor 104 has a characteristic that the inclination becomes zero at the position of ± 0.01 m (± 10 mm). Have. ^{In other words, in FIG. 14, the induced voltage constant Ke *} becomes zero in the vicinity of -10 mm, which is the bottom dead center, and in the vicinity of 10 mm, which is the top dead center. This is because the velocity of the mover 6 becomes substantially zero near the top dead center and the bottom dead center, so that the inclination of the interlinkage magnetic flux of the armature 9 with respect to the mover 6 (the rate of change of the magnetic flux) becomes zero, so that the induced voltage Is due to the fact that is almost zero.
Therefore, by setting the end induced voltage adjustment gain to substantially zero _{, even when the voltage command value V m1} ^* corresponding to the induced voltage is ^{generated while the induced voltage constant Ke *} is constant, the voltage command value creators 103 and 103a The voltage command value V _m2 ^* output from is approximately zero. As a result, the voltage applied to the linear motor 104 is adjusted according to the change in the induced voltage of the linear motor 104, and it is possible to prevent the applied voltage from becoming excessive.
Therefore, the useless current flowing through the winding 8 can be suppressed, and the linear motor 104 can be driven with high efficiency. Further, since the loss in the power conversion circuit 105 can be suppressed, a switching element and a heat radiation fin having the minimum necessary current capacity can be adopted, and the linear motor drive device 101 can be downsized.

On the other hand, when the mover 6 reciprocates near the center position, the end induced voltage adjustment gain becomes a value close to 1, so that the induced voltage constant is constant and the voltage command value V _m1 ^* (corresponding to the induced voltage). = V _m2 ^* ) is applied to the linear motor 104. That is, a voltage close to the induced voltage is applied to the linear motor 104, and unnecessary current can be suppressed.

As described above, since the voltage command value creators 103 and 103a are provided with the end induced voltage regulator 133, a voltage corresponding to the induced voltage is always applied to the linear motor 104 regardless of the magnitude of the stroke or the positive / negative asymmetry of the stroke. Will be done. In other words, a voltage having an absolute value of the voltage component corresponding to the rate of change of the magnetic flux generated by the fluctuation of the relative positions of the armature 9 and the mover (field magnet) 6 is applied to the linear motor 104. This makes it possible to realize high efficiency and miniaturization of the linear motor drive device 101.
The characteristic curve showing the relationship between the mover position and the gain shown in FIG. 14 is stored in a storage unit (not shown in advance) in the end-induced voltage regulator 133. The end-induced voltage regulator 133 outputs the gain corresponding to the input mover position xm as the end-induced voltage adjustment gain based on the characteristic curve showing the relationship between the mover position stored in the storage unit and the gain. To do.

FIG. 15 is an explanatory diagram showing a second configuration example of the end-induced voltage regulator constituting the voltage command value creators 103 and 103a shown in FIG. The end-induced voltage regulator 133a outputs a value obtained by differentiating the relationship of the armature interlinkage magnetic flux with respect to the armature position shown in FIG. 3 at the mover position as the end-induced voltage adjustment gain.
When the mover 6 is located near the center, the slope of the armature differential magnetic flux (rate of change of magnetic flux) is close to 1, so the end-induced voltage regulator 133a outputs 1 as the end-induced voltage adjustment gain. To do. When the stroke is slightly increased (for example, 8 mm), the slope of the armature differential magnetic flux (rate of change of the magnetic flux) becomes smaller, so that the end induced voltage adjustment gain is set to a value smaller than 1. When the stroke is further increased and the mover 6 reciprocates to the end (top dead center or bottom dead center), the end induced voltage adjustment gain is close to zero when the mover position is around ± 10 mm. Is output.

FIG. 16 is an explanatory diagram showing a third configuration example of the end-induced voltage regulator constituting the voltage command value creators 103 and 103a shown in FIG. The end-induced voltage regulator 133b outputs 1 as the end-induced voltage adjustment gain up to the predetermined mover position (x1), and when the mover position exceeds x1, the absolute value of the mover position is calculated. Outputs the end-induced voltage adjustment gain that decreases monotonically with increasing. FIG. 16 shows an example in which the gain is linearly reduced as an example of the monotonically decreasing gain.

Depending on the configuration of the armature 9, the armature differential magnetic flux with respect to the armature position xm may not be sinusoidal, and higher-order components may be superimposed. In that case, the relationship between the mover position and the end-induced voltage adjustment gain may be changed so that the end-induced voltage adjustment gain does not decrease monotonically even if it exceeds a predetermined mover position, but has a maximum value. ..

As described above, although the end-induced voltage regulator shows some configuration examples, the voltage waveform applied to the linear motor 104 is the voltage waveform shown in FIG. FIG. 17 is an explanatory diagram of the applied voltage waveform, the upper figure of FIG. 17 shows the voltage waveform at the time of small scroll, and the lower figure of FIG. 17 shows the voltage waveform at the time of large scroll. As shown in the upper part of FIG. 17, when the stroke is small, the linear motor drive device 101 outputs a waveform close to a cosine wave to the linear motor 104. On the other hand, as shown in the lower figure of FIG. 17, when the stroke is large, the linear motor drive device 101 outputs a distorted waveform to the linear motor 104 in the region surrounded by the dotted line, that is, in the vicinity of the zero cross. As the stroke increases, the degree of distortion increases. In other words, when the waveform of the applied voltage is frequency-analyzed by FFT (Fast Fourier Transform) or the like, the distortion factor increases as the stroke increases.
In the lower figure of FIG. 17, the distortion of the voltage waveform in the region surrounded by the dotted line (near the zero cross) is due to the inclusion of harmonic components. That is, when the mover 6 is located near the end (top dead point or bottom dead point) based on the characteristic curve showing the relationship between the mover position and the gain shown in FIGS. 14 to 16 described above, the end portion When the end induced voltage adjustment gain output by the induced voltage regulator is multiplied by the multiplier 92f (FIG. 13), the voltage waveform of the _{voltage command value V m2} ^{* output from the voltage command value creators 103 and 103a is obtained.} A waveform in which the harmonics of the third component are superimposed, that is, a waveform in which the harmonics of the third component are included in the primary component. Therefore, the linear motor drive device 101 increases the content of harmonic components in the vicinity of the zero cross as the stroke increases, and outputs the applied voltage to the linear motor 104.

<Stroke controller 153>
FIG. 18 is an explanatory diagram showing a configuration example of the stroke controller 153 constituting the voltage command value creator 103a shown in the lower figure of FIG. As shown in FIG. 18, the stroke controller 153 ^{obtains the difference between the stroke command value l *} and the stroke detection value or stroke estimation value corresponding to the mover position xm with the subtractor 91a, and is proportional to this with the multiplier 92d. The calculation result proportionally controlled by multiplying the gain Kp_astr and the calculation result obtained by multiplying the integral gain Ki_astr by the multiplier 92e and integrating the result with the integrator 94d are added by the adder 90a, and the stroke command after adjustment is performed. Output the value l ^**. The stroke controller 153 of this embodiment has a configuration of proportional integration control, but other control configurations such as proportional control and integral control can also be applied.

By configuring the stroke controller 153 in this way, the mover 6 can be controlled according to the stroke command value, and even when the ^{induced voltage constant Ke * cannot be regarded as constant due to the change in the stroke.} , A voltage corresponding to the induced voltage is applied to the linear motor 104. As a result, it is possible to realize high efficiency and miniaturization of the linear motor drive device 103a.

<PWM signal creator 134>
The PWM signal generator 134 constituting the control unit 102 shown in FIG. 1 uses a known pulse width modulation by comparing ^{the carrier signal of the triangular wave with the voltage command value Vm *,} and drives according to the ^{voltage command value Vm *.} A signal is generated, and the generated drive signal is output to the power conversion circuit 105.

<Power conversion circuit 105>
FIG. 19 is a diagram showing a configuration example of the power conversion circuit 105 constituting the linear motor drive device 101 shown in FIG. The full bridge circuit 126 switches the DC voltage source 120 according to the drive signal input by the control unit 102, and outputs a voltage to the linear motor 104. The full bridge circuit 126 includes four switching elements 122, a first upper and lower arm (hereinafter referred to as U phase) having switching elements 122a and 122b connected in series, and a second upper and lower arm having switching elements 122c and 122d. It constitutes an arm (hereinafter referred to as V phase). The switching element 122 operates according to the pulsed gate signals (124a to 124d) output by the gate driver circuit 123 based ^{on the voltage command value Vm *} generated by the control unit 102 and the drive signal by pulse width modulation. it can.
By controlling the conduction state (on / off) of the switching element 122, the DC voltage of the DC voltage source 120 can be output to the winding 8 as a voltage corresponding to the AC voltage. A DC current source may be used instead of the DC voltage source 120. As the switching element 122, for example, a semiconductor switching element such as an IGBT or MOS-FET can be adopted.

[Connection with linear motor 104]
The switching elements 122a and 122b of the first upper and lower arms and the switching elements 122c and 122d of the second upper and lower arms of the power conversion circuit 105 are connected to the linear motor 104, respectively. Although FIG. 19 shows an example in which the windings 8 of the upper and lower armatures 9 are connected in parallel, the windings 8 can also be connected in series.

As described above, according to the present embodiment, the linear motor can be driven with high efficiency by suppressing the invalid current flowing in the winding without arranging the sensor for detecting the thrust of the linear motor in the linear motor. It becomes possible to provide a motor system. Specifically, at the mover position where the rate of change of magnetic flux is small, the voltage component (absolute value of the voltage component) applied according to the change of magnetic flux among the applied voltage is reduced to achieve a highly efficient linear motor system. Can be realized. In addition, an efficient linear motor system is achieved by increasing the content of harmonic components of the voltage applied to the windings in response to an increase in the relative positions of the mover (field magnet) and armature (increase in stroke). Can be realized.

The configuration of this embodiment can be the same as that of the first embodiment except for the following points. This embodiment relates to a closed compressor 50 as an example of a device equipped with a linear motor system 200 described later.

<Sealed compressor 50>
FIG. 20 is a vertical cross-sectional view of the closed-type compressor of the second embodiment according to another embodiment of the present invention, and is an example of a vertical cross-sectional view of the closed-type compressor 50 having a linear motor 104. The closed type compressor 50 is a reciprocating compressor in which the compression element 20 and the electric element 30 are arranged in the closed container 3. The compression element 20 and the electric element 30 are elastically supported in the closed container 3 by the support spring 49. The electric element 30 includes a mover (field magnet) 6 and an armature 9.
The compression element 20 includes a cylinder block 1 that forms the cylinder 1a, a cylinder head 16 that is assembled on the end surface of the cylinder block 1, and a head cover 17 that forms a discharge chamber space. The working fluid supplied into the cylinder 1a is compressed by the reciprocating motion of the piston 4, and the compressed working fluid is sent to a discharge pipe (not shown) communicating with the outside of the compressor. As the working fluid, for example, air or a refrigerant for a refrigeration cycle can be adopted.

A piston 4 is attached to one end of the mover (field magnet) 6. In this embodiment, the mover (field magnet) 6 and the piston 4 reciprocate to compress and expand the working fluid. This corresponds to a load in which the work required for compression and expansion fluctuates. A compression element 20 is arranged at one end of the electric element 30. The cylinder block 1 has a guide rod for guiding the reciprocating motion of the mover 6 along the front-rear direction.
A resonance spring 23 (not shown in FIG. 20) is added to the mover 6, and the mover (field magnet) 6 is set at a mechanical resonance frequency determined by the mass of the mover (field magnet) 6 and the spring constant. When reciprocating, it is necessary to consider the influence of the compression element 20 on the resonance frequency. That is, the suction pressure of the compression element 20 and the pressure of the discharge space add a spring-like action of the working fluid, so that the frequency at which the resonance state occurs changes. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover (field magnet) 6 being high, and the resonance frequency is high. On the contrary, when the pressure of the cylinder 1a is low, the spring constant of the resonance spring 23 added to the mover (field magnet) 6 becomes dominant, and the resonance frequency is the mass of the mover (field magnet) 6. It is close to the mechanical resonance frequency determined by the spring constant.

In this way, when the linear motor 104 is used as the power for the compression element 20, the resonance frequency changes depending on the conditions of the compression element 20 (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency according to the change of the load and the resonance frequency.
In addition, when the ^{voltage command value is created with the induced voltage constant Ke *} constant, the applied voltage becomes excessive (applied voltage> induced voltage) near the end of the stroke (maximum positive displacement and maximum negative displacement), which is useless. There is a problem that the efficiency of the linear motor 104 is lowered due to the flow of the current.

The effect of changing the resonance frequency can be seen from the phase relationship between the voltage applied to the winding 8 Vm, the motor current Im, and the speed of the mover 6. Therefore, a highly efficient linear motor system can be configured by controlling based on these phase relationships.

Further, when the linear motor 104 is installed in the closed container 3, an airtight connector called a hermetic connector or a hermetic seal may be used. To maintain airtightness, it is desirable to minimize the number of connectors. Therefore, in the linear motor system 200 of the present embodiment described later, the position of the mover 6 is estimated by the position estimator 135 from the voltage Vm applied to the linear motor 104 and the motor current Im flowing through the linear motor 104, and the position estimation value is obtained. Based on xm ^, the resonance frequency is detected or estimated with high accuracy to provide highly efficient linear motor drive.

FIG. 21 is an overall schematic configuration diagram of the linear motor system 200 of this embodiment. As shown in FIG. 21, the linear motor system 200 has the same configuration as the linear motor system 100 in 1 in the above embodiment, but outputs a load current detector 150, a current detector 107, and a voltage command value Vm ^* . The voltage command value creator 103b and the position estimator 135 are different. Hereinafter, the load current detector 150, the current detector 107, the voltage command value creator 103b, and the position estimator 135 will be described. Since other configurations are the same as those of the linear motor system 100 in the first embodiment, the description thereof will be omitted.

<Load current detector 150>
FIG. 23 is an explanatory diagram showing a configuration example of the load current detector constituting the control unit 202 shown in FIG. 21. The load current detector 150 of this embodiment extracts the amplitude of the load current component by using the Fourier transform equation.
Similar to the configuration of the phase difference detector 130 shown in FIG. 10, the load current detector 150 is used for each of the sine calculator 81 that outputs the sine of the input value and the cosine calculator 82 that outputs the cosine of the input value. ^{, The reference phase θ *} , which is the phase command value, is input, and the sine and cosine with respect to the reference phase θ ^{* are obtained.} A value obtained by multiplying each of the sine and the cosine by the motor current Im is output from the multiplier 92, respectively. When the outputs are integrated with the integrators 94e and 94f, respectively, the first-order Fourier coefficients of the sine and the cosine are obtained. That is, since the frequency component higher than the drive frequency ω of the Fourier expansion can be eliminated, it can be robustly configured for higher-order noise.

The outputs of the integrators 94e and 94f are squared and input to the square root calculator 96. That is, the square root of the sum of squares of the sine component and the cosine component, which is the first-order Fourier coefficient of each of the sine and cosine, is obtained, and the amplitude of the fundamental wave current is obtained. Since the amplitude of the fundamental wave current also increases as the load increases, the load current Im_ld can be detected by the configuration of FIG. 23.

FIG. 24 is an explanatory diagram showing another configuration example of the load current detector constituting the control unit 202 shown in FIG. 21. ^{The load current detector 150b inputs the reference phase θ *} , which is a phase command value, to the cosine arithmetic unit 82 that outputs the cosine of the input value, and obtains the cosine with respect to the reference phase θ ^*. By multiplying the obtained cosine and the motor current Im by the multiplier 92, the cos component (Im_cos) of the fundamental frequency of the motor current Im is extracted. Next, the cos component (Im_cos) of the fundamental frequency of the motor current Im is processed by a low-pass filter (low-pass filter) with the first-order lag filter 141, and is output as the load current Im_ld.
Although a configuration example of the first-order lag filter is shown in FIG. 24, the low-pass filter may be configured not only with the first-order lag filter but also with other known configurations such as a second-order lag filter.

As shown in FIGS. 8 and 9, when the position of the mover is a sine wave, the load current is dominated by the cosine component. Therefore, in the configuration of FIG. 24, the load current can be detected by filtering the extracted cosine component. The configuration shown in FIG. 24 is effective in reducing the calculation load.

<Voltage command value creator 103b>
FIG. 25 is an explanatory diagram showing a configuration example of the voltage command value creator 103b constituting the control unit 202 shown in FIG. 21. As shown in FIG. 25, the voltage command value creator 103b transmits the reference phase θ ^* , the position estimation value xm ^ which is the output of the position estimation device 135 described later, and the load current Im_ld detected by the load current detector 150. _{The voltage command value V m1} ^* corresponding to the induced voltage generated according to the speed of the mover (field magnet) 6 _{and the voltage command value V m3} ^* corresponding to the voltage drop due to the resistance and inductance of the linear motor 104. The voltage command value V _m2 ^* is output by adding the vector.

The load current Im_ld is multiplied by a preset or estimated resistance value R _m ^* and an inductance value L _m ^* by the multiplier 92, respectively. Next, the value obtained by multiplying the load current Im_ld and the resistance value R _m ^{* is multiplied by the output of the cosine arithmetic unit 82 that outputs the cosine of the reference phase θ *} , which is the input value, by the multiplier 92, and the load current Im_ld and the inductance are multiplied. Multiply the value obtained by multiplying the value L _m ^* by the output of the sine arithmetic unit 81a that outputs the negative sine ^{of the reference phase θ * which is the input value.} Further, these are added by the adder 90 and output as _{a voltage command value V m3} ^{* corresponding to the voltage drop.}

Since both the voltage command value V _m1 ^* and the voltage command value V _m3 ^* are AC waveforms, it is equivalent to outputting the vector sum (vector addition) in the voltage command value creator 103b.

FIG. 26 is a vector diagram showing the vector sum of the voltage command value creator 103b at the time of light load and heavy load, and is an explanatory diagram showing the vector addition of the voltage command value creator 103b in the vector diagram. The left figure of FIG. 26 is a vector diagram when the load is light, that is, when the load current is small, and the right figure of FIG. 26 is a vector diagram when the load is heavy, that is, when the load current is large. Since the stroke command value is the same in both the left and right views of FIG. 26, the voltage command value V _m1 ^* corresponding to the induced voltage is the same vector. In addition, in the left figure and the right figure of FIG. 26, the counterclockwise direction is positive.
_{In this way, by vector-adding the voltage command value V m1} ^* corresponding to the induced voltage to _{the voltage command value V m3} ^* corresponding to the voltage drop due to the resistance and inductance of the linear motor 104, it corresponds to the load current Im_ld. with the voltage command value V _{m @ 2} ^* amplitude increases Te, the AC voltage command phase theta _Vm ^* minute, the phase advances, the single-phase AC voltage command values V _{m @ 2} ^* is output. That is, even if the stroke of the mover 6 is the same, in other words, the speed command value vm ^* is the same, the voltage command value V _m2 ^* is appropriately controlled according to the load condition.

_{The voltage V m2} ^* applied to the linear motor 104 can be adjusted by changing any of the stroke command value l ^* , the phase command value θ ^* , and the speed command value vm ^*. Therefore, by adjusting the amplitude and frequency of the applied voltage, it is possible to control the drive frequency to the resonance frequency and control the stroke.

Similar to the first embodiment, the voltage command value creator 103b includes an end-induced voltage regulator 133. Therefore, by multiplying an appropriate end induced voltage adjustment gain according to the position of the mover, a voltage close to the induced voltage is applied to the linear motor 104, wasteful current can be suppressed, and the linear motor 104 is driven with high efficiency. be able to.

FIG. 27 is an explanatory diagram showing another configuration example of the voltage command value creator constituting the control unit 202 shown in FIG. 21. As shown in FIG. 27, the voltage command value creator 103c _{adds a voltage command value V m3} ^* corresponding to the voltage drop due to resistance and inductance _{to the voltage command value V m1} ^* corresponding to the induced voltage. It differs from the voltage command value creator 103b shown in FIG. 25 above in that the command value is multiplied by the end induced voltage gain, which is the output of the end induced voltage regulator 133, by the multiplier 92f. The voltage command value creator 103c is effective when the change in the inductance of the linear motor 104 is large at the end.
It is also possible to apply a gain only to the value obtained by multiplying the load current Im_ld and the inductance value L _m ^{* with the same configuration as the end induced voltage regulator 133.}

<Position estimator 135>
The position of the mover (field magnet) 6 is estimated by using the voltage Vm applied to the linear motor 104 and the current Im flowing through the linear motor 104, for example, to obtain the position estimation value xm ^ by the following equation (6).

In the formula (6), Vm ^* is a voltage command value Vm ^* applied to the linear motor 104.

FIG. 28 is an explanatory diagram showing a configuration example of the position estimator 135 constituting the control unit 202 shown in FIG. 21, and is an explanatory diagram when the equation (6) is shown in a block diagram. As shown in FIG. 28, the position estimator 135 ^{inputs the voltage command value Vm *} and the motor current Im, and multiplies the motor current Im by the winding resistance value Rm ^* of the linear motor 104 to obtain the voltage command value. It ^{is subtracted from Vm *, and} the result of this subtraction is calculated by an integrator. Further, the result obtained by multiplying the motor current Im by the winding inductance value Lm ^* of the linear motor 104 is obtained, subtracted from the result calculated by the above integrator, and the reduced result is divided by the ^{induced voltage constant Ke *.} This outputs the position estimation value xm ^. In addition to the above, a known synchronous motor position estimation method can be applied to the position estimator 135. If the stroke is calculated based on the position estimated value xm ^ and input to the stroke controller 153 shown in the lower figure of FIG. 13, the desired stroke can be controlled.

<Phase difference detector 130a>
In the first embodiment described above, the phase difference detector 130 shown in FIG. 10 is ^{configured to input the position xm of the mover (field magnet) 6 and the reference phase θ *} to calculate the phase difference dlt θ. On the other hand, in this embodiment, by using the motor current Im, the phase of the motor current Im with respect ^{to the reference phase θ * can be calculated as shown in the following equation (7).} This is because, as described above, when the resonance frequency changes from the assumed value, the effect can be seen from the phase relationship of the motor current Im.

FIG. 22 is an explanatory diagram showing a configuration example of the phase difference detector 130a constituting the control unit 202 shown in FIG. 21, and is an explanatory diagram when the equation (7) is shown in a block diagram. ^{A reference phase θ *} , which is a phase command value, is input to each of the sine calculator 81 that outputs the sine of the input value and the cosine calculator 82 that outputs the cosine of the input value, and the reference phase θ ^* (phase command value) is relative to the reference phase θ * (phase command value). Obtain sine and cosine. The value obtained by multiplying each of the sine and the cosine by the motor current Im is output from the multiplier 92. Integrating the outputs with the integrators 94a and 94b gives the first-order Fourier coefficients of the sine and cosine, respectively. That is, since the frequency component higher than the drive frequency ω of the Fourier expansion can be eliminated, it can be robustly configured for higher-order noise.
The outputs of the integrators 94a and 94b are input to the inverse tangent 86. The inverse tangent device 86 outputs an inverse tangent value as a phase difference estimated value dltθ ^ based on the input sine and cosine components. The inverse tangent 86 of this embodiment outputs the inverse tangent value of the phase in which the numerator is the output of the integrator 94a and the denominator is the output of the integrator 94b, but even if the numerator and the denominator are reversed, the value is output. good.
By detecting or estimating the resonance frequency with high accuracy using the motor current Im in this way, a highly efficient linear motor drive device 201 can be realized.

[Current detector 107]
FIG. 29 is a diagram showing a configuration example of the power conversion circuit 105 constituting the linear motor drive device 201 shown in FIG. 21. Since the configuration of the power conversion circuit 105 and the connection with the linear motor 104 are the same as those in the first embodiment, the description thereof will be omitted. As shown in FIG. 29, the U-phase lower arm and the V-phase lower arm are provided with a current detector 107 such as a CT (current transformer). Thereby, the current Im flowing through the winding 8 of the linear motor 104 can be detected.

As the current detector 107, for example, instead of CT, a shunt resistor 125 is added to the lower arm of the power conversion circuit 105, and a phase shunt current method is adopted to detect the current flowing through the linear motor 104 from the current flowing through the shunt resistor 125. it can. A single shunt current detection method that detects the current on the AC side of the power conversion circuit 105 from the DC current flowing through the shunt resistor 125 added to the DC side of the power conversion circuit 105 in place of or in addition to the current detector 107. You may adopt it. The single shunt current detection method utilizes the fact that the current flowing through the shunt resistor 125 changes with time depending on the energized state of the switching element 122 constituting the power conversion circuit 105.

As described above, according to the present embodiment, in the linear motor 104 of the sealed compressor 50, the voltage amplitude and the voltage phase are controlled according to the load, and the drive frequency is adjusted in consideration of the voltage phase. The linear motor can be driven at the mechanical resonance frequency including the load, a highly efficient linear motor system can be constructed, and at the mover position where the rate of change of the magnetic flux is small, the change of the magnetic flux among the applied voltages can be obtained. By reducing the voltage component (absolute value of the voltage component) applied accordingly, a highly efficient linear motor system can be configured.

The control unit 102 that constitutes the linear motor drive device 101 in the first embodiment and the control unit 202 that constitutes the linear motor drive device 201 in the second embodiment are composed of a semiconductor integrated circuit such as a microcomputer or a DSP, and are composed of software or the like. It is often realized by. Therefore, it is assumed that it will be difficult to verify whether the linear motor drive devices 101 and 201 in the above-described first and second embodiments are correctly configured. Therefore, in this embodiment, the present invention relates to a verification system 600 for verifying whether or not the configurations of the above-mentioned linear motor drive devices 101 and 201 are operating correctly. FIG. 30 is an explanatory diagram showing a configuration example of the verification system of the third embodiment according to another embodiment of the present invention.

The linear motor drive device includes a position detector 106 (Example 1) such as a laser displacement meter that detects the position (stroke of the mover 6) of the mover (field element) 6 constituting the linear motor 104, or the linear motor drive device. Is provided with a position estimator 135 (Example 2) that outputs an estimated value xm ^ of the position of the mover (field magnet) 6 based on the current detected value Im (motor current Im) which is the output of the current detector 107. Has been done.

As shown in FIG. 30, the verification system 600 is composed of at least a control unit 102 that outputs a drive signal to the power conversion circuit 105, a power conversion circuit 105, a linear motor 104, a position detector 106, and a verification device 190. To. The control unit 102 may be replaced with the control unit 202 shown in the second embodiment, and the position detector 106 may be replaced with the position estimator 135 shown in the second embodiment.
A voltmeter 193 is inserted in the connection between the power conversion circuit 105 and the linear motor 104. Thereby, the line voltage applied to the linear motor 104 is measured. The voltmeter may detect the difference between the potential of each phase and the potential of the DC voltage source 120 on the N (minus) side as the voltage of each phase.

<Verification device 190>
A voltage is applied to the linear motor 104 by switching the DC voltage source 120 according to the drive signal input by the control unit 102. The voltage waveform detected by the voltmeter 193 is a pulse-shaped waveform with pulse width modulation. In order to verify whether the linear motor drive devices 101 and 201 are correctly configured, the fundamental wave component is important. The low-pass filter 191 provided inside the verification device 190 outputs a drive frequency component from the voltage waveform detected by the voltmeter 193.
Here, as a simple method of outputting the drive frequency component, for example, 1/5 to 1 of the triangular wave carrier signal (usually several kHz to several tens kHz) of the PWM signal creator 134 constituting the control unit 102. A low-pass filter (low-pass filter, LPF) 191 having a lower cutoff frequency of about / 10 is used.
The stroke or mover position and the voltage waveform of the drive frequency component are confirmed by the recording device 194, and the change in the voltage waveform between the small stroke and the large stroke is measured as shown in FIG. 17 above. Therefore, it is possible to verify whether or not the control unit 102 (Example 1) or the control unit 202 (Example 2) is performing a predetermined operation.

Instead of the low-pass filter 191, each frequency component is measured by FFT conversion (fast Fourier transform), and the change in the distortion factor of the applied voltage waveform can also be verified. That is, whether or not the control unit 102 or the control unit 202 is performing a predetermined operation by measuring that the distortion factor of the voltage waveform in the vicinity of the zero cross shown in FIG. 17 described above increases as the stroke is loaded. Can be verified.

According to this embodiment, it is possible to easily verify whether or not the control unit constituting the linear motor drive device is performing a predetermined operation.

The configuration of this embodiment can be the same as that of Example 1 or 2 except for the following points. This embodiment relates to an air suspension system 300 as an example of a device equipped with a linear motor system.
FIG. 31 is a circuit diagram of the air suspension system 300 of the third embodiment according to another embodiment of the present invention, and FIG. 32 is a schematic view of a vehicle equipped with the air suspension system 300 shown in FIG. 31. However, in FIG. 32, only the distribution point 309N, which will be described later, and the components on the air suspensions 301 and 302 sides are shown.

As shown in FIG. 31, the air suspension system 300 includes two air suspensions 301 and 302, a compressor 303 driven by a linear motor 104, an intake filter 304, a first tank 305, an air dryer 307, and as a valve. It has three check valves 308, 315, 317, a supply / exhaust switching valve 310, two suspension control valves 311, 312, a return passage on-off valve 314, and an exhaust passage on-off valve 319. The air suspension system 300 connects them by a passage through which air can flow.

As shown in FIG. 32, the air suspension system 300 is a system mounted on a vehicle 400, for example, and controls the air pressure in the air chambers 301C and 302C (FIG. 31) of the air suspensions 301 and 302. For example, the left wheel 410L and the right wheel 410R of the vehicle 400 are provided with an axle 420 that connects these hubs and the like. For example, air suspensions 301 and 302 are provided between the wheel 410 side and the vehicle body 430 side, such as between the left wheel 410L and the right wheel 410R and the vehicle body 430, or between the hub and the vehicle body 430, and the air chamber 301C, By controlling the air pressure inside the 302C, the vehicle height can be adjusted.

As shown in FIG. 32, the air suspensions 301 and 302 may be attached between the axle 420 on the wheel 410 side and the vehicle body 430 of the vehicle 400, and the suspension arm connecting the wheel 410 and the vehicle body 430. It may be mounted between the class (wheel 410 side) and the vehicle body 430, or between the hub of the wheel 410 (wheel 410 side) and the vicinity of the vehicle body 430 mounting portion (vehicle body 430 side) of the upper arm of the suspension. As described above, the air suspensions 301 and 302 may be provided so as to support the wheels 410 and the vehicle body 430. For example, the air suspensions 301 and 302 can be provided between the wheels 410 and the vehicle body 430 in the vertical direction, and the wheels 410 and the vehicle body can be directly provided. It is not limited to the mode of attaching to 430.

In this embodiment, the air suspension system 300 having two air suspensions will be described, but the number of air suspensions included in the air suspension system 300 is not particularly limited as long as it is one or more. The number of air suspensions can be equal to, for example, the number of wheels. For example, in the case of a four-wheeled vehicle, a total of four air suspensions can be arranged, two on the two front wheel sides and two on the two rear wheel sides. In this embodiment, an example in which the cushioning cylinders 301A and 302A and the air chambers 301C and 302C serving as air springs are integrated is shown, but as is known on the large vehicle and the rear suspension side, the cushioning cylinder The (hydraulic shock absorber) 301A, 302A and the air spring may be provided independently.

As shown in FIG. 31, in the air suspensions 301 and 302, air chambers 301C and 302C are formed between the cushioning cylinders 301A and 302A and the piston rods 301B and 302B, respectively, to form an air spring. ing. Passages, which will be described later, are connected to each of the air chambers 301C and 302C, and the pressure and vehicle height are controlled by the operation of the air suspension system 300.

The compressor 303 can compress the air sucked from the suction port 303C and discharge it from the discharge port 303D. The compressor 303 is composed of a compressor main body 303A and a linear motor 104. A pressure sensor for measuring the pressure of the suction port 303C or the discharge port 303D, or both ports is provided.

A resonance spring 23 is added to the mover (field magnet) 6 of the linear motor 104, and the mover (field magnet) 6 is set at a mechanical resonance frequency determined by the mass and spring constant of the mover (field magnet) 6. In the case of reciprocating, as shown in FIG. 20 in the above-mentioned Example 2, it is necessary to consider the influence of the compression element 20 on the resonance frequency. That is, the suction pressure of the compression element 20 and the pressure of the discharge space add a spring-like action of the working fluid, so that the frequency at which the resonance state occurs changes. That is, when the pressure of the cylinder 1a is high, it is equivalent to the spring constant of the resonance spring 23 added to the mover (field magnet) 6 being high, and the resonance frequency is high. On the contrary, when the pressure of the cylinder 1a is low, the spring constant of the resonance spring 23 added to the mover 6 becomes dominant, and the resonance frequency is a mechanical resonance frequency determined by the mass of the mover 6 and the spring constant. Close to.

In this way, when the linear motor 104 is used as the power for the compression element 20, the resonance frequency changes depending on the conditions of the compression element 20 (suction pressure, discharge pressure, pressure difference between suction and discharge, etc.). Therefore, it is necessary to change the drive frequency according to the change of the load and the resonance frequency.

FIG. 33 is an explanatory diagram showing a configuration example of the end induced voltage regulator 133d constituting the voltage command value creator. The end-induced voltage regulator 133d includes a plurality of characteristic curves showing the relationship between the mover position and the gain, and the characteristic curves showing the relationship between the plurality of mover positions and the gain are associated with each pressure in advance and the ends are associated with each other. It is stored in a storage unit (not shown) in the part-induced voltage regulator 133d. Further, the characteristic curve showing the relationship between the plurality of mover positions and the gain is a characteristic curve showing the relationship between the mover position and the gain of FIGS. 14 to 16 shown in the above-described first embodiment, and is an end portion. The induced voltage regulator 133d is movable according to the pressure of the port among the characteristic curves showing the relationship between the gain and the positions of a plurality of movers stored in the storage unit (not shown) according to the pressure of the port. A characteristic curve showing the relationship between the child position and the gain is selected, and the end-induced voltage adjustment gain corresponding to the mover position xm is output. For example, depending on the pressure of the suction port 303C or the discharge port 303D of the compressor 303, the characteristic curve showing the relationship between one mover position and the gain is derived from the name of the characteristic curve showing the relationship between the plurality of mover positions and the gain. Be selected. As a result, the optimum position-induced voltage relationship can be used according to the change in pressure, and the voltage component (absolute value of the voltage component) applied according to the change in magnetic flux among the applied voltage can be reduced. , A highly efficient linear motor system can be configured.

As described above, according to the present embodiment, in the air suspension system 300, the voltage component (absolute value of the voltage component) applied in response to the change in the magnetic flux of the applied voltage is reduced to achieve a highly efficient linear structure. A motor system can be configured.

The compressor shown in the second embodiment can be applied to a compressor for pumping refrigerant in an air conditioner including a heat exchanger that functions as a condenser or an evaporator. Further, as the linear motor drive device for controlling the drive of the compressor, the linear motor drive device shown in the above-mentioned Example 1 or Example 2 can be adopted.
Furthermore, the compressor shown in the second embodiment described above can also be applied to a compressor that pumps a liquid refrigerant in a refrigerator having a condenser and an evaporator. Further, as the linear motor drive device for controlling the drive of the compressor, the linear motor drive device shown in the above-mentioned Example 1 or Example 2 can be adopted.

The present invention is not limited to the above-described examples, and includes various modifications. For example, the above-described embodiment has been described in detail in order to explain the present invention in an easy-to-understand manner, and is not necessarily limited to the one including all the described configurations.
Further, each of the above configurations, functions, processing units, processing procedures, and the like may be realized by hardware by designing a part or all of them by, for example, an integrated circuit. Further, each of the above configurations and functions may be realized by software by the processor interpreting and executing a program that realizes each function.
Although the linear motor 104 has been described as a single-phase machine, the configuration of the present invention can be applied to the three-phase machine, and the same effect can be obtained.
The power conversion circuit 105 may be in a mode of outputting a current. In this case, a current command value creator may be provided instead of the voltage command value creator.

1 ... Cylinder block 1a ... Cylinder 2 ... Permanent magnet 3 ... Sealed container 4 ... Piston 6 ... Movable element (field magnet)
7 ... Magnetic pole 8 ... Winding 9 ... Armature 16 ... Cylinder head 17 ... Head cover 20 ... Compression element 23 ... Resonant spring (assist spring)
30 ... Electric element 50 ... Sealed compressor 100, 200 ... Linear motor system 101, 201 ... Linear motor drive device 102, 202 ... Control unit 103, 103a, 103b, 103c ... Voltage command value creator 104 ... Linear motor 105 ... Power conversion circuit 106 ... Position detector 107 ... Current detector 126 ... Full bridge circuit 130, 130a ... Phase difference detector 131 ... Drive frequency regulator 133, 133a, 133b, 133c, 133d ... End induced voltage regulator 134 ... PWM signal generator 135 ... Position estimator 140 ... Integrator 150, 150b ... Load current detector 153 ... Stroke controller 190 ... Verification device 300 ... Air suspension system 600 ... Verification system

Claims (10)

At least, it has an armature having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an armature or an elastic body connected to the field magnet, and deformation of the elastic body. a linear motor for the field element and that the relative position of the armature varies, relatively reciprocate the armature and the field element in the axial direction in response to,
A control unit that applies a voltage to the winding is provided.
The control unit
Of the voltage applied to the winding, the absolute value of the applied voltage component is set to the relative position where the rate of change of the magnetic flux is small, based on the rate of change of the magnetic flux caused by the fluctuation of the relative position of the armature and the field magnet. Make it smaller with
When the relative position of the armature and the field magnet exceeds a predetermined value, the absolute value of the voltage component applied to the winding according to the rate of change of the magnetic flux is reduced.
Further, the control unit
It is provided with a voltage command value creator that generates a first voltage command value based on at least the position of the field detected by the position detector and the reference phase and frequency command value.
The voltage command value maker has a characteristic curve showing the relationship between the position of the field magnet and the gain in advance, and the end-induced voltage is the gain corresponding to the position of the field magnet detected by the position detector. It has an end-induced voltage regulator that outputs as adjustment gain.
The voltage command value creator is a linear motor system characterized in that the end-induced voltage adjustment gain is multiplied by the first voltage command value and output to the linear motor. At least, it has an armature having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an armature or an elastic body connected to the field magnet, and deformation of the elastic body. a linear motor for the field element and that the relative position of the armature varies, relatively reciprocate the armature and the field element in the axial direction in response to,
A control unit that applies a voltage to the winding is provided.
The control unit is a linear motor system characterized in that the content of harmonic components of a voltage applied to the winding is increased in accordance with an increase in the relative positions of the field magnet and the armature. In the linear motor system according to claim 2.
The control unit is a linear motor system characterized in that when the relative positions of the armature and the field magnet exceed a predetermined value, the content of harmonic components of the voltage applied to the winding is increased. In the linear motor system according to claim 3,
The control unit
It is provided with a voltage command value creator that generates a first voltage command value based on at least the position of the field detected by the position detector and the reference phase and frequency command value.
The voltage command value maker has a characteristic curve showing the relationship between the position of the field magnet and the gain in advance, and the end-induced voltage is the gain corresponding to the position of the field magnet detected by the position detector. It has an end-induced voltage regulator that outputs as adjustment gain.
The voltage command value creator is a linear motor system characterized in that the end-induced voltage adjustment gain is multiplied by the first voltage command value and output to the linear motor. In the linear motor system according to claim 4,
The characteristic curve showing the relationship between the position of the field magnet and the gain is a linear motor system characterized in that the gain decreases in a range where the position of the field magnet exceeds a predetermined value. In the linear motor system according to claim 3,
The control unit
A position estimator that estimates the position of the field based on at least the current detection value that flows through the winding detected by the current detector.
A first voltage command value is generated based on the position estimate value of the field magnet by the position estimator, a reference phase and a frequency command value, and a load current obtained by the current detection value and a voltage based on the reference phase. Equipped with a voltage command value creator that generates a second voltage command value corresponding to the drop
The voltage command value creator has a characteristic curve showing the relationship between the position of the field magnet and the gain in advance, and adjusts the gain according to the position estimated value of the field magnet by the position estimator at the end induced voltage. It has an end-induced voltage regulator that outputs as a gain,
The voltage command value maker adds the end-induced voltage adjustment gain to the second voltage command value by multiplying the first voltage command value, or the first voltage command value and the first voltage command value. A linear motor system characterized in that the addition result of the voltage command value of 2 is multiplied by the end-induced voltage adjustment gain and output to the linear motor. In the linear motor system according to claim 6,
The characteristic curve showing the relationship between the position of the field magnet and the gain is a linear motor system characterized in that the gain decreases in a range where the position of the field magnet exceeds a predetermined value. At least, it has an armature having a magnetic body in which windings are wound, a field magnet having a permanent magnet, and an armature or an elastic body connected to the field magnet, and deformation of the elastic body. A linear motor that reciprocates the field magnet and the armature relative to each other in the axial direction by changing the relative positions of the field magnet and the armature according to the above.
A control unit that applies a voltage to the winding is provided.
A compressor that compresses the working fluid by reciprocating the piston connected to the field magnet in the cylinder.
The control unit sets the absolute value of the voltage component to be applied to the change of the magnetic flux based on the rate of change of the magnetic flux generated by the fluctuation of the relative position between the armature and the field of the voltage applied to the winding. When the relative position between the armature and the field magnet exceeds a predetermined value, the voltage component of the voltage applied to the winding is applied according to the rate of change of the magnetic flux. Reduce the absolute value of
Et al., Wherein the control unit is,
A position estimator that estimates the position of the field based on at least the current detection value that flows through the winding detected by the current detector.
A first voltage command value is generated based on the position estimate value of the field magnet by the position estimator, a reference phase and a frequency command value, and a load current obtained by the current detection value and a voltage based on the reference phase. Equipped with a voltage command value creator that generates a second voltage command value corresponding to the drop
The voltage command value creator has a characteristic curve showing the relationship between the position of the field magnet and the gain in advance, and adjusts the gain according to the position estimated value of the field magnet by the position estimator at the end induced voltage. It has an end-induced voltage regulator that outputs as a gain,
The voltage command value maker adds the end-induced voltage adjustment gain to the second voltage command value by multiplying the first voltage command value, or the first voltage command value and the first voltage command value. A compressor characterized in that the addition result of the voltage command value of 2 is multiplied by the end-induced voltage adjustment gain and output to the linear motor. In the compressor according to claim 8,
The characteristic curve showing the relationship between the position of the field magnet and the gain is a curve in which the gain decreases in a range where the position of the field magnet exceeds a predetermined value. The working fluid compressed by the compressor according to claim 8 or 9 is supplied to a plurality of air chambers interposed between the vehicle body side and the wheel side and adjusting the vehicle height according to the supply and discharge of the working fluid. Air suspension system.

Images