JP5578240B2 - Linear motor control device - Google Patents

Description

Related applications

  This application claims the priority of Japanese Patent Application No. 2010-239602 filed on Oct. 26, 2010, which is incorporated herein by reference in its entirety.

  The present invention relates to a linear motor control device applied to a synchronous ground primary-side linear arrangement linear motor used for driving a machine tool conveyance apparatus, an industrial machine conveyance apparatus, and other various devices.

  Linear motors are widely used for traveling driving and the like in transport carts and the like of physical distribution devices (for example, Patent Document 1). The linear motor includes a linear induction motor (LIM), a linear synchronous motor (LSM), a linear direct current motor, and the like, but the linear induction motor is mainly used as a long-distance traveling system. Most of the linear synchronous motors have a system in which a magnet is arranged on the ground side and moves on the coil side.

  In addition, although there is an example in which primary coils are partially discretely arranged on the ground side in a linear synchronous motor (for example, Patent Document 2), the linear synchronous motor is an auxiliary use at a curved path or a path end. Basically, a linear induction motor is used. Further, Non-Patent Document 1 describes modeling for a ground primary type intermittent arrangement (discrete arrangement) linear synchronous motor in which a stator is arranged only at a place where acceleration / deceleration is necessary.

JP-A-63-114887 JP 2007-82307 A

Suzuki Kengo, Kim Yong, Co-authored by Hideo Hyakume, “Modeling of Intermittent Stator Arrangement of Permanent Magnet Type Linear Synchronous Motor”, IEEJ Linear Drive Study Group, LD-07-35, October 2007, pp17-pp22

  The linear induction motor has a low thrust and it is difficult to improve the running performance. For this reason, a linear synchronous motor has been tried to be applied to a transfer device that becomes a loader of a machine tool. Most conventional linear synchronous motors have a magnet moving on the ground side and moving on the coil side. However, in order to move the coil side, power must be supplied to the mover. Due to the wiring to the mover, the travel route is limited and the power supply system is complicated. Or For this reason, in a linear synchronous motor, it tried to arrange | position a primary coil on the ground side over the full path length. However, when the primary coil is disposed on the ground side, if the coil is continuously disposed over the entire length of the movement path as in the conventional linear motor, the amount of use of the coil increases and the cost increases.

  As a synchronous linear motor that solves such problems, a plurality of individual motors composed of armatures that can function as the primary armature of one independent linear motor are routed in the moving direction of the mover. We considered a discretely arranged linear synchronous motor arranged at intervals over the entire length. According to this configuration, since the individual motors are discretely arranged, the amount of coil used can be reduced, and the cost can be reduced.

  However, as a problem peculiar to the discretely arranged linear motor, the inductance and the induced voltage change depending on the position of the mover with respect to the individual motor, and disturbance (for example, cogging force) occurs at the end of the individual motor. These are great disturbances in controlling the motor. Although this problem is mentioned in Non-Patent Document 1, no control has been proposed in consideration of the influence.

  The object of the present invention is to achieve smooth movement control corresponding to changes in the induced voltage with respect to the position of the mover relative to the individual motor, while adopting the ground discrete arrangement form of the individual motor which is advantageous in terms of reduction of coil usage and power supply type. It is providing the linear motor control apparatus which can perform.

The linear motor control device of the present invention will be described with reference numerals used in the embodiments.
In this specification, for the sign of “x ^· ” indicating the speed detection value, “·” is added above the letter “x” for easy understanding in the figure, but it can be used in the specification. on the character limit, indicated by ^"x," denoted by the "-" in the upper right corner of the letter "x".

The linear motor control device according to the present invention comprises a plurality of individual motors (3) each having three phases of coils arranged in a linear direction and functioning as a primary armature of one linear motor (1). (4) A device for controlling a synchronous linear motor (1) arranged at intervals along the movement path and having the mover (4) made of a permanent magnet, each of the individual motors (3 ) And a general control means (7) for distributing the position command to the individual motors (3) according to the input position command. Each individual motor control means (6) has a position / speed control means (17) that performs both position control and speed control or only position control, and a current control means (13) that performs current control, The individual motor control means (6) includes a current detection means (14) for detecting a current component of each phase of the individual motor (3) and a position for detecting the position and speed of the mover (4). A detection means (15) and a speed detection means (16) are provided. The current control means (14) uses the detected value of the current detection means (14) with respect to the q-axis current command value i _q * which is a thrust current command value given from the position / speed control means (17). A thrust current control unit (18) for outputting a q-axis voltage command value V _q ′ for controlling the q-axis current detection value i _{q of the} obtained individual motor (3) to follow, and a set magnetic flux current command value A magnetic flux current control unit (19) that outputs a d-axis voltage command value V _d ′ that controls the detected d-axis current value i _{d of} the individual motor (3) to follow a certain d-axis current value i _d *. ), A coordinate conversion unit (20) that converts the q-axis voltage command value V _q ′ and the d-axis voltage command value V _d ′ into command values of the coordinates of each phase of the individual motor (3), and the coordinates A power converter (21) that converts the output of the converter (20) into a drive current for the individual motor (3) A vector control type. In this configuration, the mover detected by the speed detector (16) with respect to the q-axis voltage command value V _q ′ output from the thrust current controller (18) and input to the coordinate converter (20). There is provided an induced voltage compensation means (31) for adding the detected voltage value x ^· of (4) and the voltage compensation value Φx ^· obtained by the determined induced voltage constant Φ.

  Vector control is a technology that grasps motor current and flux linkage as instantaneous values of vectors, and controls those vectors with instantaneous values to make the instantaneous thrust of the motor follow the command. Since it is possible, it is widely used in the control of rotary motors. In the present invention, a vector is formed by the thrust current control unit, the magnetic flux current control unit, and the coordinate conversion unit. However, in the discretely arranged linear motor (1) in which the individual motor (3) which is the armature on the primary side is installed on the ground side with an interval, the inductance varies depending on the position of the mover (4) with respect to the individual motor (3). Change and the induced voltage changes. Control for these changes cannot be appropriately performed by general vector control alone.

In this regard, according to the above configuration, the speed detection means (16) detects the q-axis voltage command value V _q ′ output from the thrust current control section (18) and input to the coordinate conversion section (20). Since the induced voltage compensation means (31) for adding the voltage compensation value Φs obtained from the speed detection value s of the movable element (4) and the predetermined induced voltage constant Φ is provided, the position of the movable element (4) The q-axis voltage command value V _q ′ is appropriately compensated for the inductance change and induced voltage change caused by the above, and smooth operation of the mover (4) can be obtained. Moreover, since the linear motor (1) to be controlled is discretely arranged with the primary side armature (3) as the fixed side, the coil usage is small, and the power is supplied to the moving side. In comparison, the advantage of the discretely arranged linear motor (1) that the power feeding system can be simplified can be obtained. The induced voltage constant Φ used in the induced voltage compensation means (31) is a constant value at the intermediate portion in the moving direction of the mover of the individual motor, and the individual motor (3) and the mover (4) face each other at both ends. The value changes according to the facing area and gradually decreases toward the end side. For example, the induced voltage constant Φ may be a value that changes to a trapezoidal shape. As a result, smooth movement control corresponding to changes in the induced voltage depending on the position of the mover (4) can be realized with simple control that can be performed even with a processing apparatus having a relatively low arithmetic processing capability.

In the present invention, the following position change inductance compensation means (32) may be provided. The position change inductance compensation means (32) includes a position detection value x of the mover (4) detected by the position detection means (15) and a mover (4) detected by the speed detection means (16). Q-axis current detection value i _q and d-axis current detection value i _d obtained by converting the current detection value s of the current and the current value detected by the current detection means (1) into the q-axis and d-axis current values. The q-axis voltage compensation value and the d-axis voltage compensation value are calculated according to the determined calculation formula. The q-axis voltage compensation value is added to the q-axis voltage command value V _q ′ output from the thrust current control unit (18) and input to the coordinate conversion unit (20), and the magnetic flux current control unit (19 ) And the d-axis voltage compensation value is added to the d-axis voltage command value V _d ′ output to the coordinate conversion unit (20).

Thus, by calculating the q-axis voltage compensation value and the d-axis voltage compensation value and compensating the q-axis voltage command value V _q ′ and the d-axis voltage command value V _d ′, the position depends on the position of the movable element (4). The q-axis voltage command value V _q ′ and the d-axis voltage command value V _d ′ are appropriately compensated for the inductance change, and a smoother operation of the mover (4) can be obtained.

In the present invention, the predetermined cogging compensation current value i _cogging is subtracted from the thrust current command value given from the position / speed control means (17), and (18) is input to the thrust current control unit. It is preferable to provide cogging compensation means (33) for setting the q-axis current command value i _q *. Cogging compensation current value i _cogging allows for adequate cogging mitigation, because uniquely determined by the position of the movable element (4), it is Toko determined in advance by tests and the like. By performing the cogging compensation with the obtained value, the cogging due to the discrete arrangement of the individual motors (3) can be reduced.

τ_p：可動子の１磁極対のピッチ
Ｌ_d：Ｌ−Ｍ
Ｌ_q：Ｌ−Ｍ
Ｌ：各相の自己インダクタンス
Ｍ：各相間の相互インダクタンス
これにより、可動子（４）の位置によるインダクタンスの変化に対応した円滑な移動制御を、演算処理能力の比較的低い処理装置を用いても可能となる簡単な制御で実現可能となる。In the present invention, the q-axis voltage compensation value added by the position change inductance compensation means (32) is a value determined by the following equations (5q) and (5d).

τ _p : pitch of one pair of magnetic poles of the mover L _d : LM
L _q : LM
L: Self-inductance of each phase M: Mutual inductance between phases Thus, smooth movement control corresponding to the change in inductance depending on the position of the mover (4) can be performed using a processing device having a relatively low processing capacity. This can be realized by simple control that is possible.

  Any combination of at least two configurations disclosed in the claims and / or the specification and / or drawings is included in the present invention. In particular, any combination of two or more of each claim in the claims is included in the present invention.

The present invention will be more clearly understood from the following description of preferred embodiments with reference to the accompanying drawings. However, the embodiments and drawings are for illustration and description only and should not be used to define the scope of the present invention. The scope of the invention is defined by the appended claims. In the accompanying drawings, the same part number in a plurality of drawings indicates the same part.
1 is a block diagram showing an overall configuration of a linear motor control device according to an embodiment of the present invention. (A) is a plan view of an individual motor in the linear motor, and (B) is a sectional view thereof. It is a block diagram of the individual motor control means in the linear motor control device. It is a block diagram which shows the detail of the current control means in the linear motor control apparatus. It is a block diagram of the basic composition which omitted the compensation means from the current control means. It is explanatory drawing which shows the change of the inductance and linkage flux in a discrete arrangement | positioning linear motor. It is explanatory drawing which shows the relationship between the grade of compensation, and the burden of a control means. (A) is explanatory drawing when the difference of each phase is not considered about the change of an inductance and a linkage flux, (B) is explanatory drawing when the difference of each phase is considered. It is explanatory drawing of an induced voltage compensation parameter. It is explanatory drawing of an induced voltage constant. It is explanatory drawing of ideal induced voltage compensation on accuracy. It is explanatory drawing of an inductance compensation parameter. It is explanatory drawing of cogging compensation. It is a front view of the processing system containing the linear motor control apparatus of the same embodiment, and the conveying apparatus using the linear motor. It is a fracture | rupture top view of the same conveying apparatus.

  An embodiment of the present invention will be described with reference to FIGS. FIG. 1 shows a linear motor system including a linear motor 1 to be controlled and a linear motor control device 2. The linear motor 1 is a linear synchronous motor (LSM), and a plurality of ground-side individual motors 3 each composed of an armature capable of functioning as a primary armature of an independent linear motor, 4 is a discretely arranged linear motor installed at intervals in the moving direction X of four. The individual motors 3 are arranged over the entire movement range of the mover 4. Each individual motor 3 is installed on a common frame 5 having a rail (not shown) of the mover 4.

  In addition to this, a sensor 15 serving as a position detector for detecting the position of the mover 4 is installed for each individual motor 3 in the frame 5. In FIG. 1, the sensor 15 is shown between the individual motors 3 for convenience of illustration. However, in actuality, the sensor 15 is disposed at the same position as the individual motor 3 in the moving element moving direction (X direction). A linear sensor 15 capable of detecting the position in a slightly longer range is used. The mover 4 is provided with a plurality of N and S magnetic poles made of permanent magnets arranged in the mover base 4a in the moving direction X, and is guided by a rail (not shown) provided on the frame 5 so as to be freely advanced and retracted. Is done. The mover 4 is formed to have a length extending between the plurality of individual motors 3.

  Each individual motor 3 is formed by arranging a plurality of coils 3a and cores 3b serving as magnetic poles in each layer in the moving direction X, as shown in FIGS. Each core 3b is constituted by a portion protruding like a comb from a common main body. In this example, it is assumed that it is driven by a three-phase alternating current, and is a three-pole primary armature provided with one electrode for each phase (U, V, W phase). The individual motor 3 may be an armature provided with a plurality of electrodes for each phase (U, V, W phase) and having electrodes that are an integral multiple of the number of phases.

  The control system will be described with reference to FIG. The control device 2 includes a plurality of individual motor control means 6 that respectively control the individual motors 3, and one overall control means 7 that gives a position command to the plurality of individual motor control means 6. The overall control means 7 is composed of a weak electric circuit element, a computer, a part of its program, and the like. The overall control means 7 stores the assigned range obtained by dividing the movement range of the entire linear motor for each individual motor 3, and the command position of the position command input from the host control means (not shown) A position command corresponding to the motor 3 is converted and given. The position command corresponding to each individual motor 3 is a command obtained by coordinate conversion to the coordinates of the individual motor 3.

  Each individual motor control means 6 includes a high-electric motor driving circuit for supplying a motor current to the individual motor, and a weak electric control unit (not shown) for controlling the motor driving circuit. The weak electric system control unit is constituted by a microcomputer and its program and circuit elements, and performs feedback control shown in FIG.

  In FIG. 3, the individual motor control means 6 has a position control means 11, a speed control means 12, and a current control means 13 for performing feedback control of position, speed, and current, respectively, and a position loop, speed loop, and Performs feedback control of cascade control having a current loop. The position control means 11 performs feedback control of a predetermined position loop gain according to the deviation between the detected value of the sensor 15 for detecting the current position of the movable element 4 with respect to the individual motor 3 and the command value of the position command. The position control means 11 outputs a speed command value as its output.

  The speed control means 12 performs feedback control of a predetermined speed loop gain according to the deviation between the speed detection value of the movable element 4 obtained through the speed detection means 16 and the speed command value. The speed control means 12 outputs a current command value as its output. In this example, the speed detection unit 16 includes a differentiation unit that obtains the speed from the position detection value of the sensor 15. However, the speed detection unit 16 may be provided separately from the sensor 15 and directly detect the speed. Note that a combination of the position control unit 11 and the speed control unit 12 is referred to as a position / speed control unit 17. In this embodiment, the position control and the speed control are performed. However, the position / speed control means 17 may perform only the position control without performing the speed control.

  The current control means 13 detects the drive current applied to the individual motor 3 by the current detection means 14 such as a current detector, and determines a current command value corresponding to the deviation between the current detection value and the current command value. Generated using current loop gain to control motor drive current. Specifically, the current detection means 14 detects components of each phase of the drive current, and includes phase current detection units 14a and 14b (FIG. 4) that detect two of the three phases. If detection for two phases can be performed, the current component of the remaining one phase can be obtained by calculation.

  FIG. 4 shows details of the current control means 13 of FIG. The current control means 13 is a control means for performing vector control, and has various compensation means that characterize the present invention and the embodiment. For simplification of the description, first, each compensation means 31 from FIG. A basic configuration of vector control in which ~ 33 is omitted will be described with reference to FIG.

The current control unit 13 includes a thrust current control unit 18, a magnetic flux current control unit 19, a coordinate conversion unit 20, and a power conversion unit 21 as a basic configuration. The thrust current control unit 18 detects the q-axis current command value i _q *, which is the thrust current command value given from the speed control unit 12 of the position / speed control unit 17, from the detection value of the current detection unit 14. It is means for controlling the q-axis current detection value i _{q of the} individual motor 3 obtained via the coordinate αβ conversion unit 22 and the detected coordinate dq conversion unit 23 to follow, and the q-axis voltage command value V _q ′ is output as an output. Output. The thrust current control unit 18 includes a subtraction unit 18a that subtracts the q-axis current detection value i _q and a calculation unit 18b that controls the output of the subtraction unit 18a.

The magnetic flux current control unit 19 detects the detected coordinate αβ conversion unit 22 from the detection value of the current detection unit 14 with respect to the d-axis current value i _d * which is the magnetic flux current command value set in the magnetic flux current command value setting unit 29. And a means for controlling the d-axis current detection value i _{d of the} individual motor 3 obtained via the detected coordinate dq conversion unit 23 to follow, and outputs a d-axis voltage command value V _d ′ as an output. The magnetic flux current command value setting means 29 is appropriately set according to the motor characteristics of the individual motor 3, but normally the d-axis electric current value i _d * is set to “0”. Flux current controller 19, consisting of a subtraction unit 19a for subtracting the d-axis current detection value i _d, a calculation section 19b for controlling the output of the subtraction unit 19a.

The detected coordinate αβ converting unit 22 converts the detected values of the currents ia, ib, and ic flowing through the U phase, V phase, and W phase of the individual motor 3 into the actual current of the stationary orthogonal two-phase coordinate component (the actual current on the α axis). , And the actual current on the β-axis). Based on the phase of the mover 4, the detection coordinate dq conversion unit 23 converts the detection values iα and iβ of the actual current of the stationary quadrature two-phase coordinate component into detection values i _q and i _d on the q and d axes. Means. The q axis is an axis in the traveling direction of the linear motor, and the d axis is an axis perpendicular to the q axis. The mover phase input to the detection coordinate dq conversion unit 23 is a detected value of the phase obtained by converting the output of the sensor 15 which is a position detector by the magnetic pole table 24 and the sin / cos conversion unit 25. The magnetic pole table 24 is a table that converts the detected value of the linear position obtained from the sensor 15 into an electrical angle θ. The sin / cos conversion unit 25 is a means for converting between the cos and sin for the electrical angle θ output from the magnetic pole table 24.

なお、Ｋ_pは比例制御のゲイン、Ｋ_iは積分制御のゲイン、Ｋ_dは微分制御のゲインである。The calculation units 18b and 19b of the thrust current control unit 18 and the magnetic flux current control unit 19 perform PID control (proportional integral differential control), for example, by a predetermined calculation formula.
As this arithmetic expression, for example, the following expression is used.

K _p is a proportional control gain, K _i is an integral control gain, and K _d is a differential control gain.

The coordinate conversion unit 20 includes an αβ conversion unit 20a and an abc conversion unit 20b. αβ conversion unit 20a, the q-axis voltage command value V _q and the d-axis voltage command value V _d, based on the armature phase command value Vα of the actual voltage of the fixed two-layer coordinate components, is a means for converting the Vβ . The mover phase is obtained from the position detection value of the sensor 15 via the magnetic pole table 24 and the sin / cos converter 25. The abc converter 20b converts the actual voltage command values Vα and Vβ output from the αβ converter 20a into voltage command values Va, Vb and Vc for controlling the U phase, V phase and W phase of the individual motor 3. It is.

  The power conversion unit 21 is a means for converting the output of the coordinate conversion unit 20 into a drive current for the individual motor 3, and includes an inverter 21a and an output control unit 21b for controlling the inverter 21a. The output control unit 21b may be any unit as long as it can control the electric power output from the inverter 21a. The output control unit 21b is not particularly limited, and is, for example, means for performing pulse width modulation (PWM).

  The current control means 13 of the embodiment shown in FIG. 4 is based on the vector control described above with reference to FIG. 5 and includes an induced voltage compensation means 31, a position change inductance compensation means 32, and a cogging compensation means 33.

The induced voltage compensator 31 detects the speed detection value of the mover 4 detected by the speed detector 16 with respect to the q-axis voltage command value V _q ′ output from the thrust current controller 18 and input to the coordinate converter 20. This is a means for adding a voltage compensation value obtained by x ^· and a predetermined induced voltage constant Φ. This induced voltage compensation value is the product of the detected velocity value x ^· of the mover and the induced voltage constant Φ, Φx ^· . Induced voltage compensating means 31, consisting of an arithmetic unit 31a for calculating a voltage compensation value [Phi] x ^·, an adder 31b for adding the calculated voltage compensation value [Phi] x ^·.

The position change inductance compensation means 32 includes a calculation unit 32a and two addition units 32b and 32c. The calculation unit 32a includes a position detection value x of the mover 4 detected by the sensor 15 as position detection means, a speed detection value x ^{· of the} mover detected by the speed detection means 16, and a current detection means 14. The q-axis voltage compensation value and the q-axis current detection value i _q and the d-axis current detection value i _d obtained by coordinate-converting the detected current value into the q-axis and d-axis current values according to a predetermined arithmetic expression, The d-axis voltage compensation value is calculated. The adder 32 b adds the q-axis voltage compensation value calculated by the calculator 32 a to the q-axis voltage command value V _q ′ output from the thrust current controller 18 and input to the coordinate converter 20. . The adder 32 c adds the d-axis voltage compensation value calculated by the calculator 32 a to the d-axis voltage command value V _d ′ output from the magnetic flux current controller 19 and input to the coordinate converter 20. .

ただし、τ_pは可動子の１磁極対のピッチ、Ｌ_dはＬ−Ｍ、Ｌ_qはＬ−Ｍ、Ｌは各相の自己インダクタンス、Ｍは各相間の相互インダクタンスである。The calculation unit 32a of the position change inductance compensation unit 32 performs calculations of the following expressions (5q) and (5d), for example.

However, the tau _p pitch of one magnetic pole pair of the movable element, the L _d L-M, the L _q L-M, L each phase of self-inductance, M is the mutual inductance between the phases.

Cogging compensation means 33, to the thrust current command value given from the position and speed control means 17, subtracts the prescribed cogging compensation current value i _cogging, q-axis current command to be input to the thrust current control unit 18 This is a means for setting the value i _q *. That is, subtraction is performed from the q-axis current command value by a current compensation value i _cogging arbitrarily determined in consideration of disturbance at the end of the individual motor 3. Thereby, cogging when the mover 4 enters or protrudes into the individual motor 3 is alleviated. The cogging compensation means 33 includes a current compensation value setting unit 33a that determines the current compensation value i _cogging and a subtraction unit 33b that performs the subtraction.

Next, the reason why each of the compensation means 31 to 33 is required and the operation of the compensation means 31 to 33 will be described. First, problems in the discretely arranged linear motor are organized.
(Problem 1)
As shown in FIG. 6, when the mover 4 made of a permanent magnet enters and exits the individual motor 3, the coil inductance and the linkage flux change depending on the position.
(Problem 2)
For example, in FIG. 6, this change changes in the order of the U phase → the V phase → the W phase when the mover 4 enters the individual motor 3 or protrudes. In reverse running, the phase changes in the order of W phase → V phase → U phase.
Note that L _u , L _v , and L _w are coil inductances of the respective phases. φf _u , φf _v , and φf _w are flux linkages in a range where the phases of the movable element 4 and the individual motor 3 are opposed to each other. In FIG. 6, the trapezoid is changed as an example. Moreover, the figure shows the change which considered the difference of each phase.

  Although the compensator for solving the above problems 1 and 2 was considered, the introduction of these compensators was introduced depending on the performance (CPU processing speed and memory capacity) of the servo amplifier constituting the individual motor control means 6. It can be difficult. Therefore, the cases are divided into four cases shown in Table 1 below. In this embodiment, the compensators (compensating means 31 to 33) are introduced in the cases (1) and (2). The four cases (1) to (4) in the table are compensators that theoretically improve accuracy. Of these, cases (1) and (2) are compensators that can be introduced even if the servo amplifier performance is low.

モータ端部とは、固定子（個別モータ）領域において、磁石（可動子）と固定子のかかり具合が完全対向でない状態のことを指す。

The motor end means a state in which the degree of engagement between the magnet (movable element) and the stator is not completely opposed in the stator (individual motor) region.

  For those that do not consider the difference between the phases, a compensator is introduced on the assumption that the parameters change as shown in FIG. When the difference between the phases is not taken into consideration, the parameter changes at the end of the individual motor 3 as shown in FIG. FIG. 8B shows changes taking into account the differences between the phases, and is shown for comparison. Considering the difference in each phase, the parameter changes even in the middle of the individual motor 3 as shown in FIG. Note that Φf is an interlinkage magnetic flux on the individual motor 3 side when the mover 4 and the individual motor 3 face each other.

  FIG. 7 shows the relationship between the load on the memory and CPU and accuracy in the above four cases (1) to (4). If it is not necessary to consider the burden on the servo amplifier, it is ideal in calculation to use the compensator (4). However, in cases (3) and (4), the burden on the memory and CPU is high. Therefore, in this embodiment, in consideration of the burden on the servo amplifiers such as the memory and the CPU, the cases (1) and (2) are compensated.

The induced voltage compensation parameters will be described with reference to FIG. In the figure, τ _p is the length of one magnet of the individual motor 3 per pole and is the same as the pitch of one magnetic pole pair. The magnetic poles have an equal pitch. In the figure, x ^· represents the speed of the carriage, but is the speed of the movable element 4, and when the movable element 4 is installed on a traveling body such as a carriage, it is the speed of the traveling body. φ _{f1 to} φ _f5 are flux linkages around one magnetic pole pair. The pitch of the electrodes of each phase of the individual motor 3 is assumed to be equal. When this individual motor 3 is modeled, the interlinkage magnetic flux φ _f and the voltage V _q ″ are expressed by the following formulas as shown in the figure. Note that n represents the relationship between the individual motor 3 and the mover 4. It is the number of interlinkage magnetic fluxes of the opposing portions, and the interlinkage magnetic flux of the magnetic poles on both ends of the individual motor 3 is calculated by Φf _n × opposed area ratio.

By performing the model as described above, the flux linkage φ _f changes in a form as shown in FIG. Further, the induced voltage constant Φ is proportional to the flux linkage φ _f, and therefore can be expressed in the form of FIG. In the figure, the speed x ^· is constant and the difference between the phases is not taken into consideration.

The induced voltage compensation will be described. Since it is difficult to directly measure the flux linkage, for example, the induced voltage constant Φ is the movable element at a speed x ^· [m / sec] when the individual motor 3 and the movable element 4 are completely opposed to each other. 4 is obtained from the induced voltage generated when the vehicle 4 is run. At this time, the mover 4 is free of control and is pulled using another drive source (not shown). In this way, the induced voltage constant Φ is determined as follows: The induced voltage constant Φ obtained by the test in this way is used for calculation in the induced voltage compensation means 31 of FIG. The method for experimentally obtaining the induced voltage constant Φ is an example.

When the MAX value of the induced voltage constant Φ is obtained, the end portion changes in accordance with the facing area, and therefore can be written in the form of the above-described diagram of FIG. Induced voltage constant Φ used in the induced voltage compensating unit 31 in FIG. 4, a value is variable in response to such a position, that is, progressively smaller values at both ends of the individual motor 3. The position data when this variable induced voltage constant Φ is used is obtained from the position detection value x.

  The ideal induced voltage compensation will be described in terms of performance. In order to use an induced voltage compensator with higher accuracy, it is necessary to use the following model shown in FIG.

φ_fu，φ_fv，φ_fwに関しては、各相のモータが完全対応したときの、各相の鎖交磁束を最大とし、モータ端部では面積比に応じて値を変化するようなモデル化を行う。

For φ _fu , φ _fv , and φ _fw , modeling is performed so that the interlinkage magnetic flux of each phase is maximized when the motor of each phase is fully supported, and the value changes according to the area ratio at the motor end. Do.

  In this case, the calculation by the induced voltage compensation means 31 in FIG. 4 shows that the induced voltage coefficient Φ varies according to the area ratio of the opposing areas at the end of the individual motor 3 as shown in FIG. The value will change in consideration of the above.

The inductance compensation parameters will be described with reference to FIG. In the figure, φ _u , φ _v , and φ _w indicate magnetic fluxes generated when a current flows in each phase. This includes mutual inductance as a term that affects each other. φ _u , φ _v , and φ _w are expressed by the following equations. The coil resistance R is, for example, the same for all phases. As an example, R _u = R _v = R _w = R.

In this embodiment, the inductance determined by the position change inductance compensating means 32 uses a value determined by the following equation.

  The inductance compensation in the above equation corresponds to the term enclosed by the line on the right side in the following relational equation. This inductance compensation is compensation that does not take into account changes in inductance, requires only a small calculation load, and can be employed even if the capacity of the memory or CPU in the servo amplifier constituting the individual motor control means 6 is low.

The ideal inductance change compensation will be described in terms of accuracy. In order to use a more accurate inductance change compensator, it is necessary to use a model of the following equation.

When formula expansion is performed for the inductance change part, the following formula is obtained.

  In the formula of FIG. 6, the part surrounded by a solid line is a part related to the reverse axis current compensation, and the part surrounded by a dotted line is a part related to L position / time change compensation. These portions are used in the position change inductance compensation means 32 of FIG. In addition, the part enclosed with a dashed-dotted line is a term regarding the time constant of control.

The cogging compensation will be described. As shown in FIG. 13, when the mover 4 enters or protrudes into the individual motor 3, a pulling force is generated between the mover 4 and the individual motor 3 and affects the control as a disturbance. This disturbance can be predicted and modeled in advance, and the disturbance due to the pulling force can be suppressed by issuing a current command in consideration of the disturbance. The cogging compensation means 33 in FIG. 4 performs subtraction from the q-axis current command value by a current compensation value i _cogging arbitrarily determined in consideration of this disturbance. Thereby, cogging when the mover 4 enters or protrudes into the individual motor 3 is alleviated.

In order to clarify the motor model and compensation, the motor circuit equation, induced voltage, inductance, etc. will be summarized and explained. When the relational expression between the voltage and current of the three-phase synchronous linear motor in this embodiment is expressed, the following expression is obtained.

In order to make it easy to control alternating current as direct current, coordinate transformation of the three-phase rotation equation into two phases yields the following equation.

When coordinate transformation is performed and summarized in the qd coordinate system, the following expression is obtained.

A portion of the above equation is expanded for clarity and is shown below.

Originally, the flux linkage of the mover 4 facing each phase of the individual motor 3 at the end of the discrete linear motor is proportional to the facing area. In consideration of this, an induced voltage compensator with higher accuracy can be created. However, depending on the amplifier performance, it is necessary to simplify the mathematical formula. Here, when the individual motor 3 and the mover 4 are completely opposed to each other, the interlinkage magnetic flux φ _fu = φ _fv = φ _fw = φ _{f of} each phase is obtained, and the term relating to the induced voltage is summarized by the following equation.

一般に、一次側電気子と永久磁石とが完全に対向したリニアモータや回転モータ等では、φ_fu＝φ_fv＝φ_fw＝φ_fと置いても、その誤差は非常に小さい。

In general, in a linear motor, a rotary motor, or the like in which the primary-side electric element and the permanent magnet are completely opposed to each other, even if φ _fu = φ _fv = φ _fw = φ _f , the error is very small.

A portion of the above equation is expanded for clarity and is shown below.

When the inductance matrix (L _{5 to} L ₈ ) is written, it can be summarized as the following first half formula. Furthermore, when the inductances (L _{1 to} L ₄ ) included in (L _{5 to} L ₈ ) are written, the following latter half expression is obtained.

From the above, L _{5 to} L ₈ are obtained from the self-inductance (L _u , L _v , L _w ) and the mutual inductance (M _uv , M _vw , M _wu ) of each coil. That is, a more accurate compensator can be obtained by appropriately modeling these six inductances.

When the self-inductance of each phase is (L _u = L _v = L _w ) and the mutual inductance is (M _uv = M _vw = M _wu ), that is, (L = L _u = L _v = L _w ), (M = M _uv = M _vw = M _wu )
L ₅ = L−M = L _d ,
L ₆ = 0,
L ₇ = 0,
L ₈ = L−M = L _q ,
And can be written as:

In this equation, voltage compensation is performed on the portion surrounded by a line as follows, so that a simple first-order lag system is formed, and control is facilitated.

  Next, with reference to FIGS. 14 and 15, a description will be given of a machining system including the conveying device 41 and the machine tool 42 to which the linear motor 1 and the linear motor control device are applied. As shown in FIG. 14, the machine tool 42 is a lathe in the illustrated example, and on a bed 51, a spindle base 53 that supports a work support means 52 that is a spindle, and a turret-type tool post 54 that is a processing means. And are installed.

  The conveying device 41 is configured such that a traveling body 43 that conveys a workpiece W that is a material for processing is installed on a rail 44 so that the traveling body 43 can freely travel, and the linear motor 1 is provided as a motor that drives the traveling body 43 to travel. Yes, the workpiece W is delivered to the workpiece support means 52 of the machine tool 42. The rail 44 is provided along a longitudinal direction on a horizontal frame 45 constructed by a column 45a.

  As shown in FIG. 15, the traveling body 43 includes traveling wheels 61 guided by the rails 44, and guide rollers 62 that are in contact with the side surfaces of the rails 44 and regulate the position in the width direction of the traveling body 43. . The linear motor 1 includes a plurality of individual motors 3 installed on the frame 46 and a mover 4 installed on the traveling body 43.

  The traveling body 43 is mounted with a front / rear moving table 46 that moves back and forth in the front / rear direction (Z direction) orthogonal to the traveling direction (X direction), and the lower end of a rod-shaped lifting body 47 that is installed on the front / rear moving table 46 so as to be movable up and down. A work holding head 48 is provided. The work holding head 48 is provided with a plurality of chucks 49 serving as conveyance object holding means. The front / rear moving table 46 is moved back and forth by a drive source (not shown) such as a motor installed on the traveling body 3, and the elevating body 47 is driven up and down by a drive source such as a motor installed on the front / rear moving table 46. The The chuck 49 has a chuck claw (not shown) that is driven to open and close by a drive source such as a cylinder device or a solenoid and holds the conveyed object W.

  By applying the discretely arranged synchronous linear motor 1 to the conveying device 41 for the machine tool 42 as described above, the advantageous effects of reducing the amount of coil usage and the power supply type are effectively exhibited. The Further, by applying this linear motor control device to the control of the linear motor 1, the effect of smooth movement control corresponding to the change of the induced voltage and the change of the inductance is effectively exhibited.

  As described above, the preferred embodiments of the present invention have been described with reference to the drawings, but various additions, modifications, or deletions can be made without departing from the spirit of the present invention. Therefore, such a thing is also included in the scope of the present invention.

DESCRIPTION OF SYMBOLS 1 Linear motor 2 Linear motor control apparatus 3 Individual motor 4 Movable element 5 Frame 6 Individual motor control means 7 General control means 11 Position control means 12 Speed control means 13 Current control means 14 Current detection means 15 Sensor (position detection means)
16 speed detection means 17 position / speed control means 18 thrust current control section 19 magnetic flux current control section 20 coordinate conversion section 21 power conversion section 31 induced voltage compensation means 32 position change inductance compensation means 33 cogging compensation means

Claims (4)

A plurality of individual motors that can function as the primary armature of one linear motor are arranged in a straight line with coils of each phase of three phases arranged at intervals along the moving path of the mover, and the movable A device for controlling a synchronous linear motor having a child composed of permanent magnets,
A plurality of individual motor control means for controlling each of the individual motors, and overall control means for distributing position commands to the individual motors in accordance with the input position commands,
Each individual motor control means has a position / speed control means for performing both position control and speed control or only position control, and a current control means for performing current control. A current detecting means for detecting, a position detecting means and a speed detecting means for detecting the position and speed of the mover, respectively;
The current control means detects a q-axis current of an individual motor obtained from a detection value of the current detection means with respect to a q-axis current command value i _q * which is a thrust current command value given from the position / speed control means. A thrust current control unit that outputs a q-axis voltage command value V _q ′ that is controlled so that the value i _q follows, and an individual motor for a d-axis current value i _d * that is a set magnetic flux current command value A magnetic flux current control unit that outputs a d-axis voltage command value V _d ′ that is controlled so that the detected d-axis current value i _d follows, and these q-axis voltage command value V _q ′ and d-axis voltage command value V _d ′. And a power conversion unit that converts the output of the coordinate conversion unit into a drive current of the individual motor, and Q-axis voltage output from the current control unit and input to the coordinate conversion unit Induced voltage compensation means for adding the detected value x ^· of the mover detected by the speed detecting means to the command value V _q ′ and the voltage compensation value Φx ^· obtained from the determined induced voltage constant Φ. The induced voltage constant Φ used in the induced voltage compensation means is set to a constant value at an intermediate portion in the moving direction of the mover of the individual motor, and according to the facing area where the individual motor and the mover face at both ends. Change to a value that gradually decreases toward the end side,
Linear motor control device. The position detection value x of the mover detected by the position detection means, the speed detection value x ^{· of the} mover detected by the speed detection means, and the current value detected by the current detection means are q-axis and d-axis. A q-axis voltage compensation value and a d-axis voltage compensation value are calculated according to a predetermined calculation formula using the q-axis current detection value i _q and the d-axis current detection value i _d obtained by converting the coordinates into the current value of
The q-axis voltage compensation value is added to the q-axis voltage command value V _q ′ output from the thrust current control unit and input to the coordinate conversion unit, and output from the magnetic flux current control unit to output the coordinate conversion unit. The linear motor control device according to claim 1, further comprising position change inductance compensation means for adding the d-axis voltage compensation value to the d-axis voltage command value V _d ′ input to. A predetermined cogging compensation current value i _cogging is subtracted from the thrust current command value given from the position / speed control means to _{obtain the} q-axis current command value i _q * to be input to the thrust current control unit. The linear motor control device according to claim 2, further comprising cogging compensation means. The q-axis voltage compensation value added by the position change inductance compensation means and the d-axis voltage compensation value are values determined by the following expressions (5q) and (5d). Linear motor control device.

τ _p : pitch of one pair of magnetic poles of the mover L _d : LM
L _q : LM
L: Self-inductance of each phase M: Mutual inductance between phases

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