JP6841018B2 - Rotating machine control device - Google Patents

Description

In the present invention, a rotary electric machine having a stator in which a winding is wound and a rotor in which magnetic poles are formed is electrically connected to the rotary electric machine, and a driving current is passed through the winding to drive the rotary electric machine. It relates to a power converter to be driven and a control device for a rotary electric machine applied to a system including.

As a control device of this type, as seen in Patent Document 1 below, a device including a storage device for storing fluctuation information of an electromagnetic force acting on a stator core and a correction information generation circuit is known. The correction information generation circuit generates correction information for correcting the fluctuation of the electromagnetic force based on the fluctuation information of the electromagnetic force read from the storage device according to the position of the magnetic pole of the rotor. Here, the fluctuation information of the electromagnetic force is the fluctuation information of the electromagnetic force component in the radial direction acting on the stator. The control device corrects the harmonic current waveform flowing through the stator winding based on the correction information. As a result, the radial electromagnetic force fluctuation acting on the stator is reduced.

Japanese Patent No. 3366858

The inventor of the present application has faced a problem that the vibration and noise of the rotating electric machine increase due to the radial electromagnetic force fluctuation acting on the rotor, apart from the radial electromagnetic force fluctuation acting on the stator. .. Therefore, it is conceivable to reduce this electromagnetic force fluctuation by passing a harmonic current that reduces the radial electromagnetic force fluctuation acting on the rotor through the winding. However, due to individual differences in the rotating electric machines, the magnetic flux characteristics of the magnetic poles vary from mass-produced rotating electric machines. If the magnetic flux characteristics vary, the appropriate harmonic current for reducing the radial electromagnetic force fluctuation may differ for each rotating electric machine. Therefore, in order to reduce the fluctuation of the electromagnetic force in the radial direction, it is required to determine an appropriate harmonic current for each rotating electric machine.

The present invention has been made to solve the above problems, and an object of the present invention is to provide a control device for a rotating electric machine capable of reducing radial electromagnetic force fluctuations acting on a rotor.

The present invention includes a rotary electric machine (10) having a stator (12) around which a winding (12U to 12W) is wound and a rotor (14) on which a magnetic pole (14a) is formed, and an electric current to the rotary electric machine. A power converter (20) that is connected to and drives the rotary electric machine by passing a drive current through the winding, and a magnetic flux information acquisition unit (30) that is applied to a system and acquires magnetic flux information of the magnetic poles. , The fundamental wave setting unit (30) for setting the fundamental wave command value for passing the fundamental wave current through the winding, and the harmonic current for reducing the radial electromagnetic force fluctuation acting on the rotor. A harmonic that is used as a current and is superimposed on the fundamental wave command value, and a harmonic command value for passing the reduced current through the winding is set based on the magnetic flux information acquired by the magnetic flux information acquisition unit. The setting unit (30) and the current obtained by superimposing the reduced current on the fundamental wave current are used as the driving current, and the fundamental wave command value set by the fundamental wave setting unit and the harmonic setting unit set by the harmonic setting unit. An operation unit (30) for operating the power converter so that the drive current flows in the winding based on the harmonic command value is provided.

If the magnetic flux characteristics of the magnetic poles vary from one rotating electric machine to another due to individual differences in the rotating electric machines, the electromagnetic force component that fluctuates with the fluctuation angular velocity of the magnetic pole magnetic flux during the rotation of the rotor varies from one rotating electric machine to another. Therefore, in order to reduce the electromagnetic force in the radial direction, it is required to grasp the electromagnetic force component that fluctuates with the fluctuating angular velocity of the magnetic pole magnetic flux for each rotating electric machine.

Here, according to the magnetic flux information of the magnetic poles, it is possible to grasp the electromagnetic force component that fluctuates with the fluctuation angular velocity of the magnetic pole magnetic flux. Therefore, even if the magnetic flux characteristics of the magnetic poles vary from one rotating electric machine to another due to individual differences in the rotating electric machines, the magnetic flux information of the magnetic poles reduces the radial electromagnetic force fluctuation acting on the rotor. The reduced current, which is a harmonic current, can be set by reflecting the individual differences of the rotating electric machine.

Therefore, in the above invention, based on the magnetic flux information acquired by the magnetic flux information acquisition unit, the reduced current is superimposed on the fundamental wave command value for flowing the fundamental wave current through the winding, and the reduced current is passed through the winding. The harmonic command value is set. Then, based on the fundamental wave command value and the harmonic command value, the power converter is operated so that the drive current, which is the current obtained by superimposing the harmonic current on the fundamental wave current, flows through the winding. As a result, even if there are individual differences in the rotating electric machine, it is possible to reduce the radial electromagnetic force fluctuation acting on the rotor.

In the above invention, the fluctuation angular velocity of the electromagnetic force in the radial direction is the same as the fluctuation angular velocity of the magnetic pole magnetic flux, whereas the fluctuation angular velocity of the electromagnetic force acting on the stator core to be reduced in Patent Document 1 is the magnetic pole. And the fluctuation angular velocity of the magnetic flux between the windings is doubled. In Patent Document 1, the magnetic flux is detected by the search coil on the surface of the tooth on which the winding is wound, and the magnetic flux detected by the search coil is the magnetic flux generated by energizing the winding and the magnetic flux magnetic flux. This is because it includes and.

The overall block diagram of the motor control system which concerns on 1st Embodiment. Sectional view of the motor. The figure which shows the resonance mode of a rotor. The figure which shows the transition of the fundamental wave current which a harmonic current is superimposed. The flowchart which shows the procedure of the correction processing of a harmonic voltage. The overall block diagram of the motor control system which concerns on 2nd Embodiment. The flowchart which shows the procedure of the generation process of a harmonic voltage. The figure which shows an example of the magnetic flux distribution. The figure which shows the change of the electromagnetic force before and after superimposing a harmonic voltage. The overall block diagram of the motor control system which concerns on 3rd Embodiment. The overall block diagram of the motor control system which concerns on 4th Embodiment. The overall block diagram of the motor control system which concerns on other embodiments.

<First Embodiment>
Hereinafter, a first embodiment in which the control device according to the present invention is applied to a blower motor constituting an in-vehicle air conditioner will be described with reference to the drawings.

As shown in FIG. 1, the in-vehicle motor control system includes a motor 10, a three-phase inverter 20 as a power converter, and a control device 30. The motor 10 is electrically connected to the battery 40 as a DC power source via the inverter 20.

The inverter 20 includes three series connections of the upper arm switches SUp, SVp, SWp and the lower arm switches SUn, SVn, SWn. The connection points of the U-phase upper and lower arm switches Sup and Sun are connected to the first end of the U-phase stator winding 12U constituting the stator 12 (stator) of the motor 10. The connection points of the V-phase upper and lower arm switches SVp and SVn are connected to the first end of the V-phase stator winding 12V, and the connection points of the W-phase upper and lower arm switches SWp and SWn are the W-phase stator winding 12W. It is connected to the first end of. The second ends of the stator windings 12U, 12V, and 12W are star-connected by being connected at the neutral point N. Freewheel diodes DUp, DVp, DWp, DUn, DVn, and DWn are connected in antiparallel to each switch SUp, SVp, SWp, SUn, SVn, and SWn. Incidentally, as each switch, for example, a voltage-controlled semiconductor switching element can be used, and specifically, for example, an IGBT or MOSFET can be used.

In this embodiment, a centralized permanent magnet synchronous machine is used as the motor 10. In particular, in the present embodiment, as the motor 10, an outer rotor type motor is used as shown in FIG. Here, FIG. 2 shows a cross-sectional view of the motor 10 cut along a plane orthogonal to the axial direction of the motor 10, that is, the rotation axis direction of the rotor 14. The center point O shown in FIG. 2 is a point through which the rotation axis passes. Further, in FIG. 2, hatching for displaying a cross section is omitted.

As shown in FIG. 2, the motor 10 includes one stator 12 and an annular rotor 14 rotatably arranged with respect to the stator 12. The rotor 14 is arranged outside the stator 12 with a gap with respect to the stator 12 in the radial direction of the rotor 14 and the stator 12. The rotor 14 includes a plurality of permanent magnets 14a and a back yoke 14b made of a soft magnetic material connecting the permanent magnets 14a. In this embodiment, the rotor 14 includes 10 permanent magnets 14a. Each of these permanent magnets 14a has the same shape as each other and constitutes one magnetic pole. The permanent magnets 14a are magnetized in the radial direction of the rotor 14, and the polarities of the permanent magnets 14a adjacent to each other in the circumferential direction are different from each other. That is, the permanent magnets 14a are arranged so that the S poles and the N poles appear alternately. In FIG. 2, the arrow portion of the arrow on the permanent magnet 14a indicates the north pole.

The stator 12 includes a plurality of teeth 12a, specifically, 12 teeth 12a. As a result, 12 slots 12b are formed in the stator 12. The 12 teeth 12a are arranged at equal pitches in the circumferential direction of the stator 12 via the slots 12b. That is, in the present embodiment, the motor 10 having the pole logarithm P of "5" and the slot number S of "12" is adopted.

Returning to the description of FIG. 1, the control device 30 is mainly composed of a microcomputer, and operates the inverter 20 to control the control amount of the motor 10 to the command value thereof. In the present embodiment, the control amount is the rotation angular velocity of the motor 10, and the command value is the command angular velocity ωtgt.

The control device 30 generates upper and lower arm operation signals gUp, gUn, gVp, gVn, gWp, and gWn in order to operate the upper and lower arm switches SUp, SUn, SVp, SVn, SWp, and SWn constituting the inverter 20. Then, the output is output to the upper and lower arm switches SUp, SUn, SVp, SVn, SWp, and SWn. Here, the upper arm operation signal and the corresponding lower arm operation signal are complementary signals to each other. That is, the upper arm switch and the lower arm switch connected in series to the upper arm switch are turned on alternately. Incidentally, the command angular velocity ωtgt is provided outside the control device 30 in the vehicle, for example, and is output from an external device higher than the control device 30.

A detection signal of the magnetic flux detection unit for detecting the magnetic pole position of the rotor 14 is input to the control device 30. In the present embodiment, the magnetic flux detection unit includes first, second, and third Hall elements 42a, 42b, and 42c as magnetic sensors. The Hall elements 42a, 42b, and 42c are arranged at positions shifted by 60 ° with respect to the mechanical angle of the motor 10. Further, the Hall elements 42a, 42b, and 42c are arranged at positions where the leakage flux can be detected among the main magnetic flux and the leakage flux from the permanent magnet 14a. In the present embodiment, the motor 10 is housed in a case, and the Hall elements 42a, 42b, and 42c are arranged in the case so that the substrate surfaces of the stator 12 and the rotor 14 face each other. Is implemented. This makes it possible to detect the leakage flux.

Subsequently, the drive control of the motor 10 executed by the control device 30 will be described. The electric angle calculator 30a calculates the rotation angle of the motor 10 based on the detection signals of the Hall elements 42a, 42b, and 42c, and specifically calculates the electric angle θe. In this embodiment, the electric angle calculator 30a corresponds to the rotation angle calculation unit.

The angular velocity calculation unit 30b calculates the rotational angular velocity ωm of the motor 10 based on the electric angle θe calculated by the electric angle calculator 30a. In the present embodiment, the rotational angular velocity ωm is the mechanical angular velocity. The deviation calculation unit 30c calculates the speed deviation Δω by subtracting the rotation angular velocity ωm calculated by the angular velocity calculation unit 30b from the command angular velocity ωtgt.

The fundamental wave voltage setting unit 30d is represented by the following equation (eq1) as an operation amount for feedback-controlling the rotation angular velocity ωm to the command angular velocity ωtgt based on the velocity deviation Δω, the electric angle θe, and the rotation angular velocity ωm. The U, V, W phase fundamental wave voltages VUB, VVB, and VWB in the three-phase fixed coordinate system are set. In the present embodiment, the fundamental wave voltage setting unit 30d calculates the U, V, W phase fundamental wave voltages VUB, VVB, and VWB by proportional integration control based on the velocity deviation Δω. More specifically, the fundamental wave voltages VUB, VVB, and VWB over one period of the electric angle θe are set by the proportional integration control. Here, the electric angular velocity ωe is used to calculate the fluctuation angular velocity of each fundamental wave voltage VUB, VVB, and VWB. The electric angular velocity ωe may be calculated as a value obtained by multiplying the input rotational angular velocity ωm by the pole logarithm P. The fundamental wave voltage setting unit 30d outputs the set fundamental wave voltages VUB, VVB, and VWB in correspondence with the input electric angle θe. The fundamental wave voltages VUB, VVB, and VWB have waveform shapes that are the same as each other and are out of phase with each other by the electric angle θe by “2π / 3”.

高調波電圧設定部３０ｅは、電気角θｅと指令角速度ωｔｇｔとに基づいて、３相固定座標系における第１Ｕ，Ｖ，Ｗ相高調波電圧ＶＵＨ１，ＶＶＨ１，ＶＷＨ１と、第２Ｕ，Ｖ，Ｗ相高調波電圧ＶＵＨ２，ＶＶＨ２，ＶＷＨ２とを算出する。本実施形態において、高調波電圧設定部３０ｅは、記憶部としてのメモリを備えている。メモリは、具体的には例えば不揮発性メモリである。各高調波電圧ＶＵＨ１，ＶＶＨ１，ＶＷＨ１，ＶＵＨ２，ＶＶＨ２，ＶＷＨ２は、指令角速度ωｔｇｔ及び電気角θｅと関係付けられてメモリに記憶されている。高調波電圧設定部３０ｅについては、後に詳述する。

The harmonic voltage setting unit 30e has the first U, V, W phase harmonic voltages VUH1, VVH1, VWH1 and the second U, V, W phases in the three-phase fixed coordinate system based on the electric angle θe and the command angle velocity ωtgt. Harmonic voltages VUH2, VVH2, and VWH2 are calculated. In the present embodiment, the harmonic voltage setting unit 30e includes a memory as a storage unit. The memory is specifically, for example, a non-volatile memory. Each harmonic voltage VUH1, VVH1, VWH1, VUH2, VVH2, VWH2 is stored in the memory in relation to the command angular velocity ωtgt and the electric angle θe. The harmonic voltage setting unit 30e will be described in detail later.

In the present embodiment, the fundamental wave voltage setting unit 30d corresponds to the fundamental wave setting unit, and the harmonic voltage setting unit 30e corresponds to the harmonic wave setting unit.

The first U, V, W phase superimposition portions 30fU, 30fV, 30fW add the first U, V, W phase harmonic voltages VUH1, VVH1, VWH1 to the U, V, W phase fundamental wave voltages VUB, VVB, VWB. .. The second U, V, W phase superimposing portions 30gU, 30gV, 30gW are added to the voltages "VUB + VUH1", "VVB + VVH1", "VWB + VWH1" output from the first U, V, W phase superimposing portions 30fU, 30fV, 30fW, and the second U. , V, W phase harmonic voltages VUH2, VVH2, VWH2 are added. The output values of the second U, V, and W phase superimposing portions 30 gU, 30 gV, and 30 gW are the command voltages VU, VV, and VW of the U, V, and W phases.

The modulator 30h as an operation unit sets each operation signal gUp, gUn, gVp, gVn for setting the output voltage of the U, V, W phase of the inverter 20 to the U, V, W phase command voltage VU, VV, VW. , GWp, gWn are generated. In the present embodiment, each operation signal gUp, gUn, gVp, gVn, gWp, gWn is generated by PWM processing based on the magnitude comparison between each command voltage VU, VV, VW and the carrier signal. As the carrier signal, for example, a triangular wave signal can be used.

By the way, when a current flows through the windings 12U, 12V, 12W of the motor 10 and a rotating magnetic field is generated in the motor 10, a radial electromagnetic force fluctuation acts on the rotor 14. This electromagnetic force is a force that fluctuates at each position in the circumferential direction of the rotor 14, and acts as an attractive force that attracts the rotor 14 toward the stator 12 and a repulsive force that separates the rotor 14 from the stator 12. This electromagnetic force becomes an exciting force that vibrates the rotor 14, which is an elastic body. When the frequency of this electromagnetic force matches the resonance frequency of the ring mode of the rotor 14, the noise (magnetic sound) of the motor 10 may increase. Further, when the frequency of the electromagnetic force matches the resonance frequency, the vibration of the motor 10 increases, so that the vibration of the in-vehicle device mechanically connected to the motor 10 increases, and as a result, the noise from the in-vehicle device is generated. May increase. Hereinafter, the annular mode will be described.

The annular mode is a mode of periodic fluctuation generated in the rotor 14 due to the exciting force applied in the radial direction of the rotor 14. FIG. 3 shows a 0 to 5th order circular mode as an example of the circular mode. FIG. 3 is a schematic view of a vertical cross section of the rotor 14. In FIG. 3, the broken line shows the shape of the rotor 14 in a state where the exciting force is not applied to the rotor 14 (hereinafter referred to as “original shape”), and the solid line shows the exciting force acting on the rotor 14. The shape of the rotor 14 in the state is shown. The alternate long and short dash line is a node line connecting two nodes separated from each other by π in a state where the rotor 14 is displaced by an exciting force acting on the rotor 14. The midpoint between adjacent nodes is the belly. At the knot portion, even if a excitation force acts on the rotor 14, the rotor 14 hardly displaces from its original shape.

The 0th-order annular mode is a mode in which the rotor 14 repeatedly expands and contracts in the radial direction while maintaining a shape similar to the original shape.

The primary ring mode is a mode in which the rotor 14 is displaced with respect to one node line while rotating. Specifically, the primary annular mode is a mode in which one belly extends in the radial direction with respect to the original shape, and one belly is contracted in the radial direction at a distance of π from the extending belly. Is. The secondary ring mode is a mode in which the rotor 14 is displaced with respect to the two node lines while rotating. Specifically, in the quadratic ring mode, two antinodes separated from each other by π with respect to the original shape extend in the radial direction, and are separated by “π / 2” from the two extending antinodes. This is a mode in which the two bellies contract in the radial direction.

The tertiary ring mode is a mode in which the rotor 14 is displaced while rotating with reference to three node lines. Specifically, in the third-order annular mode, the three vents separated in the original shape at intervals of "2π / 3" extend in the radial direction, and "π /" from the three extending vents. This is a mode in which the abdomen of three places separated by 3 ”contracts in the radial direction. The fourth-order annular mode is a mode in which the rotor 14 is displaced with reference to four node lines while rotating. Specifically, in the fourth-order annulus mode, four bellies separated from each other at "π / 2" intervals extend in the radial direction with respect to the original shape, and "π /" from the four extending bellies. This is a mode in which the bellies at four locations separated by 4 ”contract in the radial direction.

The fifth-order annular mode is a mode in which the rotor 14 is displaced with reference to five node lines while rotating. Specifically, in the fifth-order annulus mode, five bellies separated from the original shape at intervals of "2π / 5" extend in the radial direction, and "π /" from the extending five bellies. This is a mode in which the abdomen of 5 places separated by 5 "contracts in the radial direction.

Assuming that X is a natural number, the exciting force that causes the X-order annular mode is a force at which the angular distance between the point where the suction force increases and the point where the suction force decreases is "π / X".

Each of these ring modes has a unique resonance frequency (resonance angular velocity). Then, the frequency of the exciting force that causes each annular mode becomes close to the resonance frequency of each annular mode, so that the resonance phenomenon of the rotor 14 occurs. When the actual frequency of the exciting force is close to the resonance frequency, the magnetic sound of the motor 10 increases, causing problems such as an increase in the noise level in the audible frequency band.

In order to deal with such a problem, in the present embodiment, the control device 30 is provided with a harmonic voltage setting unit 30e. The harmonic voltage setting unit 30e contains first U, V, W phase harmonic voltages VUH1, VVH1, VWH1 and second U, V, W for suppressing radial electromagnetic force fluctuations that cause magnetic noise. The phase harmonic voltages VUH2, VVH2, and VWH2 are stored. Hereinafter, the harmonic voltage for suppressing the fluctuation of the electromagnetic force will be described.

The fundamental wave currents IUB, IVB, and IWB of each phase are expressed by the following equation (eq2). These fundamental wave currents IUB, IVB, and IWB have waveforms that are "2π / 3" out of phase with each other at an electric angle θe.

以下、Ｕ相を例にして説明する。モータ１０を基本波電流ＩＵＢ，ＩＶＢ，ＩＷＢによって運転した場合の結果から、騒音抑制のためには、電磁力の６次成分を低減させることが要求される。６次成分とは、巻線１２Ｕ，１２Ｖ，１２Ｗに流れる基本波電流の６倍の変動角速度を有する電磁力のことである。ここで、モータ１０の径方向の電磁力（節点力）をＦで表す。基本波電流を流すことにより、電磁力Ｆが発生するのであるから、電磁力Ｆは下式（ｅｑ３）によって表すことができる。

Hereinafter, the U phase will be described as an example. From the results when the motor 10 is operated by the fundamental wave currents IUB, IVB, and IWB, it is required to reduce the sixth component of the electromagnetic force in order to suppress noise. The sixth-order component is an electromagnetic force having a fluctuating angular velocity six times that of the fundamental wave current flowing through the windings 12U, 12V, and 12W. Here, the radial electromagnetic force (node force) of the motor 10 is represented by F. Since the electromagnetic force F is generated by passing the fundamental wave current, the electromagnetic force F can be expressed by the following equation (eq3).

電磁力Ｆの主要成分は、２次，４次，６次等の偶数次数の電磁力であることが知られている。このため、上式（ｅｑ３）におけるＧは、奇数次数の周期関数として下式（ｅｑ４）によって表すことができる。

It is known that the main component of the electromagnetic force F is an even-order electromagnetic force such as a second-order, fourth-order, or sixth-order. Therefore, G in the above equation (eq3) can be expressed by the following equation (eq4) as a periodic function of odd-order order.

ここで、通常、基本波電流を流すことにより大きな平均トルクが得られるようにモータが設計される。このことから、Ｇについて、低次のものほど係数を大きな値に設定する。このため、本実施形態では、上式（ｅｑ４）において、「ｎ＝１」とする。ここで、高調波電流を下式（ｅｑ５）で表す。

Here, the motor is usually designed so that a large average torque can be obtained by passing a fundamental wave current. For this reason, for G, the lower the order, the larger the coefficient is set. Therefore, in the present embodiment, “n = 1” is set in the above equation (eq4). Here, the harmonic current is expressed by the following equation (eq5).

上式（ｅｑ５）において、βは２以上の整数である。上式（ｅｑ４），（ｅｑ５）を上式（ｅｑ３）に代入することにより、高調波電磁力ＦＨを下式（ｅｑ６）によって表すことができる。

In the above equation (eq5), β is an integer of 2 or more. By substituting the above equations (eq4) and (eq5) into the above equation (eq3), the harmonic electromagnetic force FH can be expressed by the following equation (eq6).

ここで、「β＝６Ｍ−１」（Ｍは０以上の整数）とする場合、上式（ｅｑ６）は下式（ｅｑ７）となる。

Here, when "β = 6M-1" (M is an integer of 0 or more), the above equation (eq6) becomes the following equation (eq7).

上式（ｅｑ５），（ｅｑ７）は、「６Ｍ−１」次の高調波電流を巻線１２Ｕ，１２Ｖ，１２Ｗに流すと、「６Ｍ」次の電磁力と、「６Ｍ−２」次の電磁力とがロータ１４に作用することを表している。ここで、「６Ｍ」次の電磁力，高調波電流とは、「６Ｍ」と基本角速度との乗算値を変動角速度とする電磁力，高調波電流のことである。基本角速度とは、巻線１２Ｕ，１２Ｖ，１２Ｗに流す基本波電流の変動角速度ωｅのことである。上式（ｅｑ５），（ｅｑ７）は、「６Ｍ−１」次の高調波電流を巻線１２Ｕ，１２Ｖ，１２Ｗに流すことにより、「６Ｍ」次，「６Ｍ−２」次の電磁力を制御できることを表している。本実施形態では、「６Ｍ−２」次の電磁力を低減するように、上式（ｅｑ７）の係数ｅ，ｆを調整する。ただし、この調整により、「６Ｍ」次の電磁力が増大する。

In the above equations (eq5) and (eq7), when a harmonic current of the "6M-1" order is passed through the windings 12U, 12V, 12W, the electromagnetic force of the "6M" order and the electromagnetic force of the "6M-2" order are applied. It shows that the force acts on the rotor 14. Here, the electromagnetic force and harmonic current of the "6M" order are the electromagnetic force and harmonic current whose fluctuating angular velocity is the product of "6M" and the basic angular velocity. The basic angular velocity is the fluctuating angular velocity ωe of the fundamental wave current flowing through the windings 12U, 12V, and 12W. In the above equations (eq5) and (eq7), the electromagnetic force of the "6M-1" order and the "6M-2" order is controlled by passing the harmonic current of the "6M-1" order through the windings 12U, 12V, 12W. It shows that you can do it. In the present embodiment, the coefficients e and f of the above equation (eq7) are adjusted so as to reduce the electromagnetic force of the "6M-2" order. However, this adjustment increases the electromagnetic force of the "6M" order.

On the other hand, when "β = 6M + 1", the above equation (eq6) becomes the following equation (eq8).

上式（ｅｑ５），（ｅｑ８）は、「６Ｍ＋１」次の高調波電流を巻線１２Ｕ，１２Ｖ，１２Ｗに流すと、「６Ｍ」次の電磁力と、「６Ｍ＋２」次の電磁力とがロータ１４に作用することを表している。すなわち、上式（ｅｑ５），（ｅｑ８）は、「６Ｍ＋１」次の高調波電流を巻線１２Ｕ，１２Ｖ，１２Ｗに流すことにより、「６Ｍ」次，「６Ｍ＋２」次の電磁力を制御できることを表している。本実施形態では、「６Ｍ−１」次の高調波電流の重畳によって増大する「６Ｍ」次の電磁力を低減するように、上式（ｅｑ８）の係数ｅ，ｆを調整する。ただし、この調整により、「６Ｍ＋２」次の電磁力が増大する。

In the above equations (eq5) and (eq8), when a harmonic current of the "6M + 1" order is passed through the windings 12U, 12V, 12W, the "6M" order electromagnetic force and the "6M + 2" order electromagnetic force become a rotor. It shows that it acts on 14. That is, the above equations (eq5) and (eq8) can control the "6M" order and "6M + 2" order electromagnetic forces by passing the "6M + 1" order harmonic current through the windings 12U, 12V, 12W. Represents. In the present embodiment, the coefficients e and f of the above equation (eq8) are adjusted so as to reduce the “6M” -order electromagnetic force that increases due to the superposition of the “6M-1” -order harmonic current. However, this adjustment increases the electromagnetic force of the "6M + 2" order.

In the present embodiment, noise increases when the fluctuation angular velocity of the 10th-order electromagnetic force is close to the resonance angular velocity corresponding to the second-order annular mode. Therefore, based on the above-mentioned matters, "M = 2" is set, and the 10th-order electromagnetic force that causes noise is converted into the 12th-order electromagnetic force by superimposing the 11th-order harmonic current. Here, in the present embodiment, the fluctuation angular velocity of the twelfth-order electromagnetic force is also close to the resonance angular velocity. Therefore, the 12th-order electromagnetic force is further converted into the 14th-order electromagnetic force by superimposing the 13th-order harmonic current. The fluctuation angular velocity of the 14th-order electromagnetic force is a value sufficiently separated from the resonance angular velocity. Therefore, noise can be reduced.

Therefore, in the present embodiment, among the electromagnetic forces acting on the rotor 14, the 11th-order harmonic current (hereinafter referred to as "first harmonic current IUH1, IVH1, IWH1") capable of reducing the 10th-order electromagnetic force is used. , The 13th harmonic current (hereinafter referred to as "2nd harmonic current IUH2, IVH2, IWH2") that can reduce the 12th order electromagnetic force that increases due to the superimposition of the 11th harmonic current is the fundamental wave current. It is superimposed on IUB, IVB, and IWB. FIG. 4 illustrates the transition of the current when the U-phase harmonic currents IUH1 and IUH2 are superimposed on the fundamental wave current IUB. The following equation (eq9) shows the U-phase first harmonic current IUH1, and the following equation (eq10) shows the U-phase second harmonic current IUH2.

第１高調波電流ＩＵＨ１，ＩＶＨ１，ＩＷＨ１は、１０次の電磁力を低減させるように、上式（ｅｑ９）に示した高調波電流の位相及び振幅（各係数ｅ１，ｆ１）が調整された波形となっている。Ｕ，Ｖ，Ｗ相の高調波電流ＩＵＨ１，ＩＶＨ１，ＩＷＨ１は、波形形状が互いに同一であってかつ、電気角θｅで位相が互いに「２π／３」ずれた波形となっている。第２高調波電流ＩＵＨ２，ＩＶＨ２，ＩＷＨ２についても同様である。

The first harmonic currents IUH1, IVH1, IWH1 are waveforms in which the phase and amplitude (each coefficient e1, f1) of the harmonic current shown in the above equation (eq9) are adjusted so as to reduce the 10th-order electromagnetic force. It has become. The U, V, and W phase harmonic currents IUH1, IVH1, and IWH1 have waveform shapes that are the same as each other and that are out of phase with each other at an electric angle θe by "2π / 3". The same applies to the second harmonic currents IUH2, IVH2, and IWH2.

The first harmonic voltage VUH1, VVH1, VWH1 for passing the first harmonic currents IUH1, IVH1, IWH1 through the windings 12U, 12V, 12W and the second harmonic currents IUH2, IVH2, IWH2 are wound 12U, 12V. , The second harmonic voltage VUH2, VVH2, VWH2 for flowing to 12W is stored in advance in the memory of the harmonic voltage setting unit 30e. Specifically, as shown in the following equations (eq11) and (eq12), the matched 11th and 13th order harmonic voltages VUH1, VVH1, and VVW1 are stored in advance in the harmonic voltage setting unit 30e. Here, the conversion from the harmonic current to the harmonic voltage can be performed, for example, based on a well-known voltage equation relating the phase voltage applied to the motor 10 to the phase current.

上式（ｅｑ１１），（ｅｑ１２）におけるγ，δは、上式（ｅｑ１）の基本波電圧に対する位相差を示す。本実施形態では、上式（ｅｑ１１）におけるＶ１１，γを第１振幅，第１位相差と称すこととし、上式（ｅｑ１２）におけるＶ１３，Δを第２振幅，第２位相差と称すこととする。Ｕ，Ｖ，Ｗ相の第１高調波電圧ＶＵＨ１，ＶＶＨ１，ＶＷＨ１は、１０次の電磁力を低減させるように設定されている。具体的には、第１高調波電圧ＶＵＨ１，ＶＶＨ１，ＶＷＨ１は、１０次の電磁力を低減させるように、上式（ｅｑ１１）に示した第１位相差γ及び第１振幅Ｖ１１が調整された波形となっている。また、本実施形態において、Ｕ，Ｖ，Ｗ相の第１高調波電圧ＶＵＨ１，ＶＶＨ１，ＶＷＨ１は、波形形状が互いに同一であってかつ、電気角θｅで位相が互いに「２π／３」ずれた波形となっている。また、Ｕ，Ｖ，Ｗ相の第２高調波電圧ＶＵＨ２，ＶＶＨ２，ＶＷＨ２は、１３次の電磁力を低減させるように設定されている。具体的には、第２高調波電圧ＶＵＨ２，ＶＶＨ２，ＶＷＨ２は、１４次の電磁力を低減させるように、上式（ｅｑ１２）に示した第２位相差δ及び第２振幅Ｖ１３が調整された波形となっている。また、本実施形態において、Ｕ，Ｖ，Ｗ相の第２高調波電圧ＶＵＨ２，ＶＶＨ２，ＶＷＨ２は、波形形状が互いに同一であってかつ、電気角θｅで位相が互いに「２π／３」ずれた波形となっている。

Γ and δ in the above equations (eq11) and (eq12) indicate the phase difference with respect to the fundamental wave voltage of the above equation (eq1). In the present embodiment, V11 and γ in the above equation (eq11) are referred to as the first amplitude and the first phase difference, and V13 and Δ in the above equation (eq12) are referred to as the second amplitude and the second phase difference. To do. The first harmonic voltages VUH1, VVH1, and VWH1 of the U, V, and W phases are set to reduce the 10th-order electromagnetic force. Specifically, in the first harmonic voltage VUH1, VVH1, VWH1, the first phase difference γ and the first amplitude V11 shown in the above equation (eq11) are adjusted so as to reduce the 10th-order electromagnetic force. It is a waveform. Further, in the present embodiment, the first harmonic voltages VUH1, VVH1, and VWH1 of the U, V, and W phases have the same waveform shape and are out of phase with each other by "2π / 3" at the electric angle θe. It is a waveform. The U, V, and W phase second harmonic voltages VUH2, VVH2, and VWH2 are set to reduce the 13th-order electromagnetic force. Specifically, in the second harmonic voltage VUH2, VVH2, VWH2, the second phase difference δ and the second amplitude V13 shown in the above equation (eq12) are adjusted so as to reduce the 14th-order electromagnetic force. It is a waveform. Further, in the present embodiment, the second harmonic voltages VUH2, VVH2, and VWH2 of the U, V, and W phases have the same waveform shape and are out of phase with each other by "2π / 3" at the electric angle θe. It is a waveform.

In the present embodiment, the first and second harmonic voltages are stored as map data in the harmonic voltage setting unit 30e in relation to the command angular velocity ωtgt and the electric angle θe. The harmonic voltage setting unit 30e selects each corresponding harmonic voltage based on the command angular velocity ωtgt and the electric angle θe input each time (for example, for each control cycle of the control device 30), and each superimposing unit 30fU. Output to ~ 30fW, 30gU ~ 30gW. As a result, the first and second harmonic currents can be superimposed on the fundamental wave current.

According to such a configuration, the first and second harmonic voltages are superimposed on the fundamental wave voltage when the actual rotation angular velocity ωm approaches the resonance angular velocity. Then, when the actual rotation angular velocity ωm deviates from the resonance angular velocity, the first and second harmonic voltages superimposed on the fundamental wave voltage become smaller or become zero.

By the way, due to the individual difference of the motor 10, the magnetic flux characteristics of the permanent magnet 14a are different for each of the mass-produced motors 10. In this case, the harmonic voltage stored in the memory may deviate from the appropriate harmonic voltage for reducing the 11th and 13th order electromagnetic force fluctuations. In this case, there is a concern that noise will increase due to an increase in torque fluctuation components that do not contribute to output torque.

Therefore, the control device 30 includes the correction value calculation unit 30i shown in FIG. The correction value calculation unit 30i performs a correction process for correcting the first and second harmonic voltages stored in the memory. The correction process will be described below.

The radial electromagnetic force F of the rotor 14 is proportional to the magnetic flux φm from the permanent magnet of the rotor and the current I flowing through the stator winding, as shown in the following equation (eq13).

ここで、ロータ１４が一定速度で回転している場合において、磁束φｍのうち、正弦波で表される磁束を正弦波磁束φ０とし、正弦波磁束φ０からの歪み成分を磁束歪みΔφとする。また、電流Ｉのうち、正弦波で表される電流を正弦波電流Ｉ０とし、正弦波電流Ｉ０からの歪み成分を電流歪みΔＩとする。この場合、上式（ｅｑ１３）が下式（ｅｑ１４）で表される。

Here, when the rotor 14 is rotating at a constant speed, of the magnetic flux φm, the magnetic flux represented by the sine wave is defined as the sine wave magnetic flux φ0, and the distortion component from the sine wave magnetic flux φ0 is defined as the magnetic flux strain Δφ. Further, among the currents I, the current represented by the sine wave is defined as the sine wave current I0, and the distortion component from the sine wave current I0 is defined as the current strain ΔI. In this case, the above equation (eq13) is expressed by the following equation (eq14).

上式（ｅｑ１４）において、「Δφ×ΔＩ」は他の項と比較して非常に小さいため、無視している。上式（ｅｑ１４）において、右辺第１項の「φ０×Ｉ０」はモータ１０の出力トルクを示し、右辺第２項の「Δφ×Ｉ０＋φ０×ΔＩ」は、磁気音を生じさせる加振力を示す。第１，第２高調波電流を流すことにより、右辺第２項の加振力を０とすることができれば、磁気音が低減できる。ただし、この加振力において、磁束歪みΔφがモータ１０の個体差によりばらつく。このため、磁束歪みΔφを観測し、観測した磁束歪みΔφに基づいて、下式（ｅｑ１５）を満たす電流歪みΔＩを高調波電流として巻線に流すことにより、加振力を低減できる。

In the above equation (eq14), “Δφ × ΔI” is ignored because it is very small compared to other terms. In the above equation (eq14), “φ0 × I0” in the first term on the right side indicates the output torque of the motor 10, and “Δφ × I0 + φ0 × ΔI” in the second term on the right side indicates an exciting force that generates a magnetic sound. .. If the exciting force of the second term on the right side can be set to 0 by passing the first and second harmonic currents, the magnetic sound can be reduced. However, in this exciting force, the magnetic flux strain Δφ varies depending on the individual difference of the motor 10. Therefore, the exciting force can be reduced by observing the magnetic flux strain Δφ and passing the current strain ΔI satisfying the following equation (eq15) through the winding as a harmonic current based on the observed magnetic flux strain Δφ.

ちなみに、本実施形態において低減対象とする電磁力変動の次数は、永久磁石の磁束変動の次数と同じである。つまり、マクスウェルの応力の式により、ロータの径方向の電磁力Ｆｅは、下式（ｅｑ１６）により表される。

Incidentally, the order of the electromagnetic force fluctuation to be reduced in the present embodiment is the same as the order of the magnetic flux fluctuation of the permanent magnet. That is, according to Maxwell's stress equation, the electromagnetic force Fe in the radial direction of the rotor is expressed by the following equation (eq16).

上式（ｅｑ１６）において、μは透磁率を示し、Ｂは磁束密度を示し、Φはティース１２ａと永久磁石１４ａとの間の磁束を示す。ここで磁束Φは、下式（ｅｑ１７）に示すように、永久磁石１４ａの磁石磁束φｍと、巻線１２Ｕ，１２Ｖ，１２Ｗに電流が流れることにより発生する磁束φｅとからなる。

In the above equation (eq16), μ indicates the magnetic permeability, B indicates the magnetic flux density, and Φ indicates the magnetic flux between the teeth 12a and the permanent magnet 14a. Here, the magnetic flux Φ is composed of the magnet magnetic flux φm of the permanent magnet 14a and the magnetic flux φe generated by the current flowing through the windings 12U, 12V, and 12W, as shown in the following equation (eq17).

上式（ｅｑ１７）において、Ｌは各巻線１２Ｕ〜１２Ｗのインダクタンスを示し、Ｉは各巻線１２Ｕ〜１２Ｗに流れる電流を示す。上式（ｅｑ１７）を上式（ｅｑ１６）に代入すると、下式（ｅｑ１８）が導かれる。

In the above equation (eq17), L indicates the inductance of each winding 12U to 12W, and I indicates the current flowing through each winding 12U to 12W. Substituting the above equation (eq17) into the above equation (eq16) leads to the following equation (eq18).

上式（ｅｑ１８）において、右辺第２項が磁気音を生じさせる加振力を示す。すなわち、本実施形態では、磁石磁束φｍを観測して電磁力Ｆｅを低減するため、観測対象となる磁石磁束φの変動角速度と、低減対象とする電磁力Ｆｅの変動角速度とが同じとなる。

In the above equation (eq18), the second term on the right side indicates the exciting force that produces a magnetic sound. That is, in the present embodiment, since the magnetic flux φm is observed to reduce the electromagnetic force Fe, the fluctuation angular velocity of the magnet magnetic flux φ to be observed and the fluctuation angular velocity of the electromagnetic force Fe to be reduced are the same.

On the other hand, in the technique described in Patent Document 1, a search coil for detecting magnetic flux is provided in the teeth. In this configuration, the magnetic flux detected by the search coil is the total magnetic flux Φ, which is the sum of the magnetic flux φm of the magnet and the magnetic flux φe generated by the flow of the current. Therefore, in the technique described in Patent Document 1, as shown in the above equation (eq16), the angular velocity twice the fluctuating angular velocity of the magnet magnetic flux φ to be observed is the fluctuating angular velocity of the electromagnetic force Fe to be reduced. Become. Therefore, the fluctuation angular velocity of the electromagnetic force to be reduced in the present embodiment is different from the fluctuation angular velocity of the electromagnetic force to be reduced in Patent Document 1.

FIG. 5 shows a procedure of correction processing executed by the correction value calculation unit 30i. This process is executed at the time of pre-shipment inspection of the control device 30 in the factory or at the time of starting the motor 10.

According to this series of processes, first, in step S10, it is determined whether or not the rotor 14 is rotating.

If an affirmative determination is made in step S10, the process proceeds to step S12, and the energization of the windings 12U, 12V, and 12W is temporarily stopped. This process is a process for making it easy to grasp the magnetic flux of the permanent magnet 14a and avoiding an increase in the arithmetic processing. That is, when the windings 12U, 12V, 12W are energized, magnetic flux is generated by the current flowing through the windings 12U, 12V, 12W. Therefore, each Hall element 42a to 42c detects the total of the magnetic flux generated by the current and the magnetic flux of the magnet. In this case, in order to correct the first and second harmonic voltages, it is necessary to subtract the magnetic flux generated by the current from the detected magnetic flux. As a result, the arithmetic processing in the correction processing increases. On the other hand, the magnetic flux detection value of each Hall element 42a to 42c during the period when the energization of the windings 12U, 12V, and 12W is stopped is only the magnetic flux, and it is necessary to subtract the magnetic flux generated by the current from the magnetic flux detection value. There is no. In this embodiment, the process of this step corresponds to the energization stop portion.

In the following step S14, the magnetic flux detected by the Hall elements 42a to 42c is acquired in relation to the electric angle θe. By grasping the magnetic flux of the magnet in relation to the electric angle θe, it is possible to appropriately grasp the magnetic flux characteristics with the electric angle θe as the independent variable and the magnet magnetic flux as the dependent variable. The magnetic flux detection value used in this step may be at least one magnetic flux detection value among the Hall elements 42a to 42c.

In the following step S16, the spatial distribution of the magnetic flux, which is the magnetic flux detection value related to the electric angle θe, is calculated based on the magnetic flux detection value acquired in step S14. In this embodiment, the processes of steps S14 and S16 correspond to the magnetic flux information acquisition unit.

In the following step S18, the amplitude and phase difference of the 11th and 13th order magnetic fluxes to be reduced are extracted based on the calculated spatial distribution of the magnetic flux. Here, the phase difference of the magnetic flux may be, for example, the phase difference with respect to the fundamental wave voltage represented by the above equation (eq1) or the phase difference with respect to the reference electric angle of the motor 10. Then, in step S18, the first amplitude correction value for correcting the first amplitude V11 and the first phase correction for correcting the first phase difference γ are based on the extracted amplitude and phase difference of the 11th order magnetic flux. Calculate the value. Specifically, the first amplitude correction value and the first amplitude correction value are based on the calculated amplitude and phase difference of the 11th order magnetic flux and the amplitude and phase difference of the 11th order magnetic flux stored in the memory in advance and as a reference. Calculate the phase correction value. Further, in step S18, the second amplitude correction value for correcting the second amplitude V13 and the second phase correction for correcting the second phase difference δ are based on the extracted amplitude and phase difference of the 13th order magnetic flux. Calculate the value. Specifically, the second amplitude correction value and the second amplitude correction value and the second amplitude correction value are based on the calculated amplitude and phase difference of the 13th order magnetic flux and the amplitude and phase difference of the 13th order magnetic flux stored in the memory in advance and as a reference. Calculate the phase correction value.

Then, in step S18, the first and second amplitudes V11 and V13 are corrected based on the calculated first and second amplitude correction values, and the first and first phases are corrected based on the calculated first and second phase correction values. 2 Correct the phase difference γ and δ. Specifically, for example, to explain the correction of the first harmonic as an example, the first amplitude V11 is corrected by multiplying the first amplitude correction value by the first amplitude V11, and the first phase correction value is set to the first position. The first phase difference γ may be corrected by adding to the phase difference γ.

After the processing in step S18 is completed, the energization of the windings 12U, 12V, and 12W is resumed. After that, the harmonic voltage setting unit 30e sets the first and second harmonic voltages in which the amplitude and the phase difference are corrected.

According to the present embodiment described above, the following effects can be obtained.

The first amplitude V11 and the first phase difference γ of the first harmonic voltage were corrected based on the magnet magnetic flux component which fluctuates with the fluctuation angular velocity of the first harmonic voltage. Further, the second amplitude V13 and the second phase difference δ of the second harmonic voltage were corrected based on the magnet magnetic flux component fluctuating with the fluctuation angular velocity of the second harmonic voltage. As a result, even if there are individual differences in the motor 10, it is possible to reduce the radial electromagnetic force fluctuation acting on the rotor 14.

The first amplitude V11 and the first phase difference γ stored in the memory were corrected based on the magnet magnetic flux component which fluctuates with the fluctuation angular velocity of the first harmonic voltage. Further, the second amplitude V13 and the second phase difference δ stored in the memory were corrected based on the magnet magnetic flux component that fluctuates with the fluctuation angular velocity of the second harmonic voltage. According to this configuration, it is not necessary to generate the first and second harmonic voltages reflecting the individual differences of the motor 10 from 0, so that the control device 30 for setting the first and second harmonic voltages The calculation load can be reduced.

Based on the magnet magnetic flux φm grasped in relation to the electric angle θe, the magnetic flux magnetic flux component fluctuating at the fluctuating angular velocity of the electromagnetic force to be reduced was extracted. As a result, the actual magnet magnetic flux distribution can be properly grasped, and the correction accuracy of the first and second harmonic voltages can be improved.

The magnetic flux information detected by the Hall elements 42a to 42c was diverted for the calculation of the electric angle θe used for the motor control and for the correction of the first and second harmonic voltages. Therefore, the first and second harmonic voltages can be corrected by reflecting the individual differences of the motor 10 without adding a dedicated magnetic flux detection unit for correcting the first and second harmonic voltages.

The magnetic flux of the magnetic pole was acquired during the period when the energization of the windings 12U, 12V, and 12W was stopped. As a result, the magnetic flux component of the magnetic pole can be easily extracted without using an external device provided outside the control device 30, and an increase in the calculation load of the control device 30 can be avoided.

Correction processing was performed when the motor 10 was started. Therefore, the first and second harmonic voltages can be corrected by reflecting the influence of the aged deterioration of the motor 10. The correction process may be performed every time the motor 10 is started, or may be performed when the motor 10 is started when a predetermined period has elapsed since the correction process was performed last time.

<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings, focusing on the differences from the first embodiment. In the first embodiment, the harmonic voltages VUH1 to VWH1 and VUH2 to VWH2 are stored in advance in the memory, and the harmonic currents are superimposed based on the stored harmonic voltages VUH1 to VWH1 and VUH2 to VWH2. It was configured. In the present embodiment, the harmonic voltages VUH1 to VWH1 and VUH2 to VWH2 are not stored in the memory in advance. Therefore, the control device 30 includes a harmonic generation unit 30j instead of the harmonic voltage setting unit 30e and the correction value calculation unit 30i, as shown in FIG. In FIG. 6, the same reference numerals are given to the same configurations as those shown in FIG. 1 above for convenience.

The harmonic generation unit 30j generates each harmonic voltage VUH1 to VWH1 and VUH2 to VWH2 based on at least one magnetic flux detection value of each Hall element 42a to 42c.

FIG. 7 shows a procedure of the harmonic voltage generation process executed by the harmonic generation unit 30j. This process is executed at the time of pre-shipment inspection of the control device 30 in the factory, at the time of starting the motor 10 after the control device 30 is shipped from the factory, and at the time of driving the motor 10 after the motor 10 is started. In FIG. 7, the same step numbers are assigned to the same processes as those shown in FIG. 5 above for convenience.

In this series of processes, after the process of step S16 is completed, the process proceeds to step S20. In step S20, the amplitude and phase difference of the 11th and 13th order magnetic fluxes to be reduced are extracted based on the spatial distribution of the magnetic flux calculated in step S16. Here, the phase difference of the magnetic flux may be, for example, the phase difference with respect to the fundamental wave voltage represented by the above equation (eq1) or the phase difference with respect to the reference electric angle of the motor 10.

Then, in step S20, based on the amplitude and phase difference of the extracted 11th and 13th order magnetic fluxes, the first U, V, W phase harmonic voltages VUH1, VVH1, VWH1 are used without using the harmonic voltage stored in the memory. And the second U, V, W phase harmonic voltages VUH2, VVH2, VWH2 are generated.

Specifically, the first order is based on the amplitude and phase difference of the extracted 11th order magnetic flux so that the spatial distribution of the magnetic flux that can shift the frequency of the current electromagnetic force from the resonance frequency of the annular mode of the rotor 14 is realized. The amplitude and phase of the first harmonic currents IUH1, IVH1 and IWH1 are calculated, and the amplitude and phase of the second harmonic currents IUH2, IVH2 and IWH2 are calculated based on the amplitude and phase difference of the extracted 13th-order magnetic flux. Here, the spatial distribution of the magnetic flux is the distribution of the magnitude of the magnetic flux associated with the mechanical angle θm of the motor 10. In the present embodiment, the noise increases when the frequency of the electromagnetic force is close to the resonance frequency corresponding to the secondary ring mode. Therefore, in the present embodiment, as shown in FIG. 8, in order to realize a spatial distribution of magnetic flux that can shift the frequency of the current electromagnetic force from the resonance frequency of the ring mode of the rotor 14, a quadratic ring The amplitude and phase of the first harmonic currents IUH1, IVH1, IWH1 and the second harmonic current IUH2, such that the spatial distribution of the magnetic flux corresponding to the mode is changed to the spatial distribution of the magnetic flux corresponding to the third-order annular mode. Calculate the amplitude and phase of IVH2 and IWH2. In FIG. 8, for convenience, the amplitude of the magnetic flux corresponding to the second-order annular mode and the amplitude of the magnetic flux corresponding to the third-order annular mode are shown as the same value.

Then, the first amplitude V11 and the first phase difference γ of the first harmonic voltages VUH1, VVH1, VWH1 are calculated based on the calculated amplitude and phase of the first harmonic currents IUH1, IVH1, IWH1. Further, the second amplitude V13 and the second phase difference δ of the second harmonic voltage VUH2, VVH2, VWH2 are calculated based on the calculated amplitude and phase of the second harmonic currents IUH2, IVH2, IWH2. Here, the conversion from the harmonic current to the harmonic voltage can be performed, for example, based on a well-known voltage equation relating the phase voltage applied to the motor 10 to the phase current.

After the processing in step S20 is completed, the energization of the windings 12U, 12V, and 12W is resumed. After that, the harmonic generation unit 30j generates and outputs the first and second harmonic voltages.

Specifically, the harmonic generation unit 30j calculates the electric angular velocity ωe based on the rotation angular velocity ωm calculated by the angular velocity calculation unit 30b. The harmonic generation unit 30j generates the first harmonic voltages VUH1, VVH1, VWH1 in accordance with the calculated electric angular velocity ωe and the input electric angle θe based on the calculated first amplitude V11 and the first phase difference γ. Output. Further, the harmonic generation unit 30j has a second harmonic voltage VUH2, VVH2, corresponding to the calculated electric angular velocity ωe and the input electric angle θe based on the calculated second amplitude V13 and the second phase difference δ. Output VWH2.

In FIG. 9, the above-mentioned harmonic voltages VUH1, VVH1, VWH1, VUH2, VVH2, and VWH2 are superposed, so that the spatial distribution of the magnetic flux corresponding to the second-order annular mode corresponds to the third-order annular mode. An example is shown in which the spatial distribution of the magnetic flux is changed. By changing the spatial distribution of the magnetic flux, the frequency component of the electromagnetic force is shifted from the frequency corresponding to the second-order annular mode to the frequency corresponding to the third-order annular mode. As a result, even if there are individual differences in the motor 10, it is possible to reduce the radial electromagnetic force fluctuation acting on the rotor 14.

Further, according to the present embodiment, it is possible to eliminate the work of matching each harmonic voltage stored in the memory, so that the labor required when designing the control device 30 can be reduced.

<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings, focusing on the differences from the first embodiment. In the first embodiment, the harmonic voltages VUH1 to VWH1 and VUH2 to VWH2 in the fixed coordinate system of the motor 10 are set. In the present embodiment, the harmonic voltages VUH1 to VWH1 and VUH2 to VWH2 in the dq coordinate system, which is the two-phase rotating coordinate system of the motor 10, are set.

FIG. 10 shows a system configuration diagram according to the present embodiment. In FIG. 10, the same components as those shown in FIG. 1 above are designated by the same reference numerals for convenience.

As shown in FIG. 10, the control system includes a current detection unit 21. The current detection unit 21 detects at least two phases of the currents of each phase flowing through the motor 10. In this embodiment, the current detection unit 21 detects U, V, and W phase currents. The detected value of the current detection unit 21 is input to the control device 30.

Subsequently, the drive control of the motor 10 executed by the control device 30 will be described.

The two-phase conversion unit 30k calculates the U, V, W phase currents in the three-phase fixed coordinate system based on the phase current detected by the current detection unit 21 and the electric angle θe calculated by the electric angle calculator 30a. It is converted into a d-axis current Idr and a q-axis current Iqr in the dq coordinate system.

The command value setting unit 30m sets the d-axis command current Id * and the q-axis command current Iq *, which are the current command values in the two-phase rotating coordinate system, based on the command angular velocity ωtgt. The d-axis command current Id * and the q-axis command current Iq * are set using, for example, map information in which the command angular velocity ωtgt and the d-axis command current Id * and the q-axis command current Iq * are related. Just do it.

The fundamental wave current control unit 30n calculates the d-axis fundamental wave voltage VdB, which is a d-axis voltage component of the motor 10, as an operation amount for feedback-controlling the d-axis current Idr to the d-axis command current Id *. In this embodiment, the d-axis fundamental wave voltage VdB is a DC component in the dq coordinate system. Further, the fundamental wave current control unit 30n calculates the q-axis fundamental wave voltage VqB, which is a q-axis voltage component of the motor 10, as an operation amount for feedback-controlling the q-axis current Iqr to the q-axis command current Iq *. In this embodiment, the q-axis fundamental wave voltage VqB is a DC component in the dq coordinate system. In this embodiment, the fundamental wave current control unit 30h corresponds to the "fundamental wave setting unit".

The harmonic voltage setting unit 30p calculates the d-axis harmonic voltage Vdh, which is a d-axis voltage component, and the q-axis harmonic voltage Vqh, which is a q-axis voltage component, based on the command angular velocity ωtgt. In the present embodiment, the harmonic voltage setting unit 30p includes a memory as a storage unit. The d-axis harmonic voltage Vdh and the q-axis harmonic voltage Vqh are stored in the memory in relation to the command angular velocity ωtgt.

In the present embodiment, since the 10th-order electromagnetic force is shifted to the 14th-order electromagnetic force, the electromagnetic force to be reduced becomes the 11th and 13th-order electromagnetic force in the fixed coordinate system. The 11th and 13th order electromagnetic forces are converted into the 12th order electromagnetic force in the dq coordinate system. Therefore, the d-axis harmonic voltage Vdh and the q-axis harmonic voltage Vqh are 12th-order harmonic voltages in the dq coordinate system.

The d-axis superimposition unit 30q adds the d-axis harmonic voltage Vdh to the d-axis fundamental wave voltage VdB and outputs it. The q-axis superimposition unit 30r adds the q-axis harmonic voltage Vqh to the q-axis fundamental wave voltage VqB and outputs it.

The three-phase conversion unit 30s has U, V, and W phase command voltages based on the output value “VdB + Vdh” of the d-axis superimposition unit 30q, the output value “VqB + Vqh” of the q-axis superimposition unit 30r, and the electric angle θe. Calculate VU, VV, VW. The calculated command voltages VU, VV, and VW are input to the modulator 30h.

The control device 30 includes a correction value calculation unit 30t. The correction value calculation unit 30t according to the present embodiment performs the processing described in the following after performing the processing in steps S10 to S16 of FIG. The correction value calculation unit 30t extracts the amplitude and phase difference of the 11th and 13th order magnetic fluxes to be reduced based on the spatial distribution of the magnetic flux calculated in step S16. The correction value calculation unit 30t determines only the d-axis harmonic voltage Vdh of the d-axis harmonic voltage Vdh and the q-axis harmonic voltage Vqh, or the d-axis, based on the amplitude and phase difference of the extracted 11th and 13th order magnetic fluxes. A correction value for correcting both the harmonic voltage Vdh and the q-axis harmonic voltage Vqh is calculated. Based on the calculated correction value, the correction value calculation unit 30t has only the d-axis harmonic voltage Vdh of the d-axis harmonic voltage Vdh and the q-axis harmonic voltage Vqh, or the d-axis harmonic voltage Vdh and the q-axis harmonic. Both the voltage Vqh are corrected.

The same effect as that of the second embodiment can be obtained by the present embodiment described above.

<Fourth Embodiment>
Hereinafter, the fourth embodiment will be described with reference to the drawings, focusing on the differences from the third embodiment. In the third embodiment, the d and q-axis harmonic voltages Vdh and Vqh are stored in the memory in advance. In the present embodiment, the d and q-axis harmonic voltages Vdh and Vqh are not stored in the memory in advance. Therefore, the control device 30 includes a harmonic generation unit 30u as shown in FIG. 11 in place of the harmonic voltage setting unit 30p and the correction value calculation unit 30t. In FIG. 11, the same components as those shown in FIG. 10 above are designated by the same reference numerals for convenience.

The harmonic generation unit 30u generates d and q-axis harmonic voltages Vdh and Vqh based on at least one magnetic flux detection value of each Hall element 42a to 42c. Specifically, the harmonic generation unit 30u performs the processes described in the following after performing the processes of steps S10 to S16 of FIG. 7.

The harmonic generation unit 30u extracts the amplitude and phase difference of the 11th and 13th order magnetic fluxes to be reduced based on the spatial distribution of the magnetic flux calculated in step S16. The harmonic generation unit 30u generates a d-axis harmonic voltage Vdh and a q-axis harmonic voltage Vqh based on the amplitude and phase difference of the extracted 11th and 13th order magnetic fluxes. Specifically, the harmonic generator 30u has extracted the amplitude of the eleventh-order magnetic flux so as to realize a spatial distribution of the magnetic flux that can shift the frequency of the current electromagnetic force from the resonance frequency of the annular mode of the rotor 14. And the amplitude and phase of the first harmonic currents IUH1, IVH1, IWH1 are calculated based on the phase difference, and the second harmonic currents IUH2, IVH2, IWH2 are calculated based on the amplitude and phase difference of the extracted 13th order magnetic flux. Calculate the amplitude and phase.

The harmonic generation unit 30u calculates the d-axis harmonic voltage Vdh and the q-axis harmonic voltage Vqh based on the calculated amplitudes and phases of the respective harmonic currents IUH1, IVH1, IWH1, IUH2, IVH2, and IWH2. After that, the energization of the windings 12U, 12V, and 12W is resumed. After that, the harmonic generation unit 30u generates and outputs d, q-axis harmonic voltages Vdh, Vqh.

According to the present embodiment described above, since it is possible to eliminate the work of matching each harmonic voltage stored in the memory, it is possible to reduce the labor required when designing the control device 30.

(Other embodiments)
Each of the above embodiments may be modified as follows.

-In each of the above embodiments, the magnetic flux of the magnet may be acquired during the period in which the windings 12U, 12V, and 12W are energized to drive the motor 10. More specifically, by subtracting the magnetic flux generated by the current flowing through the windings 12U, 12V, 12W from the total magnetic flux of the magnetic flux generated by the current flowing through the windings 12U, 12V, 12W and the magnet magnetic flux, The magnetic flux of the magnet may be acquired. In this case, when the system is provided with a second magnetic flux detecting unit for detecting the interlinkage magnetic flux of the stator 12, in addition to the first magnetic flux detecting unit which is the Hall elements 42a to 42c for detecting the magnet magnetic flux, the first and first are used. 2. The magnet magnetic flux acquisition process may be performed based on the magnetic flux detection value of the magnetic flux detection unit.

Further, during the period in which the windings 12U, 12V, and 12W are energized and the motor 10 is driven, the magnet magnetic flux is acquired by using the first magnetic flux detecting unit without using the second magnetic flux detecting unit. May be good. This is based on the fact that the influence of the magnetic flux generated by the current flowing through the windings 12U, 12V, and 12W becomes small depending on the installation position of the magnetic flux detection unit, and this influence can be ignored. Hereinafter, in this configuration, FIG. 12 shows an example of the processing executed by the correction value calculation unit 30i of FIG. In FIG. 12, the same step numbers are assigned to the same processes as those shown in FIG. 5 above for convenience.

In this series of processes, if an affirmative determination is made in step S10, the process proceeds to step S22, and each harmonic voltage VUH1, VVH1, VWH1, VUH2, VVH2, VWH2 is output from the harmonic voltage setting unit 30e, and each harmonic is output. Determine if the voltage is superimposed.

If a negative determination is made in step S22, the process proceeds to step S14. On the other hand, if an affirmative determination is made in step S22, the process proceeds to step S24, and the harmonic voltage setting unit 30e is instructed to stop the generation and output of each harmonic voltage VUH1, VVH1, VWH1, VUH2, VVH2, VWH2. To do. At this time, the outputs of the fundamental wave voltages VUB, VVB, and VWB from the fundamental wave voltage setting unit 30d do not have to be stopped. When the process of step S24 is completed, the process proceeds to step S14.

-In the second embodiment, the magnetic flux strain Δφ may be calculated based on the detected magnet magnetic flux φm, and the first and second harmonic currents may be calculated based on the relationship of the above equation (eq15). As a result, the man-hours for matching the first and second harmonic voltages can be reduced. The calculated first and second harmonic currents may be converted into first and second harmonic voltages based on the voltage equation of the motor 10.

-In the first embodiment, the harmonic command value stored in the memory is not limited to the first and second harmonic voltages, but may be the first and second harmonic currents. In this case, for example, the set first and second harmonic currents are converted into the first and second harmonic voltages based on the voltage equation of the motor 10, and the converted first and second harmonic voltages are superimposed on each unit. The output may be 30 fU, 30 fV, 30 fW, 30 gU, 30 gV, 30 gW. The same applies to the third embodiment.

-The application target of the present invention is not limited to a system provided with a rotation angle detection unit that detects the rotation angle of the rotor 14. For example, the present invention may be applied to a system in which position sensorless control is performed without a rotation angle detection unit.

-The magnetic sensor is not limited to a Hall element, and may be, for example, a linear Hall IC or a search coil installed so as to interlink with a magnetic flux. Further, the magnetic sensor is not limited to the one arranged at a position where the leakage flux of the magnetic pole can be detected, but is also arranged at a position where the main magnetic flux between the magnetic pole and the tooth can be detected in the case accommodating the motor 10. May be good.

-In the above embodiment, in order to convert the 10th-order electromagnetic force into the 14th-order electromagnetic force, two harmonic currents of the 11th and 13th orders are superimposed on the fundamental wave current, but the present invention is not limited to this. For example, one or more harmonic currents may be superimposed on the fundamental current. Specifically, for example, when the electromagnetic force near the resonance angular velocity is the fourth-order electromagnetic force, the fourth-order electromagnetic force is converted from the fourth-order to the twelfth-order electromagnetic force in order to convert the fourth-order electromagnetic force into a twelfth-order electromagnetic force far away from the resonance angular velocity. It suffices to superimpose the four harmonic currents of the fifth order, the seventh order, the ninth order, and the eleventh order, which are all the odd-order harmonic currents included in

-In the second embodiment, the harmonic generation unit 30j may determine the harmonic voltage for reducing the radial electromagnetic force fluctuation acting on the rotor 14 while superimposing the harmonic voltage. Specifically, the harmonic generation unit 30j is basically superimposed on the U, V, W phase fundamental wave voltages VUB, VVB, VWB based on the first amplitude V11 and the first phase difference γ calculated in step S20 of FIG. The first harmonic voltage VUha, VVha, and VWha to be obtained are calculated. Further, the harmonic generation unit 30j is a second base that is superimposed on the U, V, W phase fundamental wave voltages VUB, VVB, and VWB based on the second amplitude V13 and the second phase difference δ calculated in step S20. The harmonic voltages VUhb, VVhb, and VWhb are calculated. The harmonic generator 30j changes the amplitude and phase of the basic first harmonic voltages VUha, VVha, VWha, and also changes the amplitude and phase of the basic second harmonic voltages VUhb, VVhb, VWhb. The amplitude and phase of each harmonic voltage VUha, VVha, VWha, VUhb, VVhb, VWhb that minimizes the radial electromagnetic force fluctuation acting on the rotor 14 are determined.

-The rotation angle detection unit that detects the rotation angle of the motor is not limited to the Hall element, and may be, for example, an encoder or a resolver. In this case, the control system may be provided with a magnetic flux detection unit separately.

Further, the control system may not be provided with a rotation angle detection unit. In this case, position sensorless control is executed without using a rotation angle detection unit such as a Hall element. In this case, the control system may be provided with a magnetic flux detection unit separately.

-The control amount of the motor is not limited to the rotational angular velocity, but may be, for example, torque. In this case, for example, in FIG. 10, the command torque may be input to the command value setting unit 30 m instead of the command angular velocity ωtgt. Further, the control amount of the motor may be, for example, a rotation angle position.

-The motor is not limited to a concentrated winding motor, and a distributed winding motor may be used. Further, the motor is not limited to the outer rotor type, and an inner rotor type motor may be used. Even if the winding method and rotor type are different, the application of the present invention is effective if noise is generated by the resonance phenomenon of the rotor.

Further, the motor is not limited to a three-phase motor, and may be a multi-phase motor having four or more phases. In addition, the motor is not limited to a permanent magnet field type synchronous machine having a permanent magnet in the rotor, and may be, for example, a winding field type synchronous machine having a field winding in the rotor. In addition, the motor is not limited to that for a blower.

10 ... motor, 20 ... inverter, 30 ... control device.

Claims (9)

A rotary electric machine (10) having a stator (12) around which a winding (12U to 12W) is wound and a rotor (14) on which a magnetic pole (14a) is formed is electrically connected to the rotary electric machine. It is applied to a system including a power converter (20) that drives a rotary electric machine by passing a drive current through the winding.
A magnetic flux information acquisition unit that acquires magnetic flux information of the magnetic poles,
Fundamental wave setting units (30 d, 30 h) for setting the fundamental wave command value for passing the fundamental wave current through the winding, and
A storage unit that stores in advance a harmonic command value for passing a reduced current , which is a harmonic current for reducing radial electromagnetic force fluctuations acting on the rotor, through the winding.
A harmonic setting unit (30 e, 30p ) that corrects the harmonic command value stored in the storage unit based on the magnetic flux information acquired by the magnetic flux information acquisition unit.
Wherein the harmonic command value corrected by the harmonic setting unit is superimposed, based on the set the fundamental wave command value by the fundamental wave setting unit, a current obtained by superimposing the reduced current to the fundamental wave current A control device (30) for a rotating electric machine including an operation unit (30) for operating the power converter so that a certain drive current flows in the winding. The rotation angle calculation unit (30 a ) for calculating the rotation angle of the rotary electric machine, and
Extraction to extract the magnetic flux component of the magnetic pole that fluctuates at the angular velocity of the electromagnetic force fluctuation to be reduced based on the rotation angle calculated by the rotation angle calculation unit and the magnetic flux information acquired by the magnetic flux information acquisition unit. With a department ,
The control device for a rotating electric machine according to claim 1 , wherein the harmonic setting unit corrects the harmonic command value stored in the storage unit based on the magnetic flux component extracted by the extraction unit. A rotation angle calculation unit (30 a ) for calculating the rotation angle of the rotary electric machine based on the magnetic flux information acquired by the magnetic flux information acquisition unit is provided.
Claim 1 or 2 that the fundamental wave setting unit sets the fundamental wave command value for controlling the control amount of the rotary electric machine to the command value based on the rotation angle calculated by the rotation angle calculation unit. The control device for the rotary electric machine described in 1. The energization stop portion for temporarily stopping the energization of the winding while the rotor is rotating is provided.
The control of a rotary electric machine according to any one of claims 1 to 3 , wherein the magnetic flux information acquisition unit acquires magnetic flux information of the magnetic poles during a period in which energization of the winding is stopped by the energization stop unit. apparatus. The system is provided with magnetic flux detection units (42a to 42c) for detecting the magnetic flux of the magnetic poles.
The control device for a rotary electric machine according to any one of claims 1 to 4 , wherein the magnetic flux information acquisition unit acquires the magnetic flux detected by the magnetic flux detection unit as the magnetic flux information. The control device for a rotary electric machine according to claim 5 , wherein the magnetic flux detection unit is arranged at a position where the leakage flux of the magnetic pole can be detected in the system. The control device for a rotary electric machine according to claim 5 or 6 , wherein the magnetic flux detection unit is a Hall element, a linear Hall IC, or a search coil. The control device for a rotary electric machine according to any one of claims 1 to 7 , wherein the rotary electric machine is a permanent magnet synchronous machine in which the magnetic pole is formed of a permanent magnet. The control device for a rotary electric machine according to any one of claims 1 to 8 , wherein the rotary electric machine is an outer rotor type.

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