JP6565087B2 - Frame error rate prediction apparatus, radio communication apparatus and radio communication system using the same - Google Patents

Description

  The present invention relates to a frame error rate prediction apparatus, a radio communication apparatus using the same, and a radio communication system.

  In recent years, offloading of mobile traffic has progressed in the ISM (industrial, scientific, and medical radio) band, and improvement in frequency utilization efficiency of a wireless local area network (LAN) is desired. To achieve high frequency utilization efficiency, it is necessary to achieve a frame error rate (FER) that is sufficiently low in practice and to use a transmission rate that is as high as possible. In general, it is conceivable to control MCS (modulation and coding scheme) according to propagation conditions so that FER is about 1% to 10%. Here, MCS refers to a table of combinations determined in advance for modulation schemes and channel coding rates (see, for example, Patent Document 1). For example, when the reception status of the receiver is poor, or transmission data that requires communication with a low error rate, MCS with a low transmission rate is used. Conversely, when the reception status of the receiver is good or a relatively high error The transmission data that allows the rate is subjected to adaptive control using a determination method such as using MCS with a high transmission rate.

As a method of controlling to an appropriate MCS in a wireless LAN, a method of adjusting an MCS according to a transmission success rate or the number of retransmissions and selecting an MCS suitable for a propagation state is known (see Non-Patent Document 1). . However, in this method, since frame transmission is performed with an MCS that does not match the propagation state until an optimal MCS is selected, there is a risk that throughput may be reduced due to retransmission or low-rate transmission. For this reason, in order to improve the transmission efficiency, it is desirable to predict the FER (Frame Error Rate) in advance and determine the MCS based on the result. However, in order to realize this, high-precision FER prediction is required.
It is known that the FER can be accurately predicted by the theoretical calculation of the FER value based on the pairwise error probability (PEP) in the region where the FER is 1% to 10% (Non-patent Document 2). .

  When the error rate is obtained by PEP in the currently popular IEEE802.11a or later wireless LAN, it is necessary to obtain the PEP in convolutional coded OFDM (COFDM). The PEP in COFDM has been studied in Non-Patent Document 3 and the like so far.

  In Non-Patent Document 3, PEP analysis is performed when a bit error after interleaving can be regarded as random. When it can be regarded as a random error, an error path whose Hamming distance is separated by a minimum free distance becomes dominant, and the frequency response of the propagation path can also be regarded as random. Non-Patent Document 3 presents a PEP derivation method using this property.

  However, when a realistic wireless LAN operation is assumed, the communication path is not always an environment that can be regarded as a random error. In a wireless LAN such as IEEE802.11a, the bandwidth and interleave size are not sufficiently wide and only a delay dispersion of about several hundreds (ns) is taken into consideration, so that the occurrence of bit errors can be sufficiently dispersed. There are cases where it is not possible. For this reason, the frequency response cannot be regarded as random equivalently, and the above-described approximate expression of Non-Patent Document 3 that does not depend on the frequency response of the propagation path is not established. Further, the premise that the error path PEP of the minimum free distance is dominant is not satisfied, and it is necessary to consider the error path PEP having a Hamming distance greater than the minimum free distance. Therefore, the PEP derivation method presented in Non-Patent Document 3 cannot accurately approximate the PEP in the wireless LAN environment.

  In order to solve such a problem, in the technique disclosed in Non-Patent Document 4, in a communication system that communicates using a convolutional coding OFDM method, a transmitter, a receiver, and a receiver receive bit error rate (BER) prediction. PEP is derived based on information representing noise attenuation by the communication path and noise power estimated in the machine or both. In the case of OFDM communication under a multipath environment, the influence of the propagation path varies depending on the frequency response of the subcarrier. Furthermore, multilevel modulation is also used in wireless LAN. Therefore, each codeword bit is affected differently and the PEP is different at each position of the trellis. In the technique disclosed in Non-Patent Document 4, the PEP is calculated at each time point in the trellis section corresponding to the interleaver size in order to consider the difference in PEP depending on the position of the trellis, and a bit error after decoding from the obtained PEP. BER prediction is performed by calculating an expected value of the number of bits. As the FER prediction, it is assumed that the error rate of each information bit is equal, and the FER is predicted based on the predicted BER.

JP 2010-41074 A

JP 2011-111433 A

JP 2013-187561 A

S. Biaz and S. Wu, “Rate adaptation algorithms for IEEE 802.11 networks: A survey and comparison,” 2008 IEEE Symposium on Computers and Communications, pp.130-136, July 2008.

A. Martinez, A. Guillen i Fabregas, and G. Caire, “Error probability analysis of bit-interleaved coded modulation,” IEEE Transactions on Information Theory, vol.52, no.1, pp.262-271, Jan. 2006.

Y. Hori and H. Ochiai, “Performance analysis and interleaver structure optimization for short-frame BICM-OFDM systems,” IEEE Transactions on Wireless Communications, vol. 15, no. 1, pp. 651-662, Jan. 2016.

Tsuji et al., “Prediction of bit error rate for convolution-coded OFDM transmission suitable for IEEE 802.11 wireless LAN,” IEICE Tech. IT 2016-125, Mar. 2017.

  In the prior art as described above, the error rate of each information bit is different by using different SNR and multilevel modulation for each subcarrier (that is, PEP is different at each position of the trellis), but they are averaged. The calculated BER is calculated. However, since a frame error occurs when any one of the information bits in the frame becomes a bit error, it is necessary to consider the difference in error rate of each information bit in order to predict FER with high accuracy. Therefore, as assumed in the conventional method, when calculating FER from an averaged value such as BER, the difference in error rate of each bit in the frame cannot be considered, so FER is calculated based on BER. It may not always be appropriate to do so.

  The present invention has been made to solve the above-described problems, and its purpose is to be able to predict the FER in a communication system in which the PEP differs depending on the position on the trellis in the convolutional code. And a wireless communication apparatus and a wireless communication system using the same.

  According to one aspect of the present invention, there is provided a FER prediction apparatus for a communication system in which communication is performed using a convolutional coding scheme decodable by a trellis diagram, and the same interleave processing is repeatedly performed on transmission bits in a frame. A storage means for storing information on a set of convolutional coding rate and modulation scheme that is set in advance as employed in the coding scheme, and a propagation path power attenuation estimated in the communication path Based on the propagation path characteristic information and noise power, for each set of coding rate and modulation scheme stored in the storage means, the error path PEP in the decoding process for convolutional coding is calculated on the trellis corresponding to the frame size. FER predicting means for predicting FER as the sum of PEPs calculated at each position. That.

Preferably, in the frame error rate predicting means, the transmission bit is divided into a plurality of blocks for each block length set so that the propagation path fluctuation with respect to the encoded sequence after the deinterleaving process can be regarded as periodic in the frame. If it is, the block error rate calculated in a section of the trellis diagram corresponding to one of the blocks as the sum of the pair-wise error rate, as the block error rate of each block are equal to each other, the block error rate and frame The frame error rate is predicted from the number of blocks included.

  Preferably, the communication system performs communication using the orthogonal frequency division multiplexing method, and the block is one symbol in the orthogonal frequency division multiplexing method.

  Preferably, the propagation path characteristic information is a power delay profile estimated in the communication path.

  Preferably, the propagation path characteristic information is an impulse response of the propagation path.

According to another aspect of the present invention, there is provided a wireless communication apparatus that communicates by a convolutional coding scheme that can be decoded by a trellis diagram and communicates by a convolutional coding scheme in which the same interleaving process is repeatedly executed on transmission bits in a frame. Te, transmitted from the receiving apparatus, receiving means for receiving the estimated value of the propagation path characteristics information representative of the attenuation of the communication route power and noise power as the feedback information, previously set as being employed by convolution coding Storage means for storing the information of the coding rate and modulation scheme set of the convolutional coding, and the throughput of the coding rate and modulation scheme set stored in the storage means based on the feedback information. Selection means for selecting a set to be maximized, and the selection means is stored in the storage means based on the propagation path characteristic information and noise power. For the each set of coding rate and modulation scheme, the pair-wise error rate of error paths in the decoding process for the convolutional coding, calculated at each position of the section of the trellis diagram corresponding to the size of the frame, calculated in each position A frame error rate predicting means for predicting a frame error rate as a sum of the pairwise error rates , and a modulation for selecting a set having the maximum throughput within a predetermined frame error rate based on the predicted frame error rate A transmission means for performing transmission by performing convolutional coding and modulation processing on transmission data according to the group selected by the modulation method selection means .

Preferably, in the frame error rate predicting means, the transmission bit is divided into a plurality of blocks for each block length set so that the propagation path fluctuation with respect to the encoded sequence after the deinterleaving process can be regarded as periodic in the frame. If it is, the block error rate calculated in a section of the trellis diagram corresponding to one of the blocks as the sum of the pair-wise error rate, as the block error rate of each block are equal to each other, the block error rate and frame The frame error rate is predicted from the number of blocks included.

Preferably, the wireless communication apparatus performs communication using the orthogonal frequency division multiplexing method, and the block is one symbol in the orthogonal frequency division multiplexing method.

  Preferably, the propagation path characteristic information is a power delay profile estimated in the communication path.

  Preferably, the propagation path characteristic information is an impulse response of the propagation path.

According to still another aspect of the present invention , there is provided a wireless communication system that communicates by a convolutional coding OFDM system that can be decoded by a trellis diagram and that repeatedly performs the same interleaving process on transmission bits in a frame. A communication path state estimating means for estimating propagation path characteristic information indicating noise attenuation of the propagation path and noise power, and estimated propagation path characteristic information and noise power in the communication path. And a transmission device, the transmission device receiving the propagation path characteristic information and the estimated value of noise power transmitted from the reception device as feedback information, and a convolutional code Stores information on a set of convolutional coding rate and modulation scheme that are set in advance for use in the OFDM system. Storage means for selecting, based on the feedback information, a selection means for selecting a set that maximizes the throughput among the combinations of coding rate and modulation scheme stored in the storage means, and the selection means includes a propagation path A trellis diagram corresponding to the frame size of the pairwise error rate of the error path in the decoding process for convolutional coding for each set of coding rate and modulation scheme stored in the storage means based on the characteristic information and noise power Frame error rate predicting means for predicting the frame error rate as the sum of the pairwise error rates calculated at each position, and a predetermined frame error rate based on the predicted frame error rate. the inner, and a modulation scheme selection means for selecting a set of maximum throughput, selecting the modulation scheme selection means Depending on the set, the transmission data, further comprising a second transmission means for transmitting running convolutional coding and modulation process.

“Pairwise error rate (PEP)” means that when a codeword C _L having a length _L is transmitted from a transmitter, when this sequence is decoded on the receiver side including C _L ′ and an error, the code word C _L refers to the probability determining mistakenly codeword C _L '.

  The “minimum free distance” is a convolutional code, where c (u) and c (v) are codewords for two information sequences u and v. This is the minimum Hamming distance.

  The “power delay profile” represents how the received power is distributed in the delay time region when it is assumed that the propagation path is composed of a number of paths having different delay times. That is, in the actual propagation path, since the path length differs for each path, the impulse response has a time spread. Therefore, the “power delay profile” is represented by the square set average value of the impulse response.

  “Diversity order” refers to the number of subcarriers in which the frequency response of each subcarrier to which different bits are mapped between a codeword corresponding to a certain error path and a transmission codeword can be regarded as random (uncorrelated). In an environment where communication channel errors can be regarded as random errors, the diversity order is maximum (the frequency response of the propagation channel through which different bits pass between the codeword corresponding to the error path and the transmission codeword is uncorrelated). If the diversity order is lower than that, a burst error is likely to occur, and the BER deteriorates.

  The “error path” refers to a path in which a bit error occurs when the path is selected by maximum likelihood decoding for the convolutional code.

  An “error pattern” is a code word of an error path corresponding to a different section between the true path and the error path. In a predetermined convolutional code, there are a finite number of error patterns that can be generated according to the generator polynomial. Therefore, a number is assigned to the error pattern, and for example, it is referred to as a “p-th error pattern”.

  “Hamming distance” refers to the number of different bits at corresponding positions in two bit strings.

  According to the present invention, it is possible to predict the FER in a communication system in which the PEP differs depending on the position on the trellis in the convolutional code.

1 is a block diagram for illustrating a configuration of a wireless communication system according to a first embodiment. It is a conceptual diagram for demonstrating typically the transmission and reception processing by the OFDM system shown in FIG. FIG. 10 is a diagram illustrating an error path at a dH = 5 of a convolutional code having a constraint length of 3 and a generator polynomial (5, 7) ₈ ; It is a block diagram for demonstrating the structure of the adaptive rate control part 1110 shown in FIG. 6 is a flowchart for illustrating adaptive rate control according to the first embodiment. It is a conceptual diagram for demonstrating a problem in case PEP changes with positions on a trellis. It is a conceptual diagram which shows the concept of Formula (4). It is a conceptual diagram for demonstrating the pattern of PEP in case the same interleaving process is repeatedly implemented by each block within the flame | frame. It is a conceptual diagram for demonstrating the procedure which estimates FER by the calculation per block.

  Hereinafter, configurations of a wireless communication system and a wireless communication apparatus according to an embodiment of the present invention will be described. In the following embodiments, components and processing steps given the same reference numerals are the same or equivalent, and the description thereof will not be repeated unless necessary.

  As will be described below, in the present embodiment, an error in a communication channel is not necessarily a random error, such as in a wireless LAN environment, and the PEP differs depending on the position in the convolutional code trellis. In such a case, a method for accurately predicting FER will be described.

  In the following, in order to make the technical significance of the present embodiment easier to understand, the FER calculation method will be described in the following order.

  (1) First, FER calculation assuming a system in which the PEP is the same at any position on the trellis will be described.

  (2) Next, as a method for calculating the FER according to the present embodiment, a method for calculating the PEP in the trellis section corresponding to the frame size and predicting the FER will be described.

(3) Further, on the assumption of (2), in a communication system such as a wireless LAN in which the same interleaving is repeatedly performed, when it can be considered that there is no propagation path variation in the frame, the propagation path received by the encoded sequence after deinterleaving By setting the block so that the fluctuation is periodic with the block length as a period, and using the block error rate equal in the block period, the section for calculating the PEP is limited to the trellis section corresponding to one block. However, a method for predicting the FER will be described (reducing the calculation cost).
[Embodiment 1]
FIG. 1 is a block diagram for explaining a configuration of a wireless communication system according to the first embodiment.

  Referring to FIG. 1, transmitting apparatus 1000 performs error correction coding section 1002 for performing error correction coding processing by convolutional code on transmission sequence data, and data after error correction coding. Interleaving section 1004 that performs interleaving processing, and serial-parallel conversion is performed on the interleaved data string, and the data string is divided into the number of subcarriers based on the MCS selected as described later, and the divided data Next, a modulation unit 1010 for performing subcarrier modulation and a digital signal output from the modulation unit 1010 are subjected to inverse Fourier transform processing and guard interval addition processing to generate OFDM symbols, and digital analog conversion processing is performed. Orthogonal Frequency Division Multiplexing (OFDM) modulator 1012 , A high-frequency processing unit (RF unit) 1014 that performs orthogonal modulation processing, up-conversion processing, power amplification processing, and the like on the signal after OFDM modulation, and an antenna 1020 for transmitting a high-frequency signal from the RF unit 1014 Including.

  Note that the RF unit 1014 also performs low-noise amplification processing, down-conversion processing, orthogonal demodulation processing, and the like on the signal received by the antenna 1020.

  Further, the modulation scheme of subcarrier modulation is not particularly limited, but for example, there are types such as BPSK, QPSK, 16QAM, and 64QAM.

  Furthermore, the transmission apparatus 1000 receives the power delay profile and noise power estimation value from the reception apparatus 2000 side via the RF unit 1014, and performs a demodulation and decoding process, a reception processing unit 1100, and the received power An adaptive rate control unit that performs control to adaptively change the MCS based on the delay profile and the estimated noise power, and controls the coding rate of the error correction coding unit 1002 and the modulation scheme in the modulation unit 1010 1110.

  Note that the communication method of the power delay profile and the estimated value of noise power from the receiving device 2000 may be the same communication method as the data transmission, or another communication method.

  The receiving apparatus 2000 includes an antenna 2002, an RF unit 2010 that performs a low-noise amplification process, a down-conversion process, an orthogonal demodulation process, and the like on the signal of the antenna 2002, and an analog-digital conversion process on the signal from the RF unit 2010, An OFDM demodulator 2012 for executing OFDM demodulation processing such as guard interval removal processing and Fourier transform processing, and a signal from the OFDM demodulator 2012 generates a received data string by inverse processing of the modulator 1010 A demodulating unit 2014, a deinterleaving unit 2016 for executing deinterleaving processing on the output of the demodulating unit 2014, and an error correcting unit 2018 for executing error correction processing by decoding the convolutional code.

  Also in receiving apparatus 2000, RF section 2010 performs orthogonal modulation processing, up-conversion processing, power amplification processing, and the like for transmission of a power delay profile and an estimated value of noise power.

  The receiving apparatus 2000 further uses a power delay profile / noise power estimation for calculating an estimated value of a power delay profile and an estimated value of noise power based on, for example, a pilot signal in a signal received via the RF unit 2010. Unit 2100 and a transmission processing unit 2110 for converting the estimated value from power delay profile / noise power estimation unit 2100 into a transmission signal and transmitting the signal from antenna 2002 to transmitting apparatus 1000 via RF unit 2010. Including.

  FIG. 2 is a conceptual diagram for schematically explaining transmission and reception processing in the OFDM scheme shown in FIG.

  As shown in FIG. 2, the convolutionally encoded and interleaved transmission signal is mapped to each subcarrier as a complex signal in a predetermined modulation method, and after inverse Fourier transform, it is converted from digital to analog to perform orthogonal modulation or the like. The frequency conversion process is performed and sent to the transmission line.

  The signal received from the transmission path undergoes frequency inverse transform processing including quadrature detection, etc., then analog-digital converted, Fourier transformed, and the inversely mapped received signal undergoes deinterleave processing and error correction processing using convolutional codes. Is done.

Consider a trellis path (hereinafter referred to as a true path) that generates codewords whose constituent bits are all 0 and a path (hereinafter referred to as an error path) that generates codewords separated by a Hamming distance d _H (the convolutional code is a linear code. The generality is not lost even if the path and error path are set as described above).

FIG. 3 is a diagram illustrating an error path in a convolutional code having a constraint length of 3 and a generator polynomial (5, 7) ₈ at d _H = 5.

  In this case, the true path corresponds to the codewords ..., 0, 0, 0, 0, 0, 0, ..., and the error path corresponds to the codewords ..., 1, 1, 0, 1, 1, 1, ... To do. Since the error path leaves the true path at a certain position on the trellis (t = 2 in FIG. 3) and then merges, only this section has a different bit pattern. In the present embodiment, this different bit pattern is called an error pattern. For example, in FIG. 3, the error pattern is [1, 1, 0, 1, 1, 1].

Further, as will be described later, the FER is predicted based on such an error pattern that can occur for a predetermined convolutional code.
[Configuration of FER prediction processing]
FIG. 4 is a block diagram for explaining the configuration of adaptive rate control section 1110 shown in FIG.

  Referring to FIG. 4, adaptive rate control section 1110 includes an MCS storage section 3020 for storing information on a set of preset MCS, an estimated value of the power delay profile transmitted from receiving apparatus 2000 side, and Based on the information of the frame size used for communication for each MCS stored in the MCS storage unit 3020 in response to the estimated noise power, the FER prediction unit 3030 predicts the FER for each MCS. And an MCS selection unit 3040 that selects an MCS that achieves the maximum throughput among MCSs that are lower than the FER set and requested in advance in the system based on the FER value.

  The operation of the FER prediction unit 3030 will be described in more detail later.

  FIG. 5 is a flowchart for explaining adaptive rate control according to the first embodiment.

  Referring to FIG. 5, first, on the receiving apparatus 2000 side, power delay profile / noise power estimation unit 2100 performs estimation of the power delay profile and noise power (S100).

  The estimation of the power delay profile and the estimation of the noise power are not particularly limited. For example, the method disclosed in the following document can be used.

Known Document 1: T. Cui and C. Tellambura, “Power delay profile and noise variance estimation for OFDM,” IEEE Communications Letters, vol. 10, no. 1, pp. 25-27, Jan 2006
The estimated value is transmitted from the receiving device 2000 (S102), received by the transmitting device 1000 (S104), and the FER predicting unit 3030 of the transmitting device 1000 determines the error path at each position in the trellis section corresponding to the frame size. By calculating the PEP and calculating the sum within the frame, a predicted value of FER is calculated for each MCS (S108), and the MCS selection unit 3040 is based on the calculated FER and is calculated from the predetermined MCS. The MCS having the maximum throughput is selected within a range that achieves the prescribed FER (S110).

  In accordance with the selected MCS, the error correction coding unit 1002 performs convolutional coding at the selected coding rate, and the modulation unit 1010 performs subcarrier modulation with the selected modulation scheme (S112). .

  The modulated data is transmitted from the transmission apparatus 1000 (S114) and received by the reception apparatus 2000 (S116).

With such processing, it is possible to adaptively change the coding rate and the modulation method when the power delay profile and noise power are estimated on the receiving device side and fed back to the transmitting device side. Therefore, by accurately predicting the FER, it is possible to reduce the number of frame transmissions in the MCS that do not match the propagation state and improve the throughput.
[Operation performed by FER prediction unit 3030]
Below, the operation | movement which the FER prediction part 3030 performs is demonstrated according to numerical formula.

1. Features of the method of the present embodiment As will be apparent from the following description, the FER prediction of the present embodiment has the following features.

  i) Diversity order that can be achieved from the impulse response length of the propagation path and the minimum free distance is calculated, and prediction is possible even in a propagation environment where only a low diversity order can be achieved.

  ii) The error rate is calculated considering an error path longer than the minimum free distance.

  iii) In the trellis section corresponding to the frame size, the FER is predicted by calculating the PEP for each position and taking the sum.

  This will be described in more detail below.

2. 2. System Model 2.1 System Model For the sake of simplicity, SISO (single-input single-output) -OFDM is assumed below. The number of data subcarriers is K _data and the FFT (fast Fourier trans-form) size is K _fft . The propagation path is assumed to be a frequency selective fading propagation path and does not vary within the OFDM symbol. When the propagation path is regarded as a finite impulse response filter, each tap coefficient is expressed by the following equation.

Here, h = [h ₁ ,..., H _L ] ^T (L is an impulse response length) is a vector in which each element is a random variable that is independent of each other and follows a complex normal distribution with mean 0 and variance 1. It is a diagonal matrix showing a power delay profile. However, (...) ^T represents transposition of a vector or a matrix, and the element of P satisfies the following expression.

L is assumed not to exceed the guard interval length. After removal of the guard interval, the received signal Y _k of the k-th subcarrier subjected to FFT is given as follows.

Here, X _k is a transmission symbol, Z _k is an FFT result of complex additive white noise having an average of 0 and a variance σ ² . H _k is the frequency response of the propagation path in the k-th subcarrier and is given by the following equation.

Here, g ^H _Kfft (k) is a vector extracted from the first element to the L-th element in the k-th row of the FFT matrix, and (...) ^H represents Hermitian transpose. In addition, the following relationship holds.

(1) Processing by the prior art to be compared with the operation of the FER prediction unit of the present embodiment Based on the technology disclosed in the following publicly known document, first, a system in which the PEP is the same at any position on the trellis The calculation method of FER when assuming the above will be described.

  That is, an interleaver in which the influence of the propagation path on which the codeword error bit is received is large enough to be considered uncorrelated and the codeword error bit is mapped with equal probability to each bit position in the modulation symbol at any position on the trellis. A method for calculating the FER when assuming a system using the above will be described.

Known Document 2: M. Pursley and D. Taipale, "Error Probabilities for Spread-Spectrum Packet Radio with Convolutional Codes and Viterbi Decoding," in IEEE Transactions on Communications, vol. 35, no. 1, pp. 1-12, Jan 1987.
Known Document 3: CY Lou and B. Daneshrad, "PER prediction for convolutionally coded MIMO OFDM systems-An analytical approach," MILCOM 2012-2012 IEEE Military Communications Conference, Orlando, FL, 2012, pp. 1-6.
First, since FER is the probability that any of the error events will occur at any position in the frame, the error event that the p-th error pattern occurs at the time t of the trellis in the convolutional code is represented by et _, Assuming _p (underscore), FER is expressed as the following equation. Here, “x (underbar)” indicates that an underbar is added under the character “x”, and indicates that this character is a vector.

  Here, it is further considered that the following union expression that holds for the upper limit of the probability of the sum event is applied to the FER expression (3).

  In that case, the FER is expressed as follows.

  Furthermore, according to the known document 2, when the PEP can be regarded as the same at any position on the trellis, the FER is as follows.

Here, L _frame is the number of bits in the frame. Further, ΣPEP (e _p (underscore)) indicates the probability of error events occur in a location that is not specified in the trellis.

2.2 Method of Predicting FER by Computing PEP in Trellis Section Equivalent to Frame Size The processing according to the prior art to be compared with the operation of the FER prediction unit of the present embodiment as described above is as follows. There are some problems.

  That is, when the PEP differs depending on the position on the trellis, it is not appropriate to calculate the FER as in the above equation (3-2), and as a result, the prediction accuracy deteriorates.

  FIG. 6 is a conceptual diagram for explaining a problem when the PEP differs depending on the position on the trellis.

  First, as shown in FIG. 6, consider a process of executing mapping of multi-level modulation for each subcarrier after interleave processing is performed on a convolutionally encoded transmission signal by an interleaver.

  In FIG. 6, for example, 16QAM is assumed as the multi-level modulation, and only the mapping in the real axis direction is extracted and shown for simplicity of explanation.

  First, when the transmission signal is divided and mapped for each subcarrier, since the SNR differs depending on the mapped subcarrier, the PEP differs depending on the trellis position at the time of decoding the convolutional code on the receiving side. Can occur.

  In the case of multi-level modulation, there are bits that are difficult to error and bits that are likely to be erroneous, and the PEP may differ depending on the mapped bit position.

  For example, when 2-bit signals “00”, “01”, “11”, “10” are considered, the left quadrant is mapped when the quadrant to be mapped is “0” according to the value of the first bit (see FIG. Middle, solid line) and “1” is divided into the right quadrant (dotted line in the figure). As a result, the first bit error is unlikely to occur in the demapping process on the receiving side.

On the other hand, with respect to the second bit, when “1”, the two signal points are adjacent to each other across the origin, whereas when “0”, the above “ In this case, the two signal points are opposed to each other with the two signal points in the case of 1 ″. For this reason, in the mapped space (constellation), the distance between signal points due to bit inversion is smaller in the second bit than in the first bit, so the second bit is the first bit. Compared to, errors are likely to occur.
That is, for example, in a communication system such as a wireless LAN, the SNR for each subcarrier is different and multilevel modulation is used (the error rate differs depending on the bit position in the mapped symbol), so the PEP differs depending on the position on the trellis. .

  When the PEP differs depending on the position on the trellis in this way, the PEP is calculated by considering the position on the trellis in the trellis section corresponding to the frame size in order to obtain the FER as in the following equation. The FER is predicted by taking the sum of the obtained PEPs.

  Therefore, specifically, the FER is calculated by the following calculation.

Here, in equation (4), L _frame means a section corresponding to the frame size. Therefore, equation (4) means that the FER is calculated by taking the sum of the PEP calculated for each frame position within the frame. However, PEP in the formula (4) The calculation of (e _{t, p} (underscore)), for example, the PEP in consideration to the position where the error pattern occurs on a trellis, such as Non-Patent Document 3 or 4 Calculation Use. In addition, when a PEP calculation method (for example, the following publicly known document 4) in which the position on the trellis cannot be considered is applied, the same PEP is substituted into equation (4) for all t for each error pattern. Thus, FER can be calculated. Therefore, equation (4) is not limited by the PEP calculation method.

Known Document 4: G. Caire, G. Taricco, and E. Biglieri, “Bit-interleaved coded modulation,” IEEE Trans. Inf. Theory, vol. 44, no. 3, pp. 927-946, May 1998.
FIG. 7 is a conceptual diagram showing the concept of equation (4).

  In FIG. 7, different error events that can occur at different positions in the trellis section corresponding to the frame size are shown with different thicknesses.

  Actually, for example, there are a plurality of error events starting from the position of t = 1.

  Therefore, to what range error events are considered is set in advance. Although not particularly limited, for example, an event to be considered is selected by considering only error events whose Hamming distance is a certain value or less.

In the following, thus, the true path, the Hamming distance error path, to be considered only error event such that the following distance d _H.

  For these error events, the PEP is calculated for each position in the frame and the sum is taken to determine the probability that the error event will occur.

  In the following, the calculation of PEP when the PEP is different according to the position of the trellis will be described.

First, for example, in the case described with reference to FIG. 3, the error pattern is [1, 1, 0, 1, 1, 1], which indicates an error path when the convolutional code has d _H = 5.

Hereinafter, it will be representative of the PEP error paths p-th error pattern starting at position t in the trellis PEP and (e _{t, p} (underscore)).

According to Non-Patent Document 4, a conditional PEP of a trellis path (hereinafter referred to as a true path) that generates codewords whose constituent bits are all 0 and a path (hereinafter referred to as an error path) that generates codewords separated by a Hamming distance d _H is Is represented by the following equation.

Error path e _{t, p} a (underscore) interleaves, the sub-carrier number w th error bit error pattern is mapped k _w, the intra-symbol bit positions for transmitting in the sub-carrier and i _w. The distance represented by Δ _w (i _w ) is the squared Euclidean distance between the signal point when there is no error and the signal point when the error bit is included for the symbol to which the error bit is assigned. H _k represents the frequency response of the k-th subcarrier, and H _kw is the frequency response of the subcarrier to which the w-th error bit is mapped. Further, H (under bar) and Δ (under bar) are vectors having H _kw and Δ _w (i _w ) as elements, respectively.

  In this case, PEP is given by the following equation by obtaining expected values for H (underbar) and Δ (underbar).

Here, E {...} _a represents the expected value calculation for a. In Equation (5), assuming that the propagation path changes for each transmission frame, an expected value is obtained for the frequency response H _kw of the subcarrier in order to derive a decoded FER when a plurality of frames are transmitted. Δ _w (i _w ) can be determined from the true path, error path, and interleaver structure. However, since the true path depends on the input bits of the encoder, Δ _w (i _w ) is a random variable. Become. Therefore, the expected value is obtained for Δ _w (i _w ) in equation (5). Furthermore, the conditional PEP in equation (5) is given by:

According to equation (5), in order to derive PEP, an expected value related to H _kw , Δ _w (i _w ) may be obtained for equation (6). In preparation for this, first, the numerator of formula (6) is modified as follows.

  Next, eigenvalue decomposition of PA′P is as follows.

Here, λ _l is a non-zero eigenvalue of PA′P having the l-th largest Euclidean norm, v _l is the l-th element of v, and its amplitude follows the Rayleigh distribution of the following probability density function.

  r is the diversity order. However, rank (...) represents the rank of the matrix, and min (...) represents the minimum value.

  Here, PEP can be said as follows.

  The transmitted bit string can be represented by one path on the trellis diagram. When a path other than the transmitted bit string is selected on the receiving side, a bit error after decoding occurs. The receiving side calculates the metric of each path and selects the path having the maximum metric. “PEP” is a probability that a metric of an error path exceeds a metric of a true path, and asymptotically approaches a probability that a bit string corresponding to the error path is obtained as a decoding result in a low FER region.

  Substituting equation (12) into equation (6) yields:

Similar to Non-Patent Documents 3 and 4, the following equation is obtained by obtaining an expected value for v _l .

Further, an expected value is obtained for Δ _w (i _w ).
Δ _w (i _w ) is a finite discrete random variable having a different probability density depending on the bit position i _w in the symbol, and λ _l is an eigenvalue of PA′P. Therefore, an error path e _{t, p} (underbar) Can be uniquely determined from Δ _w (i _w ).

Since Δ _w (i _w ) is a finite discrete random variable, λ _l is also a finite discrete random variable. Therefore, it becomes as follows.

Here, assuming that the possible values of the discrete random variable Δ _w (i _w ) are _{mi w} , the following relationship holds.

In a region where the FER is low, the SNR is sufficiently high (the amount of noise with respect to the inter-symbol distance is small), and it is expected that the symbol error is dominant in the case of an erroneous signal point.
Assuming that the minimum Euclidean distance between signal points is d _min , a term satisfying Δ _w (i _w ) = d _min ² with respect to w = 1,..., D _H is expected to be dominant among J terms. Is done. Further, P _j of this dominant term is expressed as P _min . That is, P _min represents the probability of Δ _w (i _w ) = d _min ² for w = 1,..., D _H and is determined from the modulation scheme, error path, and interleaver structure.

Λ _{l, min} is an eigenvalue of the matrix PA _{min ′} P calculated by the following equation.

  In this embodiment, the interleaver structure is known in advance on the transmitter side and the receiver side. Therefore, when executing the calculation of equation (15), the transmitter side has information on the power delay profile (necessary for calculating the matrix PA′P), noise power (necessary for calculating σ), and the configuration of the encoder configuration. If there is (that is, MCS information), it can be executed.

The total power portion of equation (15) decreases as the diversity order r increases, and the error path PEP with a large diversity order becomes lower than the error path PEP with a small diversity order. However, when a wireless LAN is assumed, there is a possibility that d _H ≧ L because delay dispersion is small. At this time, the diversity orders of all error paths are equally L, and there is a possibility that the PEPs of paths farther than the minimum free distance cannot be ignored even in a region where the FER is low. Therefore, an error path PEP that is a Hamming distance d _f + φ away from the true path is considered. Here, the value of φ is experimentally set to an appropriate value in advance.

  If the PEP can be predicted by the procedure as described above, the predicted value of FER by the FER prediction unit 3030 can be executed by the following formula.

However, the summation regarding p is performed for the case where the Hamming distance of the error path is equal to or less than d _f + φ.

According to the configuration of the present embodiment, in a communication system using a convolutional code, it is possible to realize a FER prediction apparatus that can accurately predict FER even in a situation where an error rate varies depending on the position of a trellis. In addition, by using such a FER prediction device, even when an error in a communication channel cannot be regarded as a random error, a wireless communication device and a wireless communication device that can perform communication by adaptively selecting an appropriate MCS A communication system can be realized.
[Modification of Embodiment 1]
In the above description, in the prediction of the frame error rate (FER), the PEP is calculated at each position on the trellis corresponding to the frame.

  However, for example, in a communication system such as a wireless LAN, the same interleaving is repeatedly executed in the same frame.

  In such a communication system, when it can be considered that the propagation path fluctuation is constant in the frame, the propagation path fluctuation received by the encoded sequence after deinterleaving is periodic with the interleave block length as a period.

  FIG. 8 is a conceptual diagram for explaining a PEP pattern when the same interleaving process is repeatedly performed on each block in the frame as described above.

As shown in FIG. 8, when the same interleaving process is repeatedly performed on each block in the frame, the influence of the propagation path received by each codeword error bit is periodic. As a result, the block error rate becomes equal in the interleave block period.
By utilizing this fact, it is possible to limit the interval for calculating the PEP to the interval of the trellis corresponding to one block, reduce the calculation cost, and predict the FER.

  FIG. 9 is a conceptual diagram for explaining the procedure for predicting the FER by the operation in units of blocks as described above.

  When a block is set so that the influence of the propagation path received by the encoded sequence after deinterleaving is periodic with the interleave block length as a period, PEPs having the same block period are repeated.

When the block period is L _block , the number of blocks in the frame is N _itl , and the PEP in the block is P _itl , FER is expressed as the following equation.

That is, in calculating the prediction of FER, it is sufficient to _obtain the pair-wise error rate P _itl of one block, and the prediction can be performed with less calculation cost than calculating PEP in the section corresponding to the frame size.

  For example, in the case of a wireless LAN, since propagation path fluctuation in a frame is small and interleaving is performed in units of OFDM symbols, it can be considered that PEPs are equal in a period corresponding to OFDM symbols. Therefore, the probability that an error occurs in each OFDM symbol can be regarded as the same, and FER prediction can be performed using the above formula.

For example, if the probability P _sym that one or more error events occur in an OFDM symbol is calculated, the FER of a system that performs OFDM communication is expressed by the following equation according to the following equation.

L _frame is one frame length, L _sym is the number of coding bits (= interleave size) in the OFDM1 symbol, and R represents the coding rate. The index on the right side of () represents the number of OFDM symbols for transmitting information bits for one frame.

However, more generally, the size of the interleave block is not limited to the OFDM symbol unit, and if the same interleave process is repeated over a plurality of interleave blocks within one frame, What is necessary is just to calculate PEP about a block.
[Embodiment 2]
In the first embodiment, the PEP is calculated based on the power delay profile. However, depending on the conditions of the propagation path, it is possible to calculate the PEP from the subcarrier frequency response (equivalent to the impulse response) as follows. In the second embodiment, a case where PEP is calculated from the frequency response of the subcarrier will be described.

Non-Patent Document 4 assumes that the frequency response of the propagation path does not vary within the OFDM symbol. In other words, it is allowed to vary from frame to frame or at shorter intervals. However, in a wireless LAN environment, it is assumed that the movement of the wireless device is somewhat small, and there may occur a case where the propagation path can be regarded as the same in a plurality of frames (changes in the propagation path can be ignored within the observation time). In such a case, the frequency response H _kw of the subcarrier (or the impulse response of the propagation path) can be regarded as a constant. Therefore, the conditional PEP can be calculated by treating the frequency response (or impulse response) of the subcarrier as a constant on the assumption that the frequency response (or impulse response) of the subcarrier does not change with time in Equation (6). In this case, it is only necessary to obtain an expected value for Δ _w (i _w ) as follows with respect to Equation (6).

  However, since the frequency response of the subcarrier and the impulse response of the propagation path can be equivalently converted as shown in Equation (2), Equation (17) can also be calculated from the impulse response.

Similar to the first embodiment, w = 1, ..., Δ w (i w) = d min 2 become claim respect d _f is expected to be dominant in the J terms. P _j of this dominant term is expressed as P _min . Therefore, Equation (17) is approximated as follows.

  In this way, a conditional PEP may be calculated. If such processing is performed, FER can be predicted by estimating the frequency response (or impulse response) of the subcarrier instead of the power delay profile.

  Other processes are the same as those described in the first embodiment, and thus description thereof will not be repeated.

  Accordingly, in this specification, terms are used as follows.

  “Propagation path characteristic information” refers to information representing attenuation of power in a propagation path, and typically means the power delay profile described above. Further, as described above, when the condition that the fluctuation of the propagation path can be ignored within the observation time is satisfied, instantaneous information on the propagation path estimated in the communication path, that is, each subcarrier in the convolutional coding OFDM system is used. Frequency response (equivalently, impulse response).

  As described above, even with the configuration of the present embodiment, it is possible to accurately predict the post-decoding BER and predict the frame error rate even when an error in the communication channel cannot be regarded as a random error. A frame error rate predicting device can be realized. Further, by using such a frame error rate prediction apparatus, even when an error in a communication channel cannot be regarded as a random error, a wireless communication apparatus capable of performing communication by adaptively selecting an appropriate MCS And a wireless communication system can be realized.

  Embodiment disclosed this time is an illustration of the structure for implementing this invention concretely, Comprising: The technical scope of this invention is not restrict | limited. The technical scope of the present invention is shown not by the description of the embodiment but by the scope of the claims, and includes modifications within the wording and equivalent meanings of the scope of the claims. Is intended.

  1000 transmitting apparatus, 1002 error correction coding section, 1004 interleaving section, 1010 modulation section, 1012 OFDM modulation section, 1014 RF section, 1020 antenna, 1100 reception processing section, 1110, 1110 ′ adaptive rate control section, 2000 receiving apparatus, 2002 Antenna, 2010 RF section, 2012 OFDM demodulation section, 2014 demodulation section, 2016 deinterleave section, 2018 error correction section, 2100 power delay profile / noise power estimation section, 2110 transmission processing section, 3010 BER prediction section, 3020, 3050 MCS storage Part, 3030 FER prediction part, 3040 MCS selection part.

Claims (11)

A frame error rate prediction apparatus of a communication system that communicates by a convolutional coding method that can be decoded by a trellis diagram, and that repeatedly performs the same interleaving process on transmission bits in a frame,
Storage means for storing information on a set of coding rate and modulation scheme of convolutional coding set in advance as employed in the convolutional coding scheme;
Based on the propagation path characteristic information representing the attenuation of propagation path power and noise power estimated in the communication path, the convolutional coding is performed on each of the coding rate and modulation scheme sets stored in the storage means. Is calculated at each position in the trellis diagram corresponding to the size of the frame, and the frame error rate is predicted as the sum of the pairwise error rates calculated at the positions. A frame error rate prediction apparatus comprising: a frame error rate prediction unit. In the frame, the frame error rate prediction means divides the transmission bit into a plurality of blocks for each block length set so that propagation path fluctuations for the encoded sequence after the deinterleaving process can be regarded as periodic. If it has, as before the block error rate calculated in a section of the trellis diagram corresponding to one of the Kivu lock as the sum of the pair-wise error rate, as the block error rate of each of the blocks are equal to each other, the block error The frame error rate prediction apparatus according to claim 1, wherein the frame error rate is predicted from a rate and the number of blocks included in the frame.   The frame error rate prediction apparatus according to claim 2, wherein the communication system performs communication by an orthogonal frequency division multiplexing system, and the block is one symbol in the orthogonal frequency division multiplexing system.   The frame error rate prediction apparatus according to claim 1, wherein the propagation path characteristic information is a power delay profile estimated in the communication path.   The frame error rate prediction apparatus according to claim 1, wherein the propagation path characteristic information is an impulse response of the propagation path. A wireless communication apparatus that communicates by a convolutional coding method that can be decoded by a trellis diagram and communicates by a convolutional coding method in which the same interleave processing is repeatedly performed on transmission bits in a frame ,
Transmitted from the receiving apparatus, receiving means for receiving the estimated value of the propagation path characteristics information representative of the attenuation of the communication route power and noise power as feedback information,
Storage means for storing information on a set of coding rate and modulation scheme of convolutional coding set in advance as employed in the convolutional coding scheme;
Selection means for selecting a set that maximizes throughput from a set of coding rates and modulation schemes stored in the storage means based on the feedback information;
The selection means includes
Based on the propagation path characteristic information and the noise power, a pairwise error rate of an error path in a decoding process for the convolutional coding is determined for each of the coding rate and modulation scheme sets stored in the storage unit. Frame error rate prediction means for calculating the frame error rate as a sum of the pairwise error rates calculated at each position of the trellis diagram section corresponding to the frame size;
Modulation scheme selecting means for selecting the set having the maximum throughput within a predetermined frame error rate based on the predicted frame error rate;
A wireless communication apparatus, further comprising: transmission means for performing transmission by performing convolutional coding and modulation processing on transmission data in accordance with the set selected by the modulation scheme selection means . In the frame, the frame error rate prediction means divides the transmission bit into a plurality of blocks for each block length set so that propagation path fluctuations for the encoded sequence after the deinterleaving process can be regarded as periodic. If it has, as before the block error rate calculated in a section of the trellis diagram corresponding to one of the Kivu lock as the sum of the pair-wise error rate, as the block error rate of each of the blocks are equal to each other, the block error The wireless communication apparatus according to claim 6, wherein the frame error rate is predicted from a rate and the number of blocks included in the frame. The radio communication apparatus according to claim 7, wherein the radio communication apparatus performs communication by an orthogonal frequency division multiplexing system, and the block is one symbol in the orthogonal frequency division multiplexing system.   The wireless communication apparatus according to claim 6, wherein the propagation path characteristic information is a power delay profile estimated in the communication path.   The wireless communication apparatus according to claim 6, wherein the propagation path characteristic information is an impulse response of the propagation path. A wireless communication system that can be decoded by a trellis diagram and communicates in a convolutional coding OFDM system in which the same interleaving process is repeatedly performed on transmission bits in a frame ,
A receiving device,
The receiving device is:
Communication path state estimation means for estimating propagation path characteristic information and noise power representing attenuation of propagation path power in the communication path;
First transmission means for transmitting the estimated propagation path characteristic information and noise power,
A transmission device;
The transmitter is
Receiving means for receiving, as feedback information, the propagation path characteristic information and the estimated value of the noise power transmitted from the receiving device;
Storage means for storing information of a set of coding rate and modulation scheme of convolutional coding set in advance as employed in the convolutional coding OFDM scheme;
Selection means for selecting, based on the feedback information, a set that maximizes throughput from a set of coding rates and modulation schemes stored in the storage means;
The selection means includes
Based on the propagation path characteristic information and the noise power, a pairwise error rate of an error path in a decoding process for the convolutional coding is determined for each of the coding rate and modulation scheme sets stored in the storage unit. Frame error rate prediction means for calculating the frame error rate as a sum of the pairwise error rates calculated at each position of the trellis diagram section corresponding to the frame size;
A modulation scheme selection means for selecting the set having the maximum throughput within a predetermined frame error rate based on the predicted frame error rate;
A wireless communication system further comprising second transmission means for performing transmission by performing convolutional coding and modulation processing on transmission data in accordance with the set selected by the modulation scheme selection means .