JP6452315B2 - amplifier - Google Patents

Description

  The present invention relates to an amplifier using a transistor.

  Conventionally, at a high-order frequency (so-called “harmonic frequency”) that is substantially an integer multiple of the operating frequency of a transistor, the impedance value (so-called “load impedance”) viewed from the transistor is a value that includes an appropriate reactance component. An amplifier having a harmonic processing circuit set to 1 is used. As the harmonic processing circuit, a parallel resonant circuit in which an inductor and a capacitor are connected in parallel is used (for example, see Patent Document 1).

  In the amplifier of Patent Document 1, a distributed constant line is connected to the input side of a transistor via a wire. This distributed constant line has an electrical length less than a quarter of a wavelength corresponding to a harmonic frequency (hereinafter referred to as “second harmonic frequency”) that is approximately twice the operating frequency of the transistor (so-called “electricity”). Long (electrical length) ". The distributed constant line performs the function of an inductor with a second harmonic frequency.

  A pair of comb-shaped electrodes facing each other are branched from the center of the distributed constant line through a gap. The comb electrode constitutes a so-called “interdigital capacitor” connected in parallel to the distributed constant line. A parallel resonant circuit is configured by such distributed constant lines and interdigital capacitors.

  The parallel resonance circuit is set to be in a parallel resonance state by using a second harmonic frequency of the operating frequency. In addition, at the second harmonic frequency, the parallel resonance circuit so that the value including the inductance component (inductive component) of the wire and the load impedance of the parallel resonance circuit is a value including an appropriate reactance component when viewed from the transistor. Is set. Thus, the efficiency of converting the supplied power into the output power (hereinafter referred to as “operation efficiency”) is improved by the wire and the parallel resonant circuit functioning as a harmonic processing circuit.

International Publication No. 2012/160810

  In general, a so-called “K factor” is used as an index indicating the stability of the amplification operation of a transistor. For example, when the transistor has a sufficiently high gain at a harmonic frequency and the value of the K factor is less than 1, a negative resistance component (so-called “negative resistance”) is locally generated by an unnecessary feedback component. The negative resistance causes the transistor to have a so-called “reflection gain”, and the amplitude of the signal reflected by the transistor to the harmonic processing circuit (hereinafter referred to as “reflection amplitude”) is amplified.

  On the other hand, in the conventional harmonic processing circuit, when the harmonic frequency is used, the load impedance becomes a load including only a reactance component (so-called “pure reactance load”). For this reason, the harmonic processing circuit has a reflection amplitude value of 1, and is in a state of reflecting all signals input from the transistor (so-called “total reflection state”). As a result, the conventional amplifier has a problem of causing unnecessary oscillation (so-called “parasitic oscillation”) between the harmonic processing circuit and the transistor.

  Further, the conventional harmonic processing circuit is constituted by a reactive parallel resonant circuit composed of an inductor and a capacitor. In this parallel resonant circuit, the reactance component included in the load impedance changes greatly according to the frequency, and so-called frequency dependency becomes high. Along with this, the phase of the signal reflected by the parallel resonant circuit to the transistor through the wire (hereinafter referred to as “reflection phase”) also becomes highly frequency dependent. As a result, there is a problem that the operation efficiency of the transistor can be maximized only in a narrow frequency band.

  The present invention has been made to solve the above-described problems, and provides an amplifier capable of improving the operation efficiency of a transistor over a wider frequency band and suppressing unnecessary oscillation. The purpose is to provide.

An amplifier according to the present invention includes an active element that amplifies an input signal, an inductor connected between an input terminal to which the input signal is input and the active element, and a capacitor connected in parallel to the inductor. A parallel resonant circuit set in a parallel resonant state by a harmonic frequency with respect to the operating frequency of the element and a resistance element connected in series with the capacitor are provided, and the capacitor is opposed to each other through a gap portion. It is composed of an interdigital capacitor composed of a pair of comb-shaped electrodes, and the resistance element is composed of a thin film resistor provided on one of the comb-shaped electrodes .

  According to the amplifier of the present invention, the operation efficiency of the transistor can be improved over a wider frequency band, and unnecessary oscillation can be suppressed.

It is a block diagram of the amplifier of Embodiment 1 of this invention. It is explanatory drawing which shows the equivalent circuit using the 2nd harmonic frequency of the amplifier of Embodiment 1 of this invention. It is a characteristic view which shows the operation efficiency with respect to the reflection phase of the amplifier of Embodiment 1 of this invention and the conventional amplifier. It is explanatory drawing which represented the reflection coefficient of the conventional amplifier on the Smith chart. It is explanatory drawing which represented on the Smith chart the reflection coefficient of the amplifier of Embodiment 1 of this invention. It is a characteristic view which shows the reflection phase with respect to the frequency of the amplifier of Embodiment 1 of this invention and the conventional amplifier. It is explanatory drawing which shows the equivalent circuit using the fundamental wave frequency of the amplifier of Embodiment 1 of this invention. It is explanatory drawing which represented on the Smith chart the impedance which looked at the transistor from the point X and the point Y shown in FIG. It is a block diagram of the other amplifier of Embodiment 1 of this invention. It is a block diagram of the amplifier of Embodiment 2 of this invention.

Embodiment 1 FIG.
With reference to FIG. 1, an amplifier according to a first embodiment of the present invention will be described.
In the figure, reference numeral 1 denotes a transistor (active element). The transistor 1 is composed of, for example, a field effect transistor (FET) whose source terminal is electrically grounded. The transistor 1 amplifies the amplitude of the signal input to the gate terminal and outputs it from the drain terminal.

  The gate terminal of the transistor 1 is connected to one end of the transmission line 3a via the wire 2a. The transmission line 3a is composed of a microstrip line made of a conductive foil provided on the surface of the dielectric substrate 4a. The transmission line 3a has an electrical length that is less than a quarter of a wavelength corresponding to a harmonic frequency (second harmonic frequency) that is approximately twice the operating frequency of the transistor 1. An input terminal 5 is provided at the other end of the transmission line 3a.

  A pair of comb-shaped electrodes 6a and 6b branches off from the center of the transmission line 3a. The comb-shaped electrodes 6a and 6b are opposed to each other so that the tooth portions 61a and 61b are engaged with each other through a gap portion. The interdigital capacitors connected in parallel to the transmission line 3a are constituted by the comb electrodes 6a and 6b.

  A parallel resonant circuit is configured by the transmission line 3a and the interdigital capacitor. The parallel resonance circuit is set to be in a parallel resonance state with a harmonic frequency composed of a second harmonic frequency of the operating frequency of the transistor 1.

  Of the pair of comb electrodes 6a and 6b, a thin film resistor 7 is provided on a part of the spine 62a of the comb electrode 6a closer to the transistor 1. The wire 2a, the parallel resonance circuit, and the thin film resistor 7 constitute a harmonic processing circuit.

  The drain terminal of the transistor 1 is connected to one end of the transmission line 3b through the wire 2b. The transmission line 3b is composed of a microstrip line made of a conductive foil provided on the surface of the dielectric substrate 4b. An output terminal 8 is provided at the other end of the transmission line 3b. In this way, the amplifier 100 is configured.

Next, the operation of the amplifier 100 at the second harmonic frequency will be described with reference to FIGS.
FIG. 2 shows an equivalent circuit using a second harmonic frequency of the operating frequency of the amplifier 100. Two inductors Lw1 and L1 are connected in series between the gate terminal of the transistor 1 and the input terminal 5. The inductance component (inductive component) of the inductor Lw1 corresponds to the inductance component of the wire 2a. The inductance component of the inductor L1 corresponds to the inductance component of the transmission line 3a.

  A capacitor C is connected in parallel to the inductor L1. A resistive element R is connected in series with the capacitor C. The capacitance component (capacitance component) of the capacitor C corresponds to the capacitance component of the interdigital capacitor. The resistance component of the resistance element R corresponds to the resistance component of the thin film resistor 7.

  One end of the capacitor Cp1 is connected between the inductor Lw1 and the inductor L1. The other end of the capacitor Cp1 is electrically grounded. In addition, one end of a capacitor Cp2 is connected between the inductor L1 and the input terminal 5. The other end of the capacitor Cp2 is electrically grounded. The capacitance components of the capacitors Cp1 and Cp2 correspond to the so-called “shunt” capacitance component generated between the transmission line 3a and the comb-shaped electrodes 6a and 6b and the conductor foil provided on the back surface of the dielectric substrate 4a. doing.

  Here, an equivalent circuit in which the resistance component of the resistance element R shown in FIG. 2 is 0 ohm (Ω) is an equivalent circuit of an amplifier (hereinafter referred to as “conventional amplifier”) excluding the thin film resistor 7 shown in FIG. ing.

  Hereinafter, the amplitude of the signal of the second harmonic frequency reflected by the transistor 1 to the harmonic processing circuit is expressed as “reflection amplitude | ΓTR |”, and the phase of the signal of the second harmonic frequency reflected by the transistor 1 to the harmonic processing circuit is indicated. This is expressed as “reflection phase ∠ΓTR”. The reflection coefficient ΓTR has an absolute value represented by the value of the reflection amplitude | ΓTR | and a phase angle represented by an angle of the reflection phase ∠ΓTR.

  Further, the amplitude of the second harmonic frequency signal reflected by the parallel resonant circuit to the transistor 1 via the wire 2a is represented as “reflection amplitude | ΓR |”, and the parallel resonant circuit reflects 2 to the transistor 1 via the wire 2a. The phase of the signal of the harmonic frequency is expressed as “reflection phase ∠ΓR”. The reflection coefficient ΓR has an absolute value represented by the value of the reflection amplitude | ΓR | and a phase angle represented by an angle of the reflection phase ∠ΓR.

  First, a signal having a frequency equivalent to the operating frequency of the transistor 1 (hereinafter referred to as “fundamental wave frequency”) is input to the input terminal 5. This input signal passes through the parallel resonance circuit and the wire 2a and is input to the gate terminal of the transistor 1. The transistor 1 amplifies the amplitude of the input signal and outputs it from the drain terminal.

  Transistor 1 then reflects the signal at the second harmonic frequency. This reflected signal is input to the parallel resonant circuit via the wire 2a. As a result, the parallel resonance circuit enters a parallel resonance state. The reflection coefficient ΓR is a reflection coefficient in an electrically open state, and the angle of the reflection phase ∠ΓR is approximately 0 °.

  Here, the reflection phase ∠ΓTR with respect to the reflection phase ∠ΓR is rotated by the reactance component of the capacitor Cp1 and the inductor Lw1. The angle of the reflection phase ∠ΓTR is an angle slightly larger than 180 ° (hereinafter referred to as “maximum efficiency phase angle”).

  FIG. 3 is a characteristic diagram showing the operating efficiency η of the transistor 1 with respect to the reflection phase ∠ΓTR. The horizontal axis indicates the value [°] of the reflection phase ∠ΓTR, and the vertical axis indicates the value [%] of the operating efficiency η. The characteristic line I is obtained by removing the thin film resistor 7 shown in FIG. 1 (that is, by setting the resistance component of the resistance element R shown in FIG. 2 to 0Ω), and the conventional amplifier having the reflection amplitude | ΓTR | The operating efficiency η of The characteristic line II shows the operating efficiency η of the amplifier 100 in which the reflection amplitude | ΓTR | is 0.7 by providing the thin film resistor 7.

  In general, when the angle of the reflection phase ∠ΓTR is slightly larger than 180 °, the load impedance of the transistor such as an FET slightly includes a capacitance component (capacitance component) and is electrically short-circuited. It becomes a state. For this reason, as shown in FIG. 3, the value of the operating efficiency η of the transistor 1 becomes the maximum value ηmax at the maximum efficiency phase angle.

  At this time, the value of the reflection amplitude | ΓTR | becomes less than 1 by providing the thin film resistor 7 (resistive element R shown in FIG. 2) shown in FIG. Thereby, even when the transistor 1 has a negative resistance, unnecessary oscillation can be suppressed by setting the value obtained by integrating the reflection gain to the reflection amplitude | ΓTR |

  Further, the provision of the thin film resistor 7 makes it difficult for the reflection phase ∠ΓR to change with respect to the frequency, and the frequency dependency is reduced. Accordingly, the frequency dependence of the reflection phase ∠ΓTR is also reduced. As a result, the frequency band in which the operating efficiency η of the transistor 1 is improved is widened, and the operating efficiency η of the transistor 1 can be improved over a wider frequency band.

Next, the effect of the second harmonic frequency of the amplifier 100 will be described with reference to FIGS.
FIG. 4 is an explanatory diagram showing the reflection coefficient ΓTR and the reflection coefficient ΓR of a conventional amplifier on a Smith chart. Here, the resistance component of the resistance element R is 0Ω, the capacitance component of the capacitor C is 0.56 picofarad (pF), the inductance component of the inductor L1 is 0.20 nanohenry (nH), and the inductance component of the inductor Lw1 is 0.20 nH. The capacitance component of the capacitor Cp1 is 0.36 pF, and the capacitance component of the capacitor Cp2 is 0.37 pF. The frequency of the input signal is 14 to 16 gigahertz (GHz), and the termination resistance is 50Ω.

  FIG. 5 is an explanatory diagram showing the reflection coefficient ΓTR and the reflection coefficient ΓR of the amplifier 100 on a Smith chart. Here, the resistance component of the resistance element R is 2.3Ω, the capacitance component of the capacitor C is 0.56 pF, the inductance component of the inductor L1 is 0.20 nH, the inductance component of the inductor Lw1 is 0.20 nH, and the capacitance component of the capacitor Cp1 is The capacitance component of 0.36 pF and capacitor Cp2 is 0.37 pF. The frequency of the input signal is 14 to 16 GHz, and the termination resistance is 50Ω.

  4 and 5, the length between the center O of the Smith chart and the reflection coefficient ΓTR corresponds to the value of the reflection amplitude | ΓTR |. The length between the central portion O and the reflection coefficient ΓR corresponds to the value of the reflection amplitude | ΓR |.

  The value of the reflection amplitude | ΓTR | of the amplifier 100 shown in FIG. 5 is lower than the value of the reflection amplitude | ΓTR | of the conventional amplifier shown in FIG. Therefore, even when the transistor 1 has a negative resistance, unnecessary oscillation can be suppressed by setting the value obtained by integrating the reflection gain to the reflection amplitude | ΓTR |

  FIG. 6 is a characteristic diagram showing the reflection phase ∠ΓTR with respect to the frequency of the input signal. The horizontal axis indicates the frequency value [GHz], and the vertical axis indicates the value [°] of the reflection phase ∠ΓTR. A characteristic line III indicates the reflection phase ∠ΓTR of the conventional amplifier, and a characteristic line IV indicates the reflection phase ∠ΓTR of the amplifier 100.

  As shown in FIG. 6, the frequency dependence of ∠ΓTR of the amplifier 100 is lower than that of 従 来 ΓTR of the conventional amplifier. Therefore, the operation efficiency η of the transistor 1 can be improved over a wider frequency band.

  Note that, as shown in FIG. 3, in a transistor such as an FET, the maximum value ηmax of the operating efficiency η decreases as the value of the reflection amplitude | ΓTR | decreases. However, the amplifier 100 reduces the frequency dependence of the reflection phase ∠ΓTR, so that the average value of the operating efficiency η in the entire predetermined frequency band is improved as compared with the conventional amplifier.

  At the second harmonic frequency, the same effect can be obtained even if a thin film resistor is provided on the other comb electrode 6b instead of the thin film resistor 7 provided on the comb electrode 6a.

Next, with reference to FIG. 7, the operation of the amplifier 100 at the fundamental frequency will be described.
FIG. 7 shows an equivalent circuit using the fundamental frequency of the operating frequency of the amplifier 100. The circuit elements similar to the equivalent circuit of the second harmonic frequency shown in FIG.

  At the fundamental frequency, the capacitance component between the pair of comb electrodes 6a and 6b can be ignored. For this reason, the capacitor C shown in FIG. 2 is omitted.

  On the other hand, in the case of the fundamental frequency of the operating frequency, each comb-shaped electrode 6a, 6b functions as a so-called “tip open stub”. That is, one end of the capacitor Cp1 'is connected to the junction a between the inductor Lw1 and the inductor L1 via the resistance element R. The other end of the capacitor Cp1 'is electrically grounded. The capacitance component of the capacitor Cp1 'includes the capacitance component of the capacitor Cp1 shown in FIG. 2 and the capacitance component due to the open-ended stub of the comb electrode 6a.

  Further, one end of the capacitor Cp2 'is connected to the junction b between the inductor L1 and the input terminal 5. The other end of the capacitor Cp2 'is electrically grounded. The capacitance component of the capacitor Cp2 'includes the capacitance component of the capacitor Cp2 shown in FIG. 2 and the capacitance component due to the open-ended stub of the comb electrode 6b.

Next, the effect of the fundamental frequency of the amplifier 100 will be described with reference to FIGS.
FIG. 8 shows Smith impedance Zx as seen from the point X between the inductor Lw1 and the junction a shown in FIG. 7 and impedance Zy as seen from the point Y between the junction b and the input terminal 5. It is explanatory drawing represented on the chart. In the Smith chart, the smaller the value of the resistance component included in the impedance, the closer to the left side of the axis (resistance axis) that crosses the center of the Smith chart.

  Generally, the value of the input impedance of a transistor is about several ohms. Therefore, the value of the impedance Zx when the transistor 1 is viewed only through the inductor Lw1 connected in series is a relatively small value. In FIG. 8, the impedance Zx is expressed in the vicinity of the left end portion of the resistance axis.

  On the other hand, the value of the impedance Zy when the transistor 1 is viewed through the inductors L1 and Lw1 connected in series and the capacitors Cp1 'and Cp2' connected to the shunt is larger than the value of the impedance Zx. In FIG. 8, the impedance Zy is represented closer to the center O of the Smith chart than the impedance Zx.

  As described above, as the number of circuit elements interposed between the transistor 1 and the transistor 1 increases, the impedance value viewed from the transistor 1 increases. Therefore, as the circuit portion connecting the resistance element R to the shunt is moved away from the transistor 1, the influence of the resistance element R increases and the passage loss of the harmonic processing circuit at the fundamental frequency increases.

  On the other hand, the amplifier 100 is provided with the thin film resistor 7 on the comb electrode 6a closer to the transistor 1 among the pair of comb electrodes 6a and 6b. Thereby, in the equivalent circuit of the fundamental frequency, the resistance element R is connected in series with the capacitor Cp1 'closer to the transistor 1 than the capacitor Cp2'. As a result, it is possible to reduce the passage loss of the harmonic processing circuit at the fundamental frequency and to minimize the decrease in the gain of the amplifier 100, as compared with an amplifier in which the other comb-shaped electrode 6b is provided with a thin film resistor.

  As described above, the amplifier 100 according to the first embodiment is provided with the thin film resistor 7 in a part of the interdigital capacitor constituting the parallel resonant circuit. Thereby, even when the transistor 1 has a negative resistance at the second harmonic frequency, unnecessary oscillation can be suppressed by setting the value obtained by integrating the reflection gain to the reflection amplitude | ΓTR | Further, since the frequency dependence of the reflection phase ∠ΓTR is reduced, the operating efficiency η of the transistor 1 can be improved over a wider frequency band.

  Further, the thin film resistor 7 is provided on the comb electrode 6a closer to the transistor 1 among the pair of comb electrodes 6a and 6b constituting the interdigital capacitor. As a result, it is possible to reduce the passage loss of the harmonic processing circuit at the fundamental frequency and to minimize the decrease in the gain of the amplifier 100, compared to an amplifier in which the other comb-shaped electrode 6b is provided with a thin film resistor.

  The input terminal 5 and the output terminal 8 may be connected to impedance matching circuits that match impedances at the fundamental frequency.

  The transistor 1 may be an active element that amplifies the amplitude of the input signal, and is not limited to the FET. A bipolar transistor or an integrated circuit (IC) that performs a function equivalent to that of a transistor may be used.

  In addition, the parallel resonant circuit is set to be in a parallel resonant state with a harmonic frequency that is approximately three times the operating frequency of the transistor 1 or a harmonic frequency that is approximately n times (n is an integer of 4 or more). Also good.

  Further, as shown in FIG. 9, the amplifier 101 may further include a pair of comb electrodes 6c and 6d branched from the center of the transmission line 3b and a thin film resistor 7a provided on one comb electrode 6c. The transmission line 3b may have an electrical length that is less than a quarter length of the wavelength corresponding to the harmonic frequency. The harmonic processing circuit composed of the wire 2b, the transmission line 3b, the comb electrodes 6c and 6d, and the thin film resistor 7a is a harmonic composed of the wire 2a, the transmission line 3a, the comb electrodes 6a and 6b, and the thin film resistor 7. It operates in the same way as the processing circuit.

Embodiment 2. FIG.
With reference to FIG. 10, an amplifier using a parallel resonant circuit having a shape different from that of the first embodiment will be described. The same components as those of the amplifier 100 according to the first embodiment shown in FIG.

  The gate terminal of the transistor 1 is connected to one end of the transmission line 3c via the wire 2a. The transmission line 3c is composed of a meander line line formed by bending a thin wire electrode into a rectangular wave shape.

  A pair of metal electrodes 6e and 6f are connected to the center of the transmission line 3c. An insulator layer 9 is interposed between the metal electrodes 6e and 6f. The metal electrodes 6 e and 6 f and the insulator layer 9 constitute a so-called “metal-insulator-metal (MIM) capacitor” connected in parallel to the transmission line 3 c. A parallel resonance circuit is configured by the transmission line 3c and the MIM capacitor.

  Of the pair of metal electrodes 6e and 6f, a thin film resistor 7 is connected between the transmission line 3c and the metal electrode 6e connected to a portion closer to the transistor 1 of the transmission line 3c. The wire 2a, the parallel resonance circuit, and the thin film resistor 7 constitute a harmonic processing circuit.

  The drain terminal of the transistor 1 is connected to one end of the transmission line 3d through the wire 2b. The transmission line 3d is a meander line formed by bending a thin wire electrode into a rectangular wave shape.

The amplifier 102 configured as described above operates in the same manner as the amplifier 100 of the first embodiment as follows.
That is, at the second harmonic frequency, the value of the reflection amplitude | ΓTR | becomes less than 1 by providing the thin film resistor 7 in the harmonic processing circuit. Thereby, even when the transistor 1 has a negative resistance, unnecessary oscillation can be suppressed by setting the value obtained by integrating the reflection gain to the reflection amplitude | ΓTR |

  Further, since the thin film resistor 7 is provided in the harmonic processing circuit, the reflection phase ∠ΓR is less likely to change with respect to the frequency, and the frequency dependency is reduced. Accordingly, the frequency dependence of the reflection phase ∠ΓTR is also reduced. As a result, the operating efficiency η of the transistor 1 can be improved over a wider frequency band.

Further, the thin film resistor 7 is connected to the metal electrode 6e closer to the transistor 1 among the pair of metal electrodes 6e and 6f. For this reason, in the equivalent circuit of the fundamental frequency, the resistance element R is connected in series to the capacitor Cp 1 ′ closer to the transistor 1 than to the capacitor Cp 2 ′. As a result, it is possible to reduce the passage loss of the harmonic processing circuit at the fundamental frequency and to minimize the decrease in the gain of the amplifier 102, compared to an amplifier in which a thin film resistor is connected to the other metal electrode 6f.

  The thin film resistor 7 may be provided on a part of one metal electrode 6e.

  In the present invention, within the scope of the invention, any combination of the embodiments, or any modification of any component in each embodiment, or omission of any component in each embodiment is possible. .

  1 transistor (active element), 2a, 2b wire, 3a, 3b, 3c, 3d transmission line, 4a, 4b dielectric substrate, 5 input terminal, 6a, 6b, 6c, 6d comb electrode, 6e, 6f metal electrode, 7, 7a Thin film resistor, 8 output terminal, 9 insulator layer, 61a, 61b tooth, 62a spine, 100, 101, 102 amplifier, C, Cp1, Cp2, Cp1 ′, Cp2 ′ capacitor, L1, Lw1 inductor, R resistive element.

Claims (8)

An active element for amplifying the input signal;
An inductor connected between the input terminal to which the input signal is input and the active element, and a capacitor connected in parallel to the inductor, and parallel resonance by a harmonic frequency with respect to the operating frequency of the active element A parallel resonant circuit set to a state;
A resistance element connected in series to the capacitor;
Equipped with,
The capacitor is composed of an interdigital capacitor composed of a pair of comb-shaped electrodes opposed to each other via a gap,
2. The amplifier according to claim 1, wherein the resistance element includes a thin film resistor provided on any one of the comb electrodes . The thin film resistors, a pair of claim 1, wherein the amplifier, characterized in that provided in a part of the interdigital electrode closer to the active element of the comb electrodes. An active element for amplifying the input signal;
An inductor connected between the input terminal to which the input signal is input and the active element, and a capacitor connected in parallel to the inductor, and parallel resonance by a harmonic frequency with respect to the operating frequency of the active element A parallel resonant circuit set to a state;
A resistance element connected in series to the capacitor;
Equipped with,
The capacitor comprises a metal insulator metal capacitor having an insulator layer interposed between a pair of metal electrodes,
2. The amplifier according to claim 1, wherein the resistance element includes a thin film resistor connected to any one of the metal electrodes . 4. The amplifier according to claim 3 , wherein the thin film resistor is connected to the metal electrode closer to the active element of the pair of metal electrodes. The said inductor was comprised with the transmission line which has an electrical length less than 1/4 length of the wavelength corresponding to the said harmonic frequency, The any one of the Claims 1-4 characterized by the above-mentioned. Amplifier. 6. The amplifier according to claim 5 , wherein the transmission line is a meander line formed by bending a thin wire electrode. It said parallel resonant circuit, according to any one of claims 1 to 6, characterized by comprising setting the parallel resonance state by the harmonic frequency twice or three times the operating frequency amplifier. The active element is composed of a field effect transistor having a source terminal electrically grounded,
The amplifier according to any one of claims 1 to 7 , wherein the inductor is connected to a gate terminal of the field effect transistor.

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