JP6221824B2 - Control device for power converter - Google Patents

Description

The present invention relates to a technical field of a control device for a power converter.

  An apparatus for detecting a ripple superimposed on a motor current is disclosed (see Patent Document 1). According to the ripple detection device disclosed in Patent Document 1, the digital data of the motor current is input to the digital filter, and the filter coefficient of the digital filter is changed to an optimum value for the ripple detection, thereby improving the ripple detection. It is said that you can do it.

  As a technique for switching the constants of a digital filter, there has been proposed a technique that uses integral control to suppress output instability (see Patent Document 2). As a technique for suppressing the output instability when switching the constants of the digital filter, a technique has been proposed in which all numerical values in the input and the filter are made zero (see Patent Document 3).

JP 2009-207236 A JP-A-11-112289 JP-A-2-252307

  For example, a configuration in which a power converter such as a boost converter is interposed between an electric load including a three-phase AC motor and an inverter and a DC power supply is widely known. In such a configuration, the output voltage signal of the power converter is used in a feedback manner for control of the power converter.

  On the other hand, a ripple accompanying driving of a motor that is a part of the electric load is superimposed on the output voltage signal. In recent years, the size of passive elements such as supersaturated reactors and smoothing capacitors in power converters tends to be reduced, and the influence of ripples on output voltage signals is relatively large.

  Here, in order to mitigate the influence of this ripple, in recent years, speeding up of the control of the power converter (that is, shortening of the control cycle) has been sought. That is, an effect of reducing the amplitude of the ripple component can be expected by speeding up the control (shortening the control cycle). For example, speeding up the control means speeding up about 10 times the current level.

  However, according to the applicant's research, the ripple is superimposed on the power supply current on a small scale as a new problem due to the size reduction of the passive elements and the accompanying shortening of the control cycle of the power converter. It was issued. When ripples are superimposed on the power supply current, the power supply performance may be affected, for example, the deterioration of the power supply is relatively accelerated. Therefore, in order to shorten the control cycle, it is necessary to remove ripples from the output voltage signal of the power converter.

  Here, the optimization of the signal by the digital filter is effective not only for the ripple detection disclosed in Patent Document 1, but also for the ripple removal.

  However, the ripple accompanying the driving of the three-phase motor is composed of many ripple components having different frequency bands, and the frequency band is diverse. Therefore, if filter constants are assigned to the respective ripple components one by one, the configuration of the digital filter becomes complicated, and the configuration of the control system of the power converter becomes complicated.

  That is, the conventional technique has a technical problem that it is difficult to shorten the control cycle of the power converter without complicating the control system of the power converter.

  This invention is made | formed in view of the technical problem which concerns, and makes it a subject to provide the control apparatus of the power converter which can shorten the control period of a power converter with a simple structure.

  In order to solve the above-described problems, a power converter control device according to the present invention controls a power converter that is electrically connected between an electric load including a three-phase AC motor and a DC power supply. A digital filter to which an output voltage signal of the power converter is inputted and a filter constant corresponding to a cut-off frequency can be switched; and the filter constant is: (1) control of the three-phase AC motor Setting based on the carrier frequency of the electric load when the mode is the PWM mode, and (2) Setting based on the electric sixth frequency of the three-phase AC motor when the control mode is the rectangular wave mode. Means (claim 1).

  The power converter control device according to the present invention is a device that controls a power converter such as a boost converter, and includes a digital filter and setting means such as a controller and a computer device. The power converter is electrically interposed between an electric load composed of, for example, a three-phase AC motor and a motor driving inverter, and a DC power source such as a secondary battery. The “three-phase AC motor” according to the present invention is a concept including a motor and a motor generator.

  According to the applicant's research, the ripple superimposed on the voltage signal for controlling the power converter with the driving of the three-phase AC motor is the dominant ripple component among the plurality of ripple components. It varies depending on the control mode of the AC motor.

  More specifically, when a three-phase AC motor is driven in a PWM (Pulse Width Modulation) mode, a ripple component corresponding to the carrier frequency of the electric load becomes dominant. Further, when the three-phase AC motor is driven in the rectangular wave mode, the ripple component corresponding to the electric sixth frequency of the three-phase AC motor becomes dominant.

  The power converter control device according to the present invention has a configuration in which the filter constant that defines the cutoff frequency of the digital filter is set based on the control mode of the three-phase AC motor based on the applicant's knowledge. It has become. That is, the setting means sets the filter constant based on the carrier frequency of the electric load when the control mode of the three-phase AC motor is the PWM mode. The setting means sets the filter constant based on the electrical sixth-order frequency of the three-phase AC motor when the control mode of the three-phase AC motor is the rectangular wave mode. “Electric sixth-order frequency” means the frequency of the sixth-order harmonic of the fundamental wave corresponding to one electrical angle cycle of the three-phase AC motor.

  According to the control device for the power converter according to the present invention, the filter constant of the digital filter is selectively set according to the control mode of the three-phase AC motor. Therefore, it is not necessary to configure the digital filter as a multistage filter including a plurality of filters corresponding to each of the ripple components. Further, since the filter constant is set corresponding to the most dominant ripple component in each case, the ripple is effectively reduced from the output voltage signal (so-called VH signal) of the power converter input to the digital filter. Can be removed. In other words, the ripple component can be effectively removed with a simple configuration, and the control cycle of the power converter can be shortened.

  Such an operation and other advantages of the present invention will become apparent from the embodiments described below.

1 is a system diagram of a motor drive system according to an embodiment of the present invention. It is a block diagram of the control apparatus in the motor drive system of FIG. It is a flowchart of the constant selection process performed by the constant selection part in the control apparatus of FIG.

<Embodiment of the Invention>
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
<1. Configuration of Embodiment>
First, the configuration of a motor drive system 10 according to an embodiment of the present invention will be described with reference to FIG. FIG. 1 is a system configuration diagram conceptually showing the configuration of the motor drive system 10.

  In FIG. 1, the motor drive system 10 includes a control device 100, a boost converter 200, an inverter 300, a smoothing capacitor C, a DC power source B, and a three-phase AC motor M1. The motor drive system 10 is mounted on a vehicle (not shown).

  The control device 100 is an electronic control device configured to be able to control the operation of the motor drive system 10. The control device 100 includes an HVECU 110, an MGECU 120, and a boost control device 130 which will be described later. A detailed configuration of the control device 100 will be described later.

  Boost converter 200 is an example of a “power converter” according to the present invention that includes a reactor L1, switching elements Q1 and Q2, and diodes D1 and D2.

  Reactor L1 has one end connected to a positive electrode line (not shown) connected to the positive electrode of DC power supply B, and the other end is an intermediate point between switching element Q1 and switching element Q2, that is, an emitter terminal of switching element Q1. And a connection point with the collector terminal of the switching element Q2.

  The switching elements Q1 and Q2 are connected in series between the positive electrode line and a negative electrode line (reference numeral omitted) connected to the negative electrode of the DC power source B, and the collector terminal of the switching element Q1 is connected to the positive electrode line. The emitter terminal of the switching element Q2 is connected to the negative electrode line. The diodes D1 and D2 are rectifying elements that allow only current from the emitter side to the collector side in each switching element.

  The switching elements Q1 and Q2 and each switching element (Q3 to Q8) of the inverter 300 described later are configured as, for example, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, or the like.

  The smoothing capacitor C is a voltage smoothing capacitor connected between the positive electrode line and the negative electrode line. The voltage between the terminals of the smoothing capacitor C, that is, the voltage between the positive electrode line and the negative electrode line is the output voltage of the boost converter 200 (uniquely, the input voltage of the inverter 300).

  The inverter 300 includes a U-phase arm (not shown) including a p-side switching element Q3 and an n-side switching element Q4, a V-phase arm (not shown) and a p-side switching element including a p-side switching element Q5 and an n-side switching element Q6. The power control device includes a W-phase arm (reference numeral omitted) including Q7 and the n-side switching element Q8, and constitutes an example of the “electric load” according to the present invention together with the three-phase AC motor M1. Each arm of the inverter 300 is connected in parallel between the positive electrode line and the negative electrode line.

  The switching elements Q3 to Q8 are connected to rectifying diodes D3 to D8 that allow current to flow from the emitter side to the collector side, similarly to the switching elements Q1 and Q2. Further, intermediate points between the p-side switching element and the n-side switching element of each phase arm in inverter 300 are connected to the respective phase coils of three-phase AC motor M1.

  The DC power source B is a rechargeable secondary battery device in which a plurality of various secondary battery cells such as nickel metal hydride batteries and lithium ion batteries are connected in series. As the DC power source B, an electric double phase capacitor, a large capacity capacitor, a flywheel, or the like may be used instead of or in addition to this type of secondary battery.

  The three-phase AC motor M1 is a three-phase AC motor generator in which a permanent magnet is embedded in a rotor. The three-phase AC motor M1 is mechanically connected to drive wheels of a vehicle (not shown) and is configured to generate torque for driving the vehicle. The three-phase AC motor M1 can also perform power regeneration (power generation) in response to an input of kinetic energy of the vehicle, for example, during braking of the vehicle. When this vehicle is a so-called hybrid vehicle, the three-phase AC motor M1 is mechanically connected to an engine (not shown), and can perform power regeneration by the power of the engine or assist the power of the engine. . The vehicle may be a so-called electric vehicle including only a rotating electric machine including the three-phase AC motor M1 as a power source.

  The motor drive system 10 is provided with a sensor group (not shown). By this sensor group, for example, the power supply voltage value Vb of the DC power supply B, the load current value IL of the reactor L1 in the boost converter 200, the output voltage value VHs of the boost converter 200, the v-phase current value Iv and the w-phase current value in the inverter 300 Iw, the motor rotation angle θ of the three-phase AC motor M1, the vehicle state value Sys, and the like are appropriately detected. The vehicle state value Sys is a comprehensive concept of a plurality of state values indicating the state of the vehicle. The vehicle state value Sys includes, for example, an accelerator opening value and a vehicle speed value. If the vehicle is equipped with an engine, various state values of the engine are also included. Each of the sensors constituting the sensor group is electrically connected to the control device 100, and the various sensor output values are appropriately referred to by the control device 100.

  Next, a detailed configuration of the control device 100 will be described with reference to FIG. FIG. 2 is a block diagram of the control device 100. In the figure, the same reference numerals are given to the same portions as those in FIG. 1, and the description thereof will be omitted as appropriate.

  In FIG. 2, control device 100 includes HVECU 110, MGECU 120, and boost control device 130.

  The HVECU 110 is an electronic control device that comprehensively controls the operation of the entire vehicle. The HVECU 100 is a computer system that includes, for example, one or a plurality of CPUs (Central Processing Units). The HVECU 110 appropriately includes a volatile memory that temporarily stores sensor output values, control values, and the like, and a nonvolatile memory that stores control maps.

  The MGECU 120 is an electronic control device that controls the operation of the three-phase AC motor M1 through drive control of the inverter 300 (that is, control of the switching state of each switching element). The MGECU 120 is electrically connected to the HVECU 110, and the operation is controlled by the HVECU 110.

Boost control device 130 is an example of a “power converter control device” according to the present invention that controls the operation of boost converter 200. The boost control device 130 includes an A / D converter 131, a digital filter 132, a voltage control unit 133, a current control unit 134, a gate generation unit 135, a constant selection unit 136, and a constant calculation unit 137. Details of each part of the boost control device 130 will be described in the section of operation of the following embodiment.
<2. Operation of Embodiment>
Next, the operation of the embodiment will be described.

<2-1. Control mode of three-phase AC motor M1>
In the motor drive system 10, a PWM control mode corresponding to a known PWM control and a rectangular wave control mode corresponding to a known rectangular wave control are used as control modes of the three-phase AC motor M1. These are operated under the control of the MGECU 120.

  The PWM control is current feedback control, and is a control for supplying a PWM signal to the three-phase AC motor M1 for each of the U phase, the V phase, and the W phase by comparing the voltage command value and the carrier (carrier wave).

  Specifically, in PWM control, a two-phase current command value (Idtg, Iqtg) is generated based on the torque command value TR of the three-phase AC motor M1 determined by the HVECU 110 based on the vehicle state value Sys. On the other hand, the three-phase current value is converted from the v-phase current value Iv and the w-phase current value Iw supplied as feedback information from the sensor into a two-phase current value consisting of the d-axis current value Id and the q-axis current value Iq. The Then, based on the difference between the two-phase current command values (Idtg, Iqtg) and the two-phase current values Id and Iq, a two-phase voltage command value composed of the d-axis voltage value Vd and the q-axis voltage value is generated. The The generated two-phase voltage command values Vd and Vq are converted into three-phase voltage command values Vu, Vv and Vw.

  Various known modes can be applied to the method for determining the torque command value TR. For example, when the vehicle is an electric vehicle that does not have an engine as a power source, the torque command value TR may be determined according to a vehicle request output that is determined based on the vehicle speed and the accelerator opening, for example. . Further, when the vehicle is a hybrid vehicle having an engine as a power source, for example, the torque command is further taken into consideration in consideration of the thermal efficiency of the engine, the power output ratio of both based on the SOC of the DC power supply B, etc. The value TR may be determined.

  When a three-phase voltage command value is obtained, the converted three-phase voltage command values Vu, Vv, and Vw and a carrier signal value that changes at a carrier frequency FCar (for example, variable within a range of several tens to several hundreds of kHz). The magnitude relationship with CAR is compared. Then, U-phase switching signals Gup and Gun, V-phase switching signals Gvp and Gvn, and W-phase switching signals Gwp and Gwn whose logic states change according to the comparison result are generated and supplied to the inverter 300.

  In comparison between the carrier signal value CAR and each phase voltage command value, when each phase voltage command value matches the carrier signal value CAR from a value smaller than the carrier signal value CAR, a switching signal for turning on the p-side switching element is obtained. Generated. Further, when each phase voltage command value matches the carrier signal value CAR from a value larger than the carrier signal value CAR, a switching signal for turning on the n-side switching element is generated. That is, the switching signal is a signal that is turned on and off, and one of the p-side and n-side switching elements is always on and the other is off. When the inverter 300 changes or is maintained in the driving state of each switching element defined by each phase switching signal, the three-phase AC motor M1 is driven according to the circuit state corresponding to the changed or maintained driving state. The The PWM control is executed in this way, for example.

  The rectangular wave control is a control for supplying a one-pulse switching signal to the three-phase AC motor M1 according to the motor electrical angle of the three-phase AC motor M1. In the rectangular wave control, the voltage amplitude value is fixed to the maximum value, and torque is fed back by phase control.

  The control mode is switched between the PWM control mode and the rectangular wave control mode according to, for example, the motor rotation speed value MRN indicating the rotation speed of the three-phase AC motor M1 and the torque command value TR of the three-phase AC motor M1. Is called. In this case, for example, the MGECU 120 exchanges the motor rotation speed value MRN obtained by time-processing the input motor rotation angle θ and the torque command value TR supplied from the HVECU 110 with the control mode of the three-phase AC motor M1. One control mode corresponding to the motor rotation speed value MRN and the torque command value TR at that time is selected from the control map associated with.

  The MGECU 120 supplies the boost control device 130 with a control mode information value mmod indicating whether the selected control mode is the PWM control mode or the rectangular wave control mode. Further, the MGECU 120 supplies the motor rotation speed value MRN and the carrier frequency value fcar indicating the value of the carrier frequency Fcar to the boost control device 130.

<2-2. Overview of Control of Boost Converter 200>
The operation state of boost converter 200 is controlled by boost controller 130. Specifically, boost converter 200 generates a voltage between the positive electrode line and the negative electrode line, that is, an output voltage VH that is a voltage across the terminals of smoothing capacitor C, based on signal PWC supplied from boost controller 130. The voltage can be boosted above the output voltage of the DC power supply B.

  At this time, if the output voltage VH is lower than the target voltage, the on-duty of the switching element Q2 is relatively increased, and the current flowing through the positive line from the DC power supply B side to the inverter 300 side can be increased. VH can be raised. On the other hand, if the output voltage VH is higher than the target voltage, the on-duty of the switching element Q1 is relatively increased, and the current flowing through the positive line from the inverter 300 side to the DC power source B side can be increased, and the output voltage VH Can be reduced.

<2-3. Details of Boost Controller 130>
Next, the configuration and operation of the boost control device 130 will be described with reference to FIG.

  In FIG. 2, an A / D converter 131 is a converter that converts an analog signal into a digital signal. The A / D converter 131 receives an output voltage value VHs of the boost converter 200 supplied from the sensor. The output voltage value VHs is sampled by the A / D converter 131 and converted into an output voltage sampling value VHssmp which is a discrete sampling value (digital data value). The output voltage sampling value VHssmp is input to the digital filter 132.

  The digital filter 132 is a band attenuation filter whose cutoff frequencies fc1 and fc2 (fc2> fc1) change according to a filter constant FCfix described later. The output voltage sampling value VHssmp input to the digital filter 132 is subjected to band attenuation processing according to the filter constant FCfix by the digital filter 132, and is output as the output voltage filter value VHsft. The output voltage filter value VHsft is input to the voltage control unit 133.

  The voltage control unit 133 is a controller including an inverter input calculation unit, an adder / subtracter, a voltage control calculation unit, a carrier generation unit, and a sampling hold circuit.

  The inverter input calculation unit generates a VH command value VHtg representing the target value of the output voltage VH of boost converter 200. The VH command value VHtg generates the VH command value VHtg based on, for example, the output value of the three-phase AC motor M1 calculated from the torque command value TR of the three-phase AC motor M1 and the motor rotation speed value MRN. The motor rotation speed value MRN and the torque command value TR are supplied from the HVECU 110 and the MGECU 120.

  In the addition / subtraction unit, the output voltage filter value VHsft which is the output value of the digital filter 132 is subtracted from the VH command value VHtg, and the subtraction result is output to the voltage control calculation unit.

  When receiving the subtraction result from the addition / subtraction unit, the voltage control calculation unit calculates a current command value IR for making the output voltage VH of the boost converter 200 coincide with the VH command value VHtg.

  On the other hand, in the sampling hold circuit, the load current value IL is sampled at the timing of the peak and valley of the carrier signal value generated by the carrier generation unit. By sampling the load current value IL at the timing of the peak and valley of the carrier signal value, the average load current value IL in which the fluctuation of the load current value IL is canceled can be specified.

  The voltage control calculation unit subtracts the sampled and held load current value IL from the current command value IR to obtain the load current control value IL '. The load current control value IL ′ is supplied to the current control unit 134.

  The current control unit 134 calculates the duty command value duty as a control amount for making the load current value IL coincide with the current command value IR based on the load current control value IL ′ supplied from the voltage control unit 133. At this time, for example, a known PI control calculation including a proportional term (P term) and an integral term (I term) is used. The calculated duty command value duty is supplied to the gate generation unit 135.

  In the gate generation unit 135, the magnitude relationship between the duty command value duty and the carrier signal is compared, and a signal PWC including gate signals SG1 and SG2 whose logic states change according to the magnitude relationship is generated.

  Gate signal SG1 is a signal that turns on or off switching element Q1 of boost converter 200, and gate signal SG2 is a signal that turns on or off switching element Q2 of boost converter 200. Both are signals defined so that both the switching elements Q1 and Q2 are not turned on at the same time. The generated signal PWC is supplied to the boost converter 200, and the on-duty of each switching element is controlled by each gate signal.

  The constant selection unit 136 selects the filter constant FCfix of the digital filter 132 and sets the selected filter constant FCfix as the filter constant of the digital filter 132. Selection of the filter constant FCfix by the filter selection unit 136 is realized by a constant selection process described later.

  The constant calculation unit 137 calculates the first filter constant FCc and the second filter constant FC6 as options necessary for the selection of the filter constant FCfix by the constant selection unit 136.

  The first filter constant FCc is a filter constant based on a carrier frequency FCar which is a signal frequency of the carrier signal value CAR for driving the inverter 300 (uniquely the three-phase AC motor M1). More specifically, the first filter constant FCc is experimentally determined in advance so that a relationship of fc1 <Fcar <fc2 is established between the cutoff frequencies fc1 and fc2 of the digital filter 132 and the carrier frequency Fcar. Set in an empirically or theoretically established way. That is, when the first filter constant FCc is set in the filter constant FCfix of the digital filter 132, the ripple component corresponding to the carrier frequency FCar of the inverter 300 is removed from the output voltage value VHs of the boost converter 200. The carrier frequency FCar is an example of the “carrier frequency of electric load” according to the present invention.

  The second filter constant FC6 is a filter constant based on the electric sixth-order frequency Fm6 of the three-phase AC motor M1. More specifically, the second filter constant FC6 is previously set so that a relationship of fc1 <Fm6 <fc2 is established between the cutoff frequencies fc1 and fc2 of the digital filter 132 and the electrical sixth-order frequency Fm6. It is set in an experimentally, empirically or theoretically established manner. That is, when the second filter constant FC6 is set to the filter constant FCfix of the digital filter 132, the ripple component corresponding to the electric sixth-order frequency Fm6 of the three-phase AC motor M1 is removed from the output voltage value VHs of the boost converter 200. The The electric sixth frequency Fm6 of the three-phase AC motor M1 is calculated from the motor rotational speed MRN supplied from the MGECU 120.

<2-4. Details of constant selection process>
Next, the details of the constant selection process will be described with reference to FIG. FIG. 3 is a flowchart of the constant selection process.

  In FIG. 3, the constant selection unit 136 acquires a control mode information value mmod indicating the control mode of the three-phase AC motor M1 from the MGECU 120 (step S110). The control mode information value mmod takes either “pwm” indicating that the three-phase AC motor M1 is driven in the PWM control mode or “sqr” indicating that it is driven in the rectangular wave control mode. .

  When the control mode information value mmod is acquired, it is determined whether or not the acquired control mode information value mmod is “pwm” (step S120). When the control mode information value mmod is “pwm” (step S120: YES), that is, when the three-phase AC motor M1 is driven in the PWM control mode, the constant selection unit 136 uses the filter constant FCfix of the digital filter 132. The first filter constant FCc is selected as the filter constant FCfix of the digital filter 132 (step S130).

  On the other hand, when the control mode information value mmod is not “pwm” (step S120: NO), that is, when the control mode information value mcmd is “sqr” (that is, the three-phase AC motor M1 is driven in the rectangular wave control mode. The constant selection unit 136 selects the second filter constant FC6 as the filter constant FCfix of the digital filter 132, and sets it as the filter constant FCfix of the digital filter 132 (step S140).

  When step S130 or S140 is executed, the constant selection process ends, and after a predetermined period, the process returns to step S110 again. The constant selection process is executed as described above.

<2-5. Effect of constant selection processing>
According to the applicant's research, the dominant ripple component among the ripples derived from the three-phase AC motor M1 superimposed on the output voltage value VHs of the boost converter 200 depends on the control mode of the three-phase AC motor M1.

  That is, when the control mode of the three-phase AC motor M1 is the PWM control mode, the most dominant (influenced) component of the ripple superimposed on the output voltage value VHs is a ripple component corresponding to the carrier frequency FCar. is there. Further, when the control mode of the three-phase AC motor M1 is the rectangular wave control mode, the most dominant (influenced) component of the ripple superimposed on the output voltage value VHs is the electric 6 of the three-phase AC motor M1. It is a ripple component corresponding to the next frequency Fm6.

  In the motor drive system 10, based on this research result, when the three-phase AC motor M1 is driven in the PWM control mode, the filter constant FCfix of the digital filter 132 is set to the first filter constant FCc based on the carrier frequency Fcar. The Further, when the three-phase AC motor M1 is driven in the rectangular wave control mode, the filter constant FCfix of the digital filter 132 is set to the second filter constant FC6 based on the electric sixth-order frequency Fm6 of the three-phase AC motor M1.

  Therefore, the ripple superimposed on the output voltage value VHs can be effectively removed by the digital filter 132, and the control cycle of the boost converter 200 is shortened as the reactor L1 of the boost converter 200 and the smoothing capacitor C are reduced in size. It is possible to prevent the ripple from being superimposed on the load current IL (that is, the output current of the DC power supply B), which is generated in the case where the current is changed.

  On the other hand, the digital filter 132 is a single filter (or several filters including other filters such as LPF as appropriate) that operates only with the filter constant FCfix set according to the control mode of the three-phase AC motor M1. Has a simple configuration. For example, under the above-described technical idea that is not based on the applicant's research results, the filter corresponding to each ripple component one by one with respect to the superposition of many ripple components on the load current IL resulting in shortening of the control cycle. There is no effective measure other than preparing a plurality of filters having constants. That is, it is inevitable that the digital filter 132 is multistage and complicated.

  In the motor drive system 10, the function of the boost control device 130 can avoid such multi-stages and complications. Therefore, it is possible to prevent the ripple from being superimposed on the load current IL under a simple configuration and thereby boost the voltage. Shortening of the control cycle of converter 200 can be realized.

  The present invention is not limited to the above-described embodiments, and can be appropriately changed without departing from the spirit or concept of the invention that can be read from the claims and the entire specification. The control device is also included in the technical scope of the present invention.

  DESCRIPTION OF SYMBOLS 10 ... Motor drive system, 100 ... Control apparatus, 110 ... HVECU, 120 ... MGECU, 130 ... Boost control apparatus, 200 ... Boost converter, 300 ... Inverter, C ... Smoothing capacitor, B ... DC power supply, M1 ... Three-phase AC motor .

Claims (1)

A control device for a power converter that controls a power converter electrically connected between an electric load including a three-phase AC motor and a DC power source,
A digital filter to which an output voltage signal of the power converter is input and a filter constant corresponding to a cutoff frequency can be switched;
The filter constant is set based on (1) a carrier frequency of the electric load when the control mode of the three-phase AC motor is a PWM mode, and (2) when the control mode is a rectangular wave mode. A control device for a power converter, comprising: setting means for setting based on an electrical sixth frequency of a three-phase AC motor.

Images