JP6040565B2 - Multi-phase power conversion circuit - Google Patents

Description

  The present invention relates to a multiphase power conversion circuit that performs a unidirectional operation of boosting or stepping down between a low voltage side having one or a plurality of power supplies and a high voltage side of a common bus, or performing a bidirectional operation.

In high-power DC-DC converter circuits, when high-frequency switching is difficult due to restrictions on the switching devices used, an interleaved circuit configuration may be used to improve the quality of the input current and output voltage of the converter circuit. In an interleaved DC-DC converter circuit, the current of each phase may become unbalanced due to variations in elements such as switches and coils constituting the circuit of each phase. Therefore, as shown in the example of the n-phase interleaved bidirectional DC-DC converter circuit shown in FIG. 12 or the three-phase interleaved bidirectional DC-DC converter circuit shown in FIG. Current control is performed using current values i1, i2, i3,... In obtained from the sensor. For example, there is a method disclosed in Non-Patent Document 1 for the current balance control of the inductor of each phase related to the three-phase interleaved boost chopper circuit.
Further, in a circuit having a plurality of power supplies and having a common high-voltage bus, conventionally, the current at the input portion of each power supply is generally detected using an individual current sensor.

  By the way, in an inverter for driving an AC motor, a method for detecting a value of a DC link current by a current sensor or a shunt resistor and obtaining a three-phase current is proposed in Patent Document 1 or the like. In the case of a general AC motor, estimation is easy because the sum of three-phase currents becomes zero on connection.

Takahiro Kawashima, Masayoshi Yamamoto, Shigeyuki Tsuji, Mamoru Tsuruya: “Current Balance Control of Interleaved Chopper Circuits”, 2006 China Federation of Electrical and Information Society, p.484

JP 2003-219678 A

In the interleave type DC-DC power conversion circuit, even if the switches of each phase are driven with the same duty ratio, the current of each phase is unbalanced due to variations in the characteristics of the elements constituting the circuit. As a countermeasure against this, a method has been proposed in which the current flowing in each phase is detected and controlled by a current sensor (see FIGS. 12 and 13). However, this method has a problem that the cost increases by attaching a current sensor to each phase.
Also, in applications where a plurality of voltage sources are connected and the high voltage side is used as a common bus, a plurality of current sensors are required, which also increases the cost.

  In order to solve this problem, a method of reproducing a three-phase AC current by considering the DC link current proposed in Patent Document 1 and the conduction / non-conduction state of the switch as a method for driving the AC motor can be considered. However, in a circuit having an interleaved DC-DC converter circuit and a plurality of power supplies and a common high-voltage bus, unlike a general AC motor, the sum of three-phase currents is not always zero. Therefore, the method of Patent Document 1 cannot be applied as it is.

  The present invention has been made to solve the above problems, and an object of the present invention is to control the current of each phase of a multiphase power conversion circuit using a single current sensor.

  The multi-phase power conversion circuit of the present invention is configured by connecting output terminals of converters of respective phases to each other, and boosts or reduces between a low voltage side having one or a plurality of power supplies and a high voltage side of a common bus. A step-down unidirectional operation or a bidirectional operation is performed. Each phase converter includes an inductor for each phase, one end of which is connected to a low-voltage power source, an upper switch for each phase, and a lower switch connected in series to the upper switch, and at the midpoint between the upper switch and the lower switch. The other end of the inductor is connected to serve as the input end of each phase converter, and both ends of the upper and lower switches connected in series are commonly connected to serve as the output end of each phase converter. A controller is provided for inputting the output current ih and the output voltage vh measured on the high voltage side and outputting a drive signal to the gates of the upper switch and the lower switch. The controller includes a generator that outputs a sampling signal and a carrier wave synchronized with the sampling signal, and an inductor current estimation that estimates each phase output current flowing in the inductor by sampling and calculating the output current ih based on the generated sampling signal. And a PWM circuit that inputs the output voltage vh, each phase estimated output current, and a carrier wave, and adjusts the on / off phase of each of the upper switch and the lower switch for each switch.

  According to the present invention, the current of each phase can be controlled independently with only one current sensor. For example, even when there are variations in the characteristics of the circuit elements constituting each phase, it is possible to perform control to maintain the current balance of each phase with only one current sensor. By sampling the value of the current sensor connected to the high voltage side in synchronization with the carrier wave of each phase (for example, at the peak of the carrier wave or at the peak of the peak and valley), the current value of each phase is obtained from that value. The effect that the current sensor of each phase becomes unnecessary is acquired.

It is a figure which illustrates the n phase interleaved bidirectional DC-DC converter circuit comprised as a 1st example of the multiphase power converter circuit of this invention. It is a figure which illustrates the three-phase interleave bidirectional DC-DC converter circuit comprised as the 2nd example of the multiphase power converter circuit of this invention. FIG. 3 is a circuit diagram illustrating a controller for a three-phase interleaved bidirectional DC-DC conversion circuit. It is a figure which illustrates carrier waves Vcarr1, Vcarr2, Vcarr3, and sampling signal smpl generated by shifting the phase by 120 (= 360/3) degrees by the carrier wave and sampling signal generator. It is a figure explaining the estimation principle (when the duty ratio of an upper switch is 2/3 or less) of an inductor current (estimated output current). It is a figure explaining the estimation principle of an inductor current when there is no part which overlaps with the current waveform on the high voltage side and the inductor current waveform of each phase. It is a figure which illustrates the bidirectional | two-way DC-DC converter circuit which has two power supplies comprised as a 3rd example of the multiphase power converter circuit of this invention. It is a circuit diagram which illustrates the controller for bidirectional DC-DC conversion circuits which has a plurality of power supplies. It is a graph which shows the example of estimation of an inductor current (when the duty ratio of an upper switch is 0.5). 6 is a graph showing an example of estimating an inductor current (estimation method 1 when the duty ratio of the upper switch is 0.7). It is a graph which shows the example of estimation of an inductor current (when the duty ratio of an upper switch is 0.7, estimation method 2). It is a figure which shows the n phase interleave bidirectional | two-way DC-DC converting circuit by a prior art. It is a figure which shows the three-phase interleave bidirectional DC-DC converting circuit by a prior art.

  Hereinafter, the present invention will be described based on examples. FIG. 1 is a diagram illustrating an n-phase interleaved bidirectional DC-DC converter circuit configured as a first example of the multiphase power converter circuit of the present invention. Although a circuit configuration capable of bidirectional operation will be described as an example, the present invention is not limited to a circuit configuration capable of bidirectional operation as long as the inductor current is continuous, and only unidirectional (step-up or step-down) operation. It is applicable to a power conversion circuit capable of Further, the present invention is applicable to an n-phase (n is an integer of 2 or more including 2) multi-phase power conversion circuit.

  As shown in FIG. 1, the inductor current of each phase (current flowing in each phase inductor L1, L2, L3,... Ln) is estimated using a value ih obtained by a current sensor installed on the high voltage side of the circuit. . The on / off signal of each switch of the n-phase interleaved power conversion circuit can be generated by PWM (pulse width modulation) control. At this time, a triangular wave is generally used as a carrier wave. In addition, carrier waves of each phase are shifted from each other by 360 / n degrees.

  Hereinafter, a description will be given by taking a three-phase circuit as an example in the case of n = 3. FIG. 2 is a diagram illustrating a three-phase interleaved bidirectional DC-DC conversion circuit configured as a second example of the multiphase power conversion circuit of the present invention. In FIG. 2, the inductor L1, the lower switch S1d, and the upper switch S1u form a first phase converter. Inductor L2, lower switch S2d, and upper switch S2u form a second phase converter. Inductor L3, lower switch S3d, and upper switch S3u form a third-phase converter. The input terminals and the output terminals of the first to third layer converters are connected to each other to form an interleaved converter. However, the multi-phase power conversion circuit of the present invention detects the current flowing therethrough by connecting the output terminals of each phase in common, but as will be described later with reference to FIG. The input terminals are not necessarily connected in common.

  One end of each of the inductors (reactors) L1 to L3 of the phases 1 to 3 is connected to one end of the low voltage side power supply. The other ends of the inductors L1 to L3 are respectively connected to the midpoints of the upper switches S1u, S2u, S3u and the lower switches S1v, S2v, S3v connected in a three-phase bridge. These switches are constituted by switching elements such as MOSFETs and free-wheeling diodes, for example. An output capacitor C and a load R are connected to the high-voltage side end of the three-phase bridge. The output current ih and the output voltage vh detected on the high voltage side of the three-phase bridge are led to a controller (see FIG. 3) described later. This controller generates drive signals for driving the upper switches S1u, S2u, S3u and the lower switches S1v, S2v, S3v, and outputs them to the gates of the corresponding switches.

  FIG. 3 is a circuit diagram illustrating a controller for a three-phase interleaved bidirectional DC-DC conversion circuit. The controller shown in the figure performs output voltage control of a three-phase interleaved bidirectional DC-DC converter circuit and equalization control of current of each phase. The controller inputs the output current ih and the output voltage vh, and outputs a drive signal to the gates of the switching elements constituting the three-phase bridge. The controller includes a voltage controller AVR (for example, a PI controller, but not limited to PI, P control and other control methods can be applied), an inductor current estimator, a carrier wave and sampling signal generator, a current controller ACR ( For example, a PI controller (but not limited to PI, P control and other control methods can be applied), a limiter, a PWM circuit (comparator), and a switch drive circuit including a dead time generation circuit are provided.

  The voltage controller AVR amplifies the error between the output voltage vh and the reference voltage Vh * and outputs an error amplification signal to the current controller ACR (PI controller). Since the error amplification signal decreases when the high-voltage side output voltage vh rises and rises when it falls, the duty ratio of each switch is controlled so that the high-voltage side output voltage vh is stabilized. A current controller ACR, a limiter, and a comparator are provided for each phase. The error amplification signal from the voltage controller AVR is branched for each phase via a 1/3 divider if necessary, and the same error amplification signal is supplied to each phase current controller ACR as the current command value of each inductor. Output toward. The carrier wave and sampling signal generator outputs the sampling signal smpl to the inductor current estimator and also outputs a carrier wave (for example, a triangular wave) generated in synchronization with the sampling signal to each phase comparator.

  FIG. 4 is a diagram illustrating carrier waves Vcarr1, Vcarr2, Vcarr3 and sampling signal smpl generated by shifting the phase by 120 (= 360/3) degrees by the carrier wave and sampling signal generator. The inductor current estimator estimates each phase output current by sampling and calculating the output current ih based on the generated sampling signal smpl (details will be described later). The output current estimated for each phase is output to the current controller ACR.

  The current controller ACR provided for each phase inputs an error amplification signal from the voltage controller AVR and a difference signal between each phase estimation output current, amplifies the difference signal as necessary, and performs PI control and the like. Do. That is, the deviation signal is integrated and multiplied by the integral gain, and the deviation signal multiplied by the proportional gain is added and the value is output. The output voltage from the PI controller is limited to its upper limit value via a limiter. When the controller output voltage is exceeded, the upper switch is turned off (see the upper and middle stages of FIG. 5 described later), and a drive signal for turning on the lower switch is generated from the drive circuit. That is, PWM control is performed to control the pulse width at which the switch is turned on by the output voltage from the PI controller. The lower switches S1v, S2v, and S3v are signals obtained by turning on and off the upper switches S1u, S2u, and S3u, respectively. The switch drive circuit inserts a dead time between the lower switches S1v, S2v, S3v and the upper switches S1u, S2u, S3u, as necessary. This switch drive circuit generates a drive signal for driving the switch based on the comparison signal from the comparator and outputs it to the gate of the corresponding switch S1u, S2u, S3u, S1v, S2v, S3v).

  As shown in FIG. 4, since the carrier waves of each phase are shifted from each other by 360/3 degrees, when the characteristics of the circuit elements constituting each phase are equal and the values of the estimated output currents of each phase are the same, Although the drive signals have the same duty ratio and different phases by 360/3 degrees for each phase, the phase at which the switch is turned on and off is adjusted for each switch by the feedback output voltage vh and the estimated phase output current.

  When the three-phase interleaved bidirectional DC-DC converter circuit illustrated in FIG. 2 is boosted, when the lower switch of each phase is ON, the low-voltage DC voltage is applied to the inductor of each phase. Excitation current flows through. When the lower switch is turned off and the upper switch is turned on, current flows from the low-voltage side DC power source through each phase inductor and the upper switch to demagnetize the corresponding inductor. However, current may flow through the diode during the demagnetization period of the inductor.

  In the case of the step-down operation, contrary to the above, when the upper switch of each phase is on, the high-voltage side DC voltage is applied to the inductor of each phase, and an exciting current flows through the inductor of each phase. When the upper switch is turned off and the lower switch is turned on, a current flows through each phase inductor and the lower switch toward the low-voltage DC power supply side, and the corresponding inductor is demagnetized. However, current may flow through the diode during the demagnetization period of the inductor.

  FIG. 5 is a diagram for explaining the estimation principle of the inductor current (estimated output current) (when the duty ratio of the upper switch is 2/3 or less). As shown in the upper part of FIG. 5, in the case of three phases, carrier waves Vcarr1, Vcarr2, and Vcarr3 whose phases are shifted by 120 (= 360/3) degrees are used. Vcarr1, Vcarr2, and Vcarr3 are carriers corresponding to phases 1 to 3. Vref shown in the figure is a voltage command signal, but actually, as described above, a voltage obtained by finely adjusting the reference voltage Vh * by the output voltage vh and each phase estimated output current is compared with the carrier voltage. However, the voltage command signal Vref displayed in FIG. 5 is actually Vref1, Vref2, and Vref3 that are different for each phase. The on / off signals corresponding to the upper switches S1u, S2u, S3u of phases 1 to 3 are shown in the middle part of FIG. The lower switches S1v, S2v and S3v (illustration of the on / off signal is omitted) of the phases 1 to 3 are on / off signals obtained by inverting the upper switches S1u, S2u and S3u, respectively.

  The present invention samples the high-voltage side current in a phase synchronized with the carrier wave of each phase. This sampling is performed at least once in each period of the carrier wave for each phase. Ih1b, ih2b, and ih3b shown in the figure represent current values obtained by sampling the current value ih on the high voltage side at the timings indicated by the vertical lines in the figure. In the present invention, using the sampled current values ih1b, ih2b, and ih3b, the inductor current values i1, i2, and i3 flowing in the inductors L1, L2, and L3 of each phase are calculated and estimated. When the duty ratio of the upper switch is 2/3 = 0.66 or less, sampling at the valley portion of the carrier wave of each phase makes it easy to estimate the inductor current (estimated output current). As an example corresponding to this condition, there is a case where the voltage command signal Vref is 2/3 or less of the peak of the carrier wave.

  The high-voltage side current ih is the sum of the phase inductor currents i1, i2, and i3. At the valley timing of the carrier wave Vcarr1 in FIG. 5 (vertical line labeled ih1b), only the switch S1u is on and the switches S2u and S3u are off. Therefore, when sampling is performed at the timing ih1b of the valley portion of the carrier wave Vcarr1, only the inductor current i1 can be detected. Similarly, when sampling is performed at timings ih2b and ih3b of valley portions of the carrier waves Vcarr2 and Vcarr3, inductor currents i2 and i3 can be detected.

Thus, when the voltage command signal Vref is 2/3 or less of the peak of the carrier wave, there is a portion where the current waveform on the high voltage side and the inductor current waveform of each phase overlap. The current waveforms overlap each other when the carrier wave of each phase is a trough portion. Here, the inductor current of the phase and the current value on the high voltage side coincide with each other. At this timing, it can be confirmed that only the switch of the corresponding phase among the upper switches is conductive. Therefore, the current value on the high voltage side is sampled at the valley of the carrier wave, and the inductor current value of the phase can be detected. If the values obtained by the high-voltage current sensor sampled at the valleys of Vcarr1, Vcarr2, and Vcarr3 are ih1b, ih2b, and ih3b, respectively, the inductor current values i1, i2, and i3 of each phase are obtained by the following equations: .
i1 = ih1b (1)
i2 = ih2b (2)
i3 = ih3b (3)

In the case of n phase, if there is an overlap between the current waveform on the high voltage side and the inductor current waveform on each phase, if the value obtained by the high voltage side current sensor is ihkb, the inductor current ik of phase k is Can be obtained at
ik = ihkb (4)

  FIG. 6 is a diagram illustrating the principle of estimating the inductor current when there is no portion overlapping the current waveform on the high voltage side and the inductor current waveform of each phase. In the figure, the ON / OFF signal of each phase upper switch, the inductor current, and the high-voltage side current are shown. With reference to FIG. 6, a description will be given of a method for obtaining the inductor current value of each phase when the upper switches of all the phases are turned on in the valley portion of the carrier wave of each phase. As an example corresponding to this condition, in the case of a three-phase circuit, the voltage command signal Vref is 2/3 or more of the peak of the carrier wave. As can be seen from the inductor current of each phase shown in the figure and the current waveform on the high voltage side, there is no portion where the current values overlap (there is no timing at which each phase inductor current matches the current on the high voltage side), see FIG. Thus, the estimation method described above cannot be used. In this case, the following estimation method can be used.

(Estimation method 1)
In the carrier valley, all the upper switches are conducting. If the current on the high voltage side is detected at this time, the sum of inductance currents i1 + i2 + i3 of each phase is obtained. On the other hand, when sampling is performed at the peak portion of the carrier wave, the upper switches of the phases other than the phase corresponding to the peak portion of the carrier wave are turned on. For example, in the peak portion of the carrier wave of phase 1 (vertical line labeled ih1t), the upper switches S2u and S3u except for the switch S1u of phase 1 are in the conductive state. The sum of inductance currents (i2 + i3) excluding the inductance current i1 of 1 is obtained.

Therefore, the values obtained with the high-voltage side current sensors sampled at the valleys of Vcarr1, Vcarr2, and Vcarr3 are the values obtained with the high-voltage side current sensors sampled at ih1b, ih2b, ih3b, and the peaks, respectively. If ih1t, ih2t, and ih3t, respectively, the inductor current values i1, i2, and i3 of each phase are obtained by the following equations.
i1 = ih1b −ih1t (5)
i2 = ih2b −ih2t (6)
i3 = ih3b −ih3t (7)

Note that it is not always necessary to sample the current on the high-voltage side at the carrier valleys of all phases, and it may be used by sampling at any one of the carrier wave valleys (i1 + i2 + i3). If this value is ihb, the inductor current values i1, i2, and i3 of each phase are obtained by the following equations.
i1 = ihb −ih1t (8)
i2 = ihb −ih2t (9)
i3 = ihb −ih3t (10)

In the case of n phase, the inductor current of phase k can be obtained by the following equation.
ik = ihkb −ihkt (11)
Or
ik = ihb −ihkt (12)

(Estimation method 2)
As described above, when sampling is performed at the peak portion of the carrier wave, the switch on the upper side of the phase other than the phase corresponding to the peak portion of the carrier wave becomes conductive. If the values obtained by the high-voltage side current sensor sampled at the peak are ih1t, ih2t, and ih3t, respectively, the inductor current values i1, i2, and i3 of each phase can also be obtained by the following equations.
i1 = (ih2t + ih3t −ih1t) / 2 (13)
i2 = (ih1t + ih3t −ih2t) / 2 (14)
i3 = (ih1t + ih2t −ih3t) / 2 (15)

In the case of n phase, the inductor current of phase k can be obtained by the following equation.

  In the case of n phase, specifically, when a carrier wave having a phase difference of 180 ° when n> 2 is generated, only the method of sampling with a trough of the carrier wave with the command signal being 2 / n or less can be used. Specifically, it corresponds to 4 phases, 6 phases, and the like. The 5 and 7 phases do not violate the above conditions, but if the command signal is 2 / n or less, the valley only method can be used, and if it is (n-1) / n or more, A scheme is available.

(Estimation method 3)
Further, according to the present invention, as another method of estimating the inductor current, it is possible to sample at the center of on and off of the on / off signal of each switch, equivalent to the above-described triangular wave comparison. For this purpose, switches Su1, Su2, Su3, etc. sampled at regular intervals (for example, a sample signal is generated at a frequency that is 2 × the number of phases of the switching frequency, and the switch on / off signal is sampled at this timing by the sampler). It is also possible to detect the signal and the high-voltage side current ih and obtain the inductor current by calculation. (Su3, Su2, Su1) can be expressed as (000) to (111), where 1 is when the switch such as Su1, Su2, Su3 is on and 0 is when the switch is off. If this is considered in binary and converted to decimal, it corresponds to 0-7. For example, the inductor current connected to Su1 is a combination of 111 (7) and 110 (6) switches (7-6 = 1), a combination of 001 (1) and 000 (0) (1-0 = 1), It is possible to calculate with the difference of ih detected in the combination of 011 (3) and 010 (2) (3-2 = 1).

  For example, in the case of three phases, if phase is (001), only phase 1 is on, if phase is (010), only phase 2 is on, and if phase (100), only phase 3 is on. For example, the current of phase 1 can be calculated by (111) − (110) = (001) as described above so that this numerical value is obtained. This is the same as estimation method 1 or estimation method 2 described above, but when the currents in the patterns (101) and (100) are sampled, the current in phase 1 is (101)-(100) = ( 001). This is not only to use two timing values, but also to calculate the four timings of the multiple timings, ih so that the answer is (100) = 4, (010) = 2, (001) = 1 If the value of is calculated, the inductor current of each phase can be calculated. If it is a phase 2 switch, it is calculated by a combination of switches that are 2 when considered in decimal, and 4 in the case of phase 3. However, paying attention to a certain carrier wave, calculation is performed after 360 ° samples (that is, T [s] if the switching period of each switch is T [s]).

  FIG. 7 is a diagram illustrating a bidirectional DC-DC conversion circuit having a plurality of power supplies configured as a third example of the multiphase power conversion circuit of the present invention. The second example shown in FIG. 2 has one low-voltage side power supply, whereas the third example shown in FIG. 7 has two or more low-voltage power supplies Vs1 and Vs2. It differs in that it is configured. Similar to the second example shown in FIG. 2, the output current ih and the output voltage vh detected on the high voltage side connected to the common bus are led to the controller.

  FIG. 8 is a circuit diagram illustrating a controller for a bidirectional DC-DC conversion circuit having a plurality of power supplies. This controller generates drive signals for driving the upper switches S1u, S2u, S3u and the lower switches S1v, S2v, S3v, and outputs them to the gates of the corresponding switches, as in the example of FIG. 3 described above. However, in the second example shown in FIG. 2, the output voltage control and the current equalization control of each phase are performed, whereas in the third example shown in FIG. There is no need to be equal. When controlling the output voltage, the current command value of each inductor which is not necessarily equal for each phase is output from the voltage controller AVR in consideration of the ratio of how much power is supplied from each power supply. Based on the information, the current is controlled by the current controller ACR. The description of the operation of the other circuits of the controller shown in FIG. 8 is the same as that illustrated in FIG.

  The estimation method of the present invention was applied when a three-phase interleaved bidirectional DC-DC converter circuit (see FIG. 2) in which the equivalent series resistance of the phase 1 inductor is 1/5 times that of the other phase is in a step-up operation. The simulation results in this case are shown in FIG. 9, FIG. 10, and FIG. FIG. 9 is a graph showing an example of estimating the inductor current (when the duty ratio of the upper switch is 0.5). FIG. 10 is a graph showing an example of estimating the inductor current (estimation method 1 when the duty ratio of the upper switch is 0.7). FIG. 11 is a graph showing an example of estimating the inductor current (estimation method 2 when the duty ratio of the upper switch is 0.7). It can be seen that the inductor current can be estimated appropriately. Note that i1est, i2est, and i3est shown in the figure represent estimated current values of the respective phase inductors.

L1, L2, L3, Ln: Inductor for each phase
S1u, S2u, S3u, Snu: Upper switches for each phase
S1d, S2d, S3d, Snd: Lower switches for each phase
i1, i2, i3, in: Detected value of current flowing in inductor of each phase
ih: Detected value of current flowing on the high voltage side
Vcarr1, Vcarr2, Vcarr3: Carrier signal of each phase in PWM control
ih1b, ih2b, ih3b: Value of ih in the valley of the carrier wave of each phase
ih1t, ih2t, ih3t: The value of ih at the peak of the carrier of each phase

Claims (5)

By connecting the output terminals of the converters of each phase to each other, perform unidirectional operation of boosting or stepping down between the low voltage side with one or more power supplies and the high voltage side of the common bus, or bidirectional In the multi-phase power conversion circuit that operates,
Each of the phase converters includes an inductor for each phase, one end of which is connected to a low-voltage side power source, an upper switch for each phase, and a lower switch connected in series to the upper switch, and the middle of the upper switch and the lower switch. The other end of the inductor is connected to a point to be an input end of each phase converter, and both ends of an upper switch and a lower switch connected in series are commonly connected as an output end of each phase converter,
A controller that inputs the output current ih and the output voltage vh measured on the high-voltage side and outputs a drive signal to each gate of the upper switch and the lower switch,
The controller amplifies an error between the output voltage vh and the reference voltage Vh * and outputs an error amplification signal, and a voltage controller AVR that controls the duty ratio of each switch so that the high-voltage side output voltage vh is stabilized. And a generator that outputs a sampling signal and a carrier wave synchronized with the sampling signal, and an estimation method according to the duty ratio, and the output current ih is sampled and calculated by the generated sampling signal to flow to the inductor An inductor current estimator that estimates the output current of each phase, a phase signal that inputs the difference signal between the error amplification signal and the estimated output current of each phase from the voltage controller AVR, and outputs voltage command signals Vref1, Vref2, and Vref3 each phase current controller ACR with each, the voltage command signal Vref1, Vref2, Vref3 as compared with the carrier wave voltage output from the generator the upper switch And and a PWM circuit for adjusting for each switch each of the off phase of the lower switch,
A multi-phase power conversion circuit.
When there is a portion that overlaps the current waveform on the high voltage side and the inductor current waveform on each phase, the inductor current estimator samples the output current ih measured on the high voltage side at the valley portion of the carrier wave of each phase. The multi-phase power conversion circuit according to claim 1, wherein the obtained value is estimated to be an output current of each phase flowing through an inductor of each phase. When there is no overlapping part between the current waveform on the high voltage side and the inductor current waveform on each phase, the inductor current estimator samples the output current ih measured on the high voltage side at the valley portion of the carrier wave of each phase. When the obtained value is ihkb and the value obtained by sampling the output current ih measured on the high voltage side at the peak of the carrier wave of each phase is ihkt, in the case of n phase, the phase k The output current ik flowing through the inductor of
ik = ihkb −ihkt
Or the value obtained by sampling the output current ih measured on the high voltage side at the valley portion of the carrier wave of any one phase as ihb,
ik = ihb −ihkt
The multiphase power conversion circuit according to claim 1, which is estimated to be When there is no overlapping part between the current waveform on the high voltage side and the inductor current waveform of each phase, the inductor current estimator samples the output current ih measured on the high voltage side at the peak of the carrier wave of each phase. Assuming that the obtained value is ihkt, in the case of n phase, the output current ik flowing through the inductor of phase k is

The multiphase power conversion circuit according to claim 1, which is estimated to be The inductor current estimator detects the switch signal and the high-voltage side current ih sampled at the on center and the off center of the on / off signal of each switch, and calculates and outputs each phase output current flowing through the inductor of each phase. The multiphase power conversion circuit according to claim 1, wherein

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