JP6015647B2 - Control device for motor drive device and motor drive system - Google Patents

Description

  The present invention relates to a control device for an electric motor drive device and an electric motor drive system, for example, an electric motor drive device that applies an AC voltage to a synchronous motor.

  Non-Patent Document 1 describes a method for controlling a synchronous motor. The synchronous motor has an armature having windings and a field. In Patent Document 1, the primary magnetic flux of the synchronous motor is controlled. More specifically, the γ-axis and δ-axis that are orthogonal to each other are adopted as the control axes, and the γ-axis component of the primary magnetic flux is controlled to zero.

  In order to realize such control, in the voltage equation of the synchronous motor, when the zero is adopted as the γ-axis component of the primary magnetic flux and the command value is adopted as the δ-axis component, the γ-axis voltage and the δ-axis voltage are expressed as respective commands. Adopt as. Further, the δ-axis current is feedback-controlled to improve control accuracy. This point will be described in detail later.

  Moreover, patent documents 1-4 are shown as a technique relevant to this application.

Japanese Patent No. 3672761 Japanese Patent No. 4988374 Japanese Patent No. 3298267 Japanese Patent No. 4881038

Hamada, Yamamura, Tsunehiro, "General-purpose inverter for driving synchronous machines", IEEJ Transactions D, 1999, Vol.119, No.5, p.707-712

  However, the technique described in Patent Document 1 causes instability of control due to manufacturing variations of the synchronous motor, as will be described in detail later.

  Therefore, an object of the present invention is to provide a control technique for an electric motor drive device that contributes to stable control.

According to a first aspect of the control device for an electric motor drive device of the present invention, an AC voltage is applied to a synchronous motor (2) having an armature (21) including an armature winding (22) and a field (23). A device (3) for controlling the driving device (1) to apply and output an alternating current (iu, iv, iw), wherein the alternating current is directed in a predetermined advance direction with respect to the δ axis and the δ axis. And a δ-axis current (iδ) and γ in a rotating coordinate system having a phase (φ) with respect to the flux linkage (Λ0) to the armature by the field. A coordinate conversion unit (33) for converting into a shaft current (iγ), a product of the sum of the resistance value (R0) of the resistance component of the armature winding and a preset constant (kγγ) and the γ-axis current The primary magnetic flux command (Λδ *), which is a combination of the first term, the angular velocity (ω1) of the AC voltage, the linkage flux (Λ0), and the magnetic flux of the armature reaction generated by the AC current flowing through the armature. ) Calculate the sum ((R0 + kγγ) · iγ + L0 · ω1 · Iδ *) with the second term to be calculated, and calculate the γ-axis voltage command (vγ *) for the γ-axis voltage (vγ) of the AC voltage. A γ-axis voltage command generating unit (31) to be generated; a third term obtained by multiplying the resistance value (R0) and the δ-axis current (iδ); a δ-axis current command for the δ-axis current; and the δ-axis current And the sum of the product of the current deviation (iδ * −iδ) and the first proportional gain (Kδ) (R0 · iδ + Kδ · (iδ * −iδ)), and the δ-axis voltage of the AC voltage ( δ-axis voltage command generation unit (32) that generates a δ-axis voltage command (vδ *) for vδ), and based on the γ-axis voltage command and the δ-axis voltage command, the AC voltage of the drive device A control signal generation unit (30) for generating a control signal for controlling application, and adopting a value that makes the sum of the resistance value and the constant (kγγ) substantially zero .

A second aspect of the motor drive device control device according to the present invention is the motor drive device control device according to the first aspect, wherein the γ-axis voltage command generator (31) includes the first and , The sum ((R0 + kγγ) · iγ + L0 · ω1 · Iδ * +) of the second term and the integral term based on the first integral gain (K (ω1)) with respect to the current deviation (iδ * −iδ) K (ω1) · 1 / s · (iδ * −iδ)) is generated to generate the γ-axis voltage command, and the polarity of the first integral gain is determined by the advance direction and the rotational direction of the synchronous motor. Are positive when they are the same, and negative when the advance direction and the rotation direction are opposite to each other.

A third aspect of the motor drive device control device according to the present invention is the motor drive device control device according to the first or second aspect, wherein the γ-axis voltage command generator (31) A value (Iδ *) obtained by dividing the magnetic flux command (Λδ *) by the armature winding inductance (L0) is obtained by dividing the interlinkage magnetic flux (Λ0) to the armature by the field by the inductance. The γ-axis voltage command (vγ *) is generated using a relationship (Iδ * = I0 + iδ *) that is equal to the sum of the value (I0) and the δ-axis current command (iδ *).

A fourth aspect of the motor drive device control device according to the present invention is the motor drive device control device according to any one of the first to third aspects, wherein the absolute value of the angular velocity (ω1) is predetermined. The command (Λδ *) of the primary magnetic flux when the value is smaller than the value is set larger than the value when the absolute value of the angular velocity is equal to a predetermined value.

A first aspect of an electric motor drive system according to the present invention includes an electric motor drive device control device (3) according to any one of the first to fourth aspects, the drive device (1), and the synchronous electric motor ( 2).

  A second aspect of the motor drive system according to the present invention is the motor drive system according to the first aspect, wherein the field (23) includes a core and a permanent magnet attached to the surface of the core. Prepare.

According to the first aspect of the control device for the electric motor drive device of the present invention, it contributes to the stability of the control. In addition, the calculation can be simplified.

According to the second aspect of the control device for the electric motor drive device of the present invention, it is possible to reduce the steady deviation (current deviation generated in the steady state) for the δ-axis current regardless of the rotation direction of the synchronous motor. .

According to the third aspect of the control device for the electric motor drive device of the present invention, the γ-axis voltage command can be generated using a relatively simple expression.

According to the fourth aspect of the control device for the motor drive device of the present invention, the torque output from the synchronous motor is larger as the δ-axis component of the primary magnetic flux is larger, and the dq-axis rotation coordinate and the γ-δ axis are increased. The closer to 90 degrees, the greater the phase difference angle with the rotating coordinate. The dq axis rotational coordinates here are formed by the d axis and q axis orthogonal to each other in electrical angle, and the d axis is an axis in phase with the magnetic pole center of the field.

  When the absolute value of the angular velocity is small, a command for the δ-axis component of the primary magnetic flux is set large. Thereby, if the load torque does not fluctuate, the phase difference angle moves away from 90 degrees. Since the phase difference angle is once moved away from 90 degrees, the phase difference angle can approach 90 degrees even if the load torque increases thereafter. Therefore, the synchronous motor can output torque according to the increased load torque.

  According to the first and second aspects of the electric motor drive system according to the present invention, it is possible to provide an electric motor drive system that contributes to stable control.

It is a figure which shows an example of a notional structure of a power converter device. It is a figure which shows an example of a rotation coordinate and a magnetic flux typically. It is a figure which shows notionally an example of an internal structure of a (delta) axis voltage command generation part.

First embodiment.
<1. Configuration of power conversion device>
Referring to FIG. 1, the electric motor drive system includes a drive device 1, a synchronous motor 2, and a control unit 3. The driving device 1 applies an alternating voltage to the synchronous motor 2 and outputs an alternating current. Thereby, the synchronous motor 2 is driven. The synchronous motor 2 includes an armature 21 and a field 23. The armature 21 has an armature winding 22. In the illustration of FIG. 1, one end of each of the three armature windings 22 is connected to the driving device 1 and the other ends are connected to each other. Such a connection is called a so-called star connection. When a three-phase AC voltage is applied to these armature windings 22, the armature windings 22 apply a rotating magnetic field to the field magnets 23. The field magnet 23 supplies a field magnetic flux to the armature 21 and rotates with respect to the armature 21 in synchronization with the rotating magnetic field. For example, the field magnet 23 includes a core and a permanent magnet that is provided on the outer peripheral surface of the core and supplies a field magnetic flux. Such a synchronous motor is also called a surface magnet synchronous motor.

  In the illustration of FIG. 1, the driving device 1 is an inverter, and inputs a DC voltage Vdc between the positive DC power supply line LH and the negative DC power supply line LL, and converts the DC voltage Vdc into an AC voltage. For example, the drive device 1 includes switching elements Sup, Svp, Swp, Sun, Svn, Swn and diodes Dup, Dvp, Dwp, Dun, Dvn, Dwn. Switching elements Sxp, Sxn (x represents u, v, w, and so on) are connected in series between DC power supply lines LH, LL. The diodes Dxp and Dxn are connected in parallel with the switching elements Sxp and Sxn, respectively. The forward direction of the diodes Dxp and Dxn is a direction from the DC power supply line LL toward the DC power supply line LH. A connection point Px connecting the switching elements Sxp and Sxn is connected to the synchronous motor 2 (armature winding 22).

  The driving device 1 receives a control signal from the control unit 3. Such a control signal is a signal for controlling application of an alternating voltage by the driving device 1. In other words, the driving device 1 is controlled by the control unit 3. Referring to FIG. 1, the control unit 3 gives a control signal (switching signal) to the switching elements Sxp and Sxn. The switching elements Sxp and Sxn are turned on / off based on the switching signal. By applying appropriate switching signals to the switching elements Sxp and Sxn, the driving device 1 can convert the DC voltage Vdc into an AC voltage and apply it to the synchronous motor 2.

  The control unit 3 includes a control signal generation unit 30, a γ-axis voltage command generation unit 31, a δ-axis voltage command generation unit 32, and a coordinate conversion unit 33. Each configuration of the control unit 3 will be described in detail later.

  Here, the control unit 3 includes a microcomputer and a storage device. The microcomputer executes each processing step (in other words, a procedure) described in the program. The storage device is, for example, a ROM (Read-Only-Memory), a RAM (Random-Access-Memory), a rewritable nonvolatile memory (EPROM (Erasable-Programmable-ROM), etc.), and various storage devices such as a hard disk device. One or more can be configured. The storage device stores various information, data, and the like, stores a program executed by the microcomputer, and provides a work area for executing the program. It can be understood that the microcomputer functions as various means corresponding to each processing step described in the program, or can realize that various functions corresponding to each processing step are realized. The control unit 3 is not limited to this, and various procedures executed by the control unit 3 or various means or various functions implemented may be realized by hardware.

<2. Control Method of Drive Device>
<2-1. Overview of Control Method of Non-Patent Document 1>
FIG. 2 is a diagram schematically showing the field 23 and the rotation coordinates. In FIG. 2, the field 23 viewed electrically is schematically illustrated, and is schematically illustrated by an N-pole magnetic pole 231 and an S-pole magnetic pole 232. In FIG. 2, an axis in phase with the magnetic pole center of the field magnet 23 is shown as a d-axis, and an axis orthogonal to the d-axis by an electrical angle is shown as a q-axis. The d axis and the q axis form a dq axis rotation coordinate, and the dq axis rotation coordinate rotates in accordance with the rotation of the synchronous motor 2.

  FIG. 2 also shows γ-δ axis rotation coordinates having a phase (hereinafter referred to as a phase difference angle) φ with respect to the dq axis rotation coordinates. The γ-δ axis rotation coordinates are formed by the γ axis and the δ axis. The γ-axis advances 90 degrees with respect to the δ-axis in a predetermined advance direction (clockwise direction in FIG. 2). Further, the δ axis advances by the phase difference angle φ with respect to the d axis, and the γ axis advances by the phase difference angle φ with respect to the q axis.

  Here, the direction of rotation of the synchronous motor 2 (the direction of rotation of the field 23 with respect to the armature 21) is defined as the forward rotation direction when it coincides with a predetermined advance direction.

  In such γ-δ axis rotation coordinates, the number of primary linkage fluxes (hereinafter referred to as primary flux) of the synchronous motor 2 is considered. The primary magnetic flux here is a combination of the interlinkage magnetic flux Λ 0 to the armature 21 by the field 23 and the magnetic flux of the armature reaction generated by the alternating current flowing through the armature 21.

  As illustrated in FIG. 2, since the d-axis is an axis in phase with the magnetic pole center of the field 23, the linkage flux Λ0 to the armature 21 by the field 23 is along the d-axis. Therefore, the γ-axis component λ1γ and the δ-axis component λ1δ of the primary magnetic flux are the γ-axis current iγ that is the γ-axis component of the alternating current flowing through the armature winding 22, the δ-axis current iδ that is the δ-axis component, and the armature winding Using the inductance L of the line 22, it is expressed by the following equation.

λ1γ = L · iγ + λ2γ (1)
λ2γ = −Λ0 · sinφ (2)
λ1δ = L · iδ + λ2δ (3)
λ2δ = Λ0 · cosφ (4)

  Here, λ2γ and λ2δ are a γ-axis component and a δ-axis component of the linkage flux Λ0, respectively. When the synchronous motor 2 has a saliency like a permanent magnet embedded synchronous motor, a known relational expression with the d-axis inductance Ld, the q-axis inductance Lq, and the phase difference angle φ as variables is used as the inductance L. Thus, the present embodiment can be applied to a motor having saliency, such as a permanent magnet embedded synchronous motor. In simple terms, it can be approximated by an average value of the d-axis inductance Ld and the q-axis inductance Lq.

  The output torque τ is calculated based on the outer product of the primary magnetic flux and the current. More specifically, the output torque τ is expressed by the following equation.

  τ = np · (λ1δ · iγ−λ1γ · iδ) (5)

  Here, np indicates the number of pole pairs of the field 23.

  According to the equation (5), when the primary magnetic flux (λ1γ, λ1δ) is controlled by the following equation, the synchronous motor 2 generates an output torque τ proportional to the γ-axis current iγ.

λ1γ = 0 (6)
λ1δ = Λδ * (7)

  Here, Λδ * is a command value and is set as appropriate. When the primary magnetic flux is controlled by the equations (6) and (7), the δ axis becomes an axis in phase with the primary magnetic flux.

  Next, in order to control the drive device 1 so that the primary magnetic flux satisfies the expressions (6) and (7), the command value of the AC voltage output from the drive device 1 will be considered. First, the voltage equation of the synchronous motor 2 is expressed by the following equation.

vγ = R · iγ + L · P · Iγ ′ + L · ω1 · Iδ ′ (8)
vδ = R · iδ + L · P · Iδ'-L · ω1 · Iγ '(9)

  Here, vγ and vδ are the γ-axis component and δ-axis component of the AC voltage output from the driving device 1, respectively, and are referred to as γ-axis voltage and δ-axis voltage, respectively. R indicates the resistance value of the resistance component of the armature winding 22, and P indicates a differential operator. ω1 indicates the angular velocity of the AC voltage output from the driving device 1. Iγ ′ and Iδ ′ are values obtained by dividing the γ-axis component λ1γ and δ-axis component λ1δ of the primary magnetic flux by the inductance L, respectively.

  It can be approximated that the γ-axis component λ1γ and the δ-axis component λ1δ of the primary magnetic flux are constant in the steady state. Therefore, in the steady state, the second terms on the right side in equations (8) and (9) are both zero. Further, from the equation (6), Iγ ′ = 0 is substituted into the equations (8) and (9), and from the equation (7), Iδ ′ = Iδ * (= Λδ * / L) is substituted into the equations (8) and (9). By substituting, the γ-axis voltage vγ and the δ-axis voltage vδ when the expressions (6) and (7) are satisfied can be obtained by the following expressions.

vγ = R · iγ + L · ω1 · Iδ * (10)
vδ = R · iδ (11)

  Although the γ-axis voltage vγ and δ-axis voltage vδ calculated in this way can be adopted as the γ-axis voltage command vγ * and the δ-axis voltage command vδ *, such control is so-called feedforward control. Further, in Non-Patent Document 1, the current is feedback controlled. Therefore, the current command value will be considered. From the equations (1), (2) and the equations (3), (4), the following equations are derived respectively.

Iγ ′ = iγ−I0 · sinφ (12)
Iδ ′ = iδ + I0 · cosφ (13)

  Here, I0 is a value obtained by dividing the flux linkage Λ0 by the inductance L. By substituting Iγ ′ = 0 from equation (6) into equation (12) and substituting Iδ ′ = Iδ * from equation (7) into equation (13), a γ-axis current command iγ * for the γ-axis current iγ is obtained. And the δ-axis current command i δ * for the δ-axis current i δ is derived by the following equation.

iγ * = I0 · sinφ (14)
iδ * = Iδ * −I0 · cosφ (15)

  That is, the γ-axis current iγ and the δ-axis current iδ when the expressions (6) and (7) are satisfied are expressed by the expressions (14) and (15), respectively.

  However, here, feedback control is performed for the δ-axis current iδ, and feedback control is not performed for the γ-axis current iγ. More specifically, the γ-axis voltage command vγ * and the δ-axis voltage command vδ * are calculated by the following equations.

vγ * = R · iγ + L · ω1 · Iδ * + Kγ · (iδ * −iδ) / ω1 (16)
vδ * = R · iδ + Kδ · (iδ * −iδ) (17)

  In Equation (16), the reciprocal of the angular velocity ω1 is provided. Thereby, as described in Non-Patent Document 1, the angular velocity ω1 can be canceled in the constant term of the characteristic equation of the magnetic flux control system, and the stability can be improved.

  On the other hand, when the angular velocity ω1 is zero, the reciprocal of the angular velocity ω1 cannot be calculated.

  1 / ω1 = ω1 / {(ω1 *) ^ 2+ (0.05ωr) ^ 2} (18)

  Here, ω1 * indicates a command for the angular velocity ω1, and ωr indicates a rated angular velocity. A ^ B means a power with A and B as bases and exponents, respectively.

  As described above, the γ-axis voltage command vγ * and the δ-axis voltage command vδ * are calculated using the equations (15) to (18). Then, a known coordinate transformation is applied to these to generate three-phase phase voltage commands Vu *, Vv *, Vw *, and a driving device based on the phase voltage commands Vu *, Vv *, Vw * by a known method. 1 is output as a switching signal. Thereby, the drive device 1 ideally outputs a phase voltage equal to the phase voltage commands Vu *, Vv *, and Vw *.

  However, the resistance value R and the inductance L that are employed in the calculation of the γ-axis voltage command vγ * and the δ-axis voltage command vδ * are not actually measured values but, for example, nominal values. Here, the inductance L of the armature winding 22 and the resistance value R of its resistance component are represented by the following equations, respectively.

R = R0−ΔR (19)
L = L0−ΔL (20)

  R0 and L0 are resistance values (for example, nominal values) and inductances (for example, nominal values) used for control, respectively, and ΔR and ΔL correspond to variations due to manufacturing variations and temperature changes. When the equations (19) and (20) are substituted into the equations (8) and (9), the voltage equation of the synchronous motor 2 is expressed by the following equation.

vγ = (R0−ΔR) · iγ + (L0−ΔL) · P · Iγ ′ + (L0−ΔL) · ω1 · Iδ ′ (21)
vδ = (R0−ΔR) · iδ + (L0−ΔL) · P · Iδ ′ − (L0−ΔL) · ω1 · Iγ ′ (22)

  On the other hand, in the equations (16) and (17), the resistance value R0 and the inductance L0 are adopted. Therefore, in Non-Patent Document 1, the δ-axis voltage command vδ * and the γ-axis voltage command vγ * are expressed by the following equations.

vγ * ＝ R0 ・ iγ + L0 ・ ω1 ・ Iδ * + Kγ ・ (iδ * −iδ) ・ ω1 / {(ω1 *) ^ 2+ (0.05ωr) ^ 2} (23)
vδ * = R0 · iδ + Kδ · (iδ * −iδ) (24)

  Then, as described in Non-Patent Document 1 (see also Patent Document 1), when the characteristic equation for the case where vγ = vγ * and vδ = vδ * is established, the following characteristic equation including manufacturing variations is obtained. Derived.

  When the angular velocity ω1 is zero, the coefficient α2 is −ΔR · Kδ / (L0) ^ 2, and may take a negative value depending on the manufacturing variation ΔR. At this time, the stability condition of Rouss-Fillwitz (α0> 0 and α1> 0 and α2> 0) is not satisfied. Therefore, the control becomes unstable.

<2-2. Control method of the present embodiment>
The γ-axis voltage command generation unit 31 and the δ-axis voltage command generation unit 32 generate a γ-axis voltage command vγ * and a δ-axis voltage command vδ * using the following equations, respectively.

vγ * = (R0 + kγγ) ・ iγ + L0 ・ ω1 ・ Iδ * (26)
vδ * = R0 · iδ + Kδ · (iδ * −iδ) (27)

  Here, kγγ is a predetermined value, and is hereinafter also referred to as a stability constant.

  The generation of such a command can also be expressed as follows. That is, the γ-axis voltage command generator 31 generates a term that is a product of the sum of the resistance value R0 of the resistance component of the armature winding 22 and the stability constant kγγ and the γ-axis current iγ, the angular velocity ω1, and the primary magnetic flux. The sum of the δ-axis component command Λδ * (= L0 · Iδ *) and the product term is calculated to generate the γ-axis voltage command vγ *. Further, the δ-axis voltage command generation unit 32 calculates the sum of a term obtained by multiplying the resistance value R0 and the δ-axis current iδ and a term obtained by multiplying the current deviation (iδ * −iδ) and the proportional gain Kδ, A voltage command vδ * is generated.

  In Expressions (26) and (27), the resistance value R0, the inductance L0, the stabilizing constant kγγ, and the proportional gain Kδ are preset values, and are recorded on a recording medium (not shown), for example.

  The δ-axis current command iδ * in the equation (27) is a command for the δ-axis current iδ and is expressed by the equation (15). Further, as in Non-Patent Document 1, cosφ may be approximated by (1−iγ ^ 2/2) to calculate the δ-axis current command iδ *. More specifically, the δ-axis current command iδ * may be calculated based on the following equation.

  iδ * = Iδ * −I0 + iγ ^ 2 · I0 / 2 (28)

  The value I0 (= Λ0 / L0) in the equation (28) is a preset value, and is recorded on the recording medium, for example. The command Iδ * is input to the control unit 3 from the outside, for example. It should be noted that the δ-axis current command iδ * is input to the control unit 3 from the outside, and the command Iδ * is calculated based on Iδ * = iδ * + I0−iγ ^ 2 · I0 / 2 (see Expression (28)). It doesn't matter.

  In equations (26) and (27), the δ-axis current iδ and the γ-axis current iγ are detected values. The δ-axis current iδ and the γ-axis current iγ are detected as follows. First, alternating currents (line currents) iu, iv, iw output to the synchronous motor 2 are detected. In the illustration of FIG. 1, the two-phase alternating currents iu, iv out of the three-phase alternating currents iu, iv, iw are detected by the current detection unit 4. Since the sum of the three-phase alternating currents iu, iv, iw is ideally zero, the remaining one-phase alternating current iw can be calculated from the two-phase alternating currents iu, iv. The DC current idc input to the driving device 1 may be detected, and the DC current idc may be detected as the AC current based on the switching pattern of the driving device 1 at the time of detection. Since such a detection method is well-known, detailed description is abbreviate | omitted.

  The detected alternating currents iu and iv are input to the control unit 3 (more specifically, the coordinate conversion unit 33). A known coordinate transformation is applied to the alternating currents iu and iv by the coordinate transformation unit 33 to calculate a δ-axis current iδ and a γ-axis current iγ. As is well known, the angle necessary for coordinate conversion can be obtained by integrating the angular velocity ω1.

  The angular velocity ω1 is calculated by a known method. For example, the angular velocity command ω1 *, which is a command for the angular velocity ω1, may be adopted as the angular velocity ω1, or the angular velocity ω1 may be obtained by the following equation in the same manner as in Non-Patent Document 1, for example.

  Here, Km and Tm are constants set in advance.

  In the illustration of FIG. 1, a rotational speed command ωm * for the rotational speed ωm of the synchronous motor 2 is input to the control unit 3. Since the angular velocity ω1 is represented by the product of the rotational speed ωm and the pole pair number np, the angular velocity command ω1 * is calculated by the product of the rotational speed command ωm * and the pole pair number np. The rotational speed ωm is positive in the forward direction and negative in the reverse direction.

  As described above, all the parameters on the right side of Expressions (26) and (27) are acquired. Therefore, the γ-axis voltage command vγ * and the δ-axis voltage command vδ * can be calculated by equations (26) and (27).

  The coefficients α0, α1, α2 of the characteristic equations in the equations (26), (27) are expressed by the following equations, respectively.

  Therefore, the following equation should be satisfied within the range of manufacturing variation ΔR so that both the coefficients α1 and α2 always take positive values when the angular velocity ω1 is zero.

Kδ−kγγ> 2 · ΔR · L0 (31)
−ΔR · (Kδ−kγγ) −Kδ · kγγ> 0 (32)

  When the manufacturing variation ΔR takes the maximum value ΔRmax, the right side of the equation (31) becomes the maximum, and if the proportional gain Kδ and the stability constant kγγ are set so as to satisfy the equation (31) at this time, the manufacturing variation. Expression (31) always holds regardless of ΔR. Since (Kδ−kγγ) is positive at this time, the first term on the left side of Equation (32) is minimum when the manufacturing variation ΔR takes the maximum value ΔRmax. Accordingly, when the production variation ΔR takes the maximum value ΔRmax, if the proportional gain Kδ and the stability constant kγγ are set so as to satisfy the equation (32), the equation (32) is always established regardless of the production variation ΔR. Therefore, the following expression should be satisfied.

Kδ−kγγ> 2 · ΔRmax · L0 (33)
Kδ · kγγ <−ΔRmax · (Kδ−kγγ) (34)

  Using equation (33), equation (34) can be modified as follows.

  Kδ · kγγ <−2 · ΔRmax ^ 2 · L0 (35)

  As described above, if the proportional gain Kδ and the stability constant kγγ are set so as to satisfy both the expressions (33) and (35), the stability condition of Rous-Fillbits can be satisfied regardless of the manufacturing variation ΔR. . Therefore, the control can be stabilized. Therefore, control based on the γ-axis voltage command vγ * and the δ-axis voltage command vδ * generated using the equations (26) and (27) contributes to the improvement of the stability.

  In Equation (26), a value that makes the sum of the resistance value R0 and the stability constant kγγ substantially zero (for example, the product of the resistance value R0 and −1) may be employed as the stability constant kγγ. In this case, the γ-axis voltage command vγ * is expressed by the following equation.

  vγ * = L0 · ω1 · Iδ * (36)

  According to the γ-axis voltage command vγ * in Expression (36), it is possible to reduce the arithmetic processing as compared with the case where Expression (26) is adopted.

  The coefficients α0, α1, α2 of the characteristic equations in the equations (36), (27) are expressed by the following equations, respectively.

  According to Expression (37), the proportional gain Kδ can be set so that both the coefficients α1 and α2 always take positive values when the angular velocity ω1 is zero. That is, a proportional gain Kδ that satisfies both of the following expressions may be employed.

Kδ> 2ΔRmax · L0−R0 (38)
Kδ> ΔRmax · R0 / (R0−ΔRmax) (39)

  As described above, the control based on the γ-axis voltage command vγ * and the δ-axis voltage command vδ * generated using the equations (36) and (27) contributes to the improvement of the stability.

  For the γ-axis voltage command vγ *, a proportional process based on the proportional gain Kγ may be performed on the current deviation (iδ * −iδ). More specifically, the γ-axis voltage command vγ * may be calculated using the following equation.

  vγ * = (R0 + kγγ) ・ iγ + L0 ・ ω1 ・ Iδ * + Kγ / ω1 ・ (iδ * −iδ) (40)

  In addition, in equation (40), the reciprocal of the angular velocity ω1 is not necessarily required. However, providing the reciprocal of the angular velocity ω1 contributes to stable control. Moreover, Formula (18) can be employ | adopted as a reciprocal number of angular velocity (omega) 1.

  The generation of a command based on the formula (40) can be expressed as follows. That is, the γ-axis voltage command generator 31 generates a term that is a product of the sum of the resistance value R0 of the resistance component of the armature winding 22 and the stability constant kγγ and the γ-axis current iγ, the angular velocity ω1, and the primary magnetic flux. A sum of a product term of the δ-axis component command Λδ * (= L0 · Iδ *) and a proportional term based on the current deviation (iδ * −iδ) and the proportional gain Kγ is calculated to obtain a γ-axis voltage command vγ. * Is generated.

  If equation (40) is adopted, proportional processing {Kγ / ω1 · (iδ * −iδ)} is added. Therefore, the current deviation (iδ * −iδ) can be quickly reduced by this proportional processing.

  Further, the coefficients α0, α1, and α2 of the characteristic equations in the equations (40) and (27) are respectively expressed by the following equations. Here, Expression (18) is used.

  Since the coefficients α1 and α2 when the angular velocity ω1 is zero are the same as those in the equation (30), the proportional gain Kδ and the stability constant kγγ are set so that both the coefficients α1 and α2 always take positive values. can do.

  In the formula (40), -R0 may be adopted as the stability constant kγγ. This can simplify the arithmetic processing.

  The coefficients α0, α1, α2 of the characteristic equation at this time are expressed by the following equations.

  Since the coefficients α1 and α2 when the angular velocity ω1 is zero are the same as in the equation (37), the proportional gain Kδ can be set so that both the coefficients α1 and α2 always take positive values.

Second embodiment.
The conceptual configuration of the electric motor drive system according to the second embodiment is the same as that shown in FIG. However, in the second embodiment, when the absolute value of the angular velocity ω1 is lower than the predetermined value ωref2, the command Λδ * (or the command Iδ *) for the δ-axis component of the primary magnetic flux is set as the absolute value of the angular velocity ω1. It is set larger than the command Λδ * (Iδ *) when it is equal to the value ωref2. The significance is described below. Here, the angular velocity ω1 takes a positive or negative value according to the rotation direction of the synchronous motor 2, for example, takes a positive value in the forward rotation direction (clockwise in FIG. 2), and takes a negative value in the reverse rotation direction.

  As described in the first embodiment, if the primary magnetic flux (λ1γ, λ1δ) satisfies the equations (6) and (7), the output torque τ can be made proportional to the γ-axis current iγ. At this time, the output torque τ is expressed by the following equation (see also equation (5)).

  τ = np · Λδ * · iγ (43)

  Further, when Expression (6) is satisfied, since Iγ ′ = λ1γ / L = 0 in Expression (12), the following expression is derived.

  τ = np · Λδ * · I0 · sinφ (44)

  Here, the number n of pole pairs and the value I0 (= Λ0 / L) are constants determined by the configuration of the synchronous motor 2, and the command Λδ * is a set value. The phase difference angle φ is a variable, and when the phase difference angle φ takes 90 degrees, the output torque τ takes the maximum value.

  Now, when the absolute value of the angular velocity ω1 is lower than the predetermined value ωref2 or at the time of start-up, the load torque is likely to fluctuate compared to the steady state. Therefore, when the absolute value of the angular velocity ω1 is lower than the predetermined value ωref2, the command Λδ * is increased to increase the range (maximum value) that the output torque τ can take at low speed or at startup. Thereby, even if the load torque increases, the output torque τ corresponding to the load torque can be output. As understood from the equation (44), by increasing the command Λδ *, the influence of the fluctuation of the output torque τ on the fluctuation of the phase difference angle φ is reduced. That is, even if the output torque τ increases with the increase of the load torque at low speed or at the time of start-up, the fluctuation of the phase difference angle φ is suppressed.

  In addition, when the absolute value of the angular velocity ω1 is higher than the predetermined value ωref2, that is, when the load torque is relatively small, if a smaller command Λδ * is employed, the current flowing through the armature winding 22 (for example, the δ-axis current iδ) ) Can be reduced, so that power consumption can be reduced.

  Such a command Λδ * (or Iδ *) can be set as follows, for example. That is, the command Λδ ** is generated by correcting the command Λδ ** before correction for the δ-axis component of the primary magnetic flux according to the absolute value of the angular velocity ω1. More specifically, when the absolute value of the angular velocity ω1 is lower than the predetermined value ωref2, the command Λδ ** is increased to generate the command Λδ *, and the command is issued when the absolute value of the angular velocity ω1 is higher than the predetermined value ωref2. Λδ ** is adopted as the command Λδ * as it is.

Third embodiment.
The γ-axis voltage command vγ * may be calculated by the following formula.

  vγ * = (R0 + kγγ) ・ iγ + L0 ・ ω1 ・ Iδ * + K (ω1) ・ 1 / s ・ (iδ * −iδ) (45)

  K (ω1) is an integral gain, and s is a Laplace operator. The third term on the right side of Equation (45) is an integral term obtained by performing an integration process based on the integral gain K (ω1) for the current deviation (iδ * −iδ).

  The integral gain K (ω1) is positive or negative depending on the rotation direction of the synchronous motor 2. More specifically, the integral gain K (ω1) is positive when the rotation direction is the forward rotation direction, and the integral gain K (ω1) is negative when the rotation direction is the reverse rotation direction. Here, the forward rotation direction is the advance direction of the γ axis with respect to the δ axis in FIG. 2, and is the clockwise direction in FIG. With such an integral gain K (ω1), it is possible to appropriately reduce a steady deviation (steady current deviation) with respect to the δ-axis current according to the rotation direction of the synchronous motor 2. Hereinafter, the steady deviation will be described in detail and the reduction will be described.

  Assuming that the δ-axis current iδ and the δ-axis current command iδ * are equal to each other, Iδ * = iδ + I0 · cosφ is established from the equation (15). By substituting this into the equation (10), the γ-axis voltage command vγ * can be expressed by the following equation.

  vγ * = R · iγ + ω1 · (L · iδ + Λ0 · cosφ) (46)

  If the angular velocity ω1 is negative when the rotation direction of the synchronous motor 2 is the reverse rotation direction, the equation (46) is established even when the rotation direction is the reverse rotation direction.

  According to Expression (46), when the rotation direction of the synchronous motor 2 is the forward rotation direction, the higher the δ-axis current iδ, the higher the γ-axis voltage command vγ *. Conversely, an increase in the γ-axis voltage command vγ * contributes to an increase in the δ-axis current iδ. On the other hand, when the rotation direction of the synchronous motor 2 is the reverse rotation direction, the angular velocity ω1 is negative. Therefore, the higher the δ-axis current iδ, the smaller the γ-axis voltage command vγ *. Conversely, an increase in the γ-axis voltage command vγ * contributes to a decrease in the δ-axis current iδ. In particular, in the region where the absolute value of the angular velocity ω1 is high, the first term of the equation (46) can be ignored, and thus the correlation between the γ-axis voltage command vγ * and the δ-axis current is strengthened.

  In the present embodiment, the integral gain K (ω1) is positive when the rotation direction is the forward rotation direction and is negative when the rotation direction is the reverse rotation direction. Therefore, if the rotation direction is the forward rotation direction, the third term of Expression (45) becomes positive when the δ-axis current is smaller than the δ-axis current command iδ *. Therefore, at this time, the γ-axis voltage command vγ * increases. Since the increase in the γ-axis voltage command vγ * contributes to the increase in the δ-axis current iδ in the forward rotation direction, the steady deviation with respect to the δ-axis current iδ can be reduced. The third term of the formula (45) is negative when the δ-axis current is larger than the δ-axis current command iδ *. Therefore, at this time, the γ-axis voltage command vγ * is reduced. Since the reduction of the γ-axis voltage command vγ * contributes to the reduction of the δ-axis current command iδ * in the forward rotation direction, the steady deviation for the δ-axis current iδ can be reduced.

  Similarly, even when the rotational direction is the reverse direction, the steady deviation with respect to the δ-axis current iδ can be reduced. Therefore, according to the γ-axis voltage command vγ * in Expression (45), the steady-state deviation for the δ-axis current iδ can be appropriately reduced according to the rotation direction.

  In Non-Patent Document 1, the resistance value R and the value I0 are automatically corrected using these estimated values in order to reduce the error about the resistance value R and the value I0 (= Λ0 / L). In the present embodiment, such automatic correction is not necessary. Further, fixed values set in advance as the resistance value R and the value I0 can be adopted.

  Next, the magnitude of the steady deviation due to manufacturing variation will be considered. In the equation (46), the γ-axis voltage command vγ * has a product (L · ω1) of the inductance L and the angular velocity ω1 as a term. Therefore, the steady-state deviation of the δ-axis current iδ caused by the manufacturing variation ΔL with respect to the inductance L is larger as the absolute value of the angular velocity ω1 is larger.

  Therefore, the absolute value of the integral gain K (ω1) may be set in advance so as to increase as the absolute value of the angular velocity ω1 increases. As a result, the larger the absolute value of the angular velocity ω1, the larger the variation amount of the γ-axis voltage command vγ * by the integration process, and thus the steady-state deviation can be appropriately reduced.

  For example, the integral gain K (ω1) is set by the product (Kγ ′ · ω1) of the angular velocity ω1 and the positive constant gain Kγ ′. The positive and negative polarities of the gain Kγ ′ are the same regardless of the rotation direction, and are positive, for example. According to this, the integral gain K (ω1) can be easily calculated. If the angular velocity ω1 is positive in the forward rotation direction and negative in the reverse rotation direction, the sign of the integral gain K (ω1) also satisfies the above description.

  In order to further reduce the steady-state deviation of the δ-axis current iδ, the δ-axis voltage command generation unit 32 performs an integration process based on the integral gain Kδ ′ on the current deviation (iδ * −iδ) to obtain a δ-axis voltage command vδ. * May be generated. However, the integral gain Kδ ′ does not need to depend on the angular velocity ω1. This is because Iγ ′ = 0 if the primary magnetic flux control is performed, and the δ-axis voltage vδ does not depend on the angular velocity ω1 according to the equation (11). Therefore, unlike the integration process for the γ-axis voltage command vγ *, the integration process for the δ-axis voltage command vδ * employs a constant integral gain Kδ ′ that does not depend on the angular velocity ω1.

  Therefore, the δ-axis voltage command generation unit 32 generates the δ-axis voltage command vδ * using the following equation.

  vδ * = R0 · iδ + Kδ · (iδ * −iδ) + Kδ '· 1 / s · (iδ * -iδ) (47)

  As described above, the steady deviation of the δ-axis current iδ can be further reduced by performing the integration process on the δ-axis voltage command vδ *.

  Further, the δ-axis voltage command generation unit 32 may perform integration processing on the δ-axis voltage command vδ * only when the absolute value of the angular velocity ω1 is lower than a predetermined value. In other words, when the absolute value of the angular velocity ω1 is lower than the predetermined value ωref, the δ-axis voltage command generation unit 32 performs integration processing to generate the δ-axis voltage command vδ *, and the absolute value of the angular velocity ω1 is the predetermined value ωref. If higher, the δ-axis voltage command vδ ** may be used as it is as the δ-axis voltage command vδ *.

  Such a process can be realized, for example, by adopting an integral gain Kδ ′ that becomes zero when the absolute value of the angular velocity ω1 is higher than a predetermined value ωref.

  Alternatively, as illustrated in FIG. 3, the δ-axis voltage command generation unit 32 may include a pre-correction command generation unit 321, a switch 322, a comparison unit 323, and an integration processing unit 324. The pre-correction command generation unit 321 generates a δ-axis voltage command vδ **. The comparison unit 323 compares the absolute value of the angular velocity ω <b> 1 and the predetermined value ωref <b> 1 and outputs the comparison result to the switch 322. The δ-axis voltage command vδ ** is input to the input terminal 322a of the switch 322. The switch 322 connects the input terminal 322a and the output terminal 322b when the absolute value of the angular velocity ω1 is larger than the predetermined value ωref1, and the input terminal 322a and the output terminal when the absolute value of the angular velocity ω1 is smaller than the predetermined value ωref1. 322c is connected. The integration processing unit 324 receives the output terminal 322b, performs integration processing, and outputs a δ-axis voltage command vδ *. The output terminal 322b is connected to the output terminal of the integration processing unit 324.

  Such generation of the δ-axis voltage command vδ * can suppress an unnecessary increase in the δ-axis voltage vδ that occurs when the absolute value of the angular velocity ω1 is high. This will be described in detail below.

  When the absolute value of the angular velocity ω1 greatly decreases in a situation where the absolute value of the angular velocity ω1 is higher than the predetermined value ωref, the current deviation (iδ * −iδ) continues to take a relatively large value, and this is integrated to integrate the integral term {Kδ '· 1 / s · (iδ * −iδ)} increases. Such an integral term unnecessarily increases the δ-axis voltage command vδ *.

  Therefore, only when the absolute value of the angular velocity ω1 is lower than the predetermined value, integral control is performed to generate the δ-axis voltage command vδ *, thereby avoiding such an increase in the δ-axis voltage vδ.

  Even in such a case, the absolute value of the integral gain K (ω1) with respect to the γ-axis voltage command vγ * takes a smaller value as the absolute value of the angular velocity ω1 is lower, so that the γ-axis voltage vγ increases. Hateful.

Fourth embodiment.
The conceptual configuration of the electric motor drive system according to the fourth embodiment is the same as that shown in FIG. Here, the calculation process is simplified to generate the γ-axis voltage command vγ *. This will be described in detail below.

  The δ-axis current command i δ * and the command I δ * satisfy the relationship represented by Expression (15). Here, it is considered to approximate the phase difference angle φ to zero. This approximation is particularly appropriate when the load torque is small, for example. On the other hand, an error may occur due to this approximation, such as when the load torque is large. When this error is a problem, the integration control described above is preferably performed. This is because this error is reduced by the integration control.

  Substituting φ = 0 in equation (15), the following equation is derived.

  Iδ * = I0 + iδ * (48)

  The γ-axis voltage command generation unit 31 generates a γ-axis voltage command vγ * using the equation (48). In other words, the γ-axis voltage command generation unit 31 determines that the command Iδ * obtained by dividing the command Λδ * of the δ-axis component of the primary magnetic flux by the inductance L determines the flux linkage number Λ0 to the armature 21 by the field 23 as the inductance L The γ-axis voltage command vγ * is generated using a relationship that is equal to the sum of the value I0 divided by δ and the δ-axis current command iδ *.

  Further, as can be understood by multiplying the right side and the left side of the equation (48) by the inductance L, the voltage command generation unit 31 determines that the command Λδ * (= (L · Iδ *) of the δ-axis component of the primary magnetic flux is the field 23 Using the relationship that the number of flux linkages Λ0 (= L · I0) to the armature 21 due to is equal to the sum of the magnetic flux (= L · iδ *) of the armature reaction generated by the δ-axis current command iδ *. It can also be explained that the γ-axis voltage command vγ * is generated.

  More specifically, for example, a δ-axis current command iδ * is input from the outside to the γ-axis voltage command generation unit 31, and a command Iδ * is generated based on Expression (43). Then, using the command Iδ * and the δ-axis current command iδ *, a γ-axis voltage command vγ * is generated in the same manner as in the first embodiment.

  Alternatively, for example, a command Iδ * is input from the outside, and a δ-axis current command iδ * is generated based on the equation (43), and a γ-axis voltage command is used in the same manner as in the first or second embodiment. vγ * may be generated.

  According to the third embodiment, the arithmetic processing can be simplified as compared with the case where Expression (28) is adopted.

DESCRIPTION OF SYMBOLS 1 Drive device 2 Synchronous motor 21 Armature 22 Armature winding 23 Field vδ *, vγ * Voltage command iδ * Current command

Claims (6)

Driving device for applying an AC voltage to a synchronous motor (2) having an armature (21) including an armature winding (22) and a field (23) and outputting an AC current (iu, iv, iw) A device (3) for controlling (1),
The alternating current has a δ-axis and a γ-axis that advances 90 degrees in a predetermined advance direction with respect to the δ-axis, and the δ-axis is related to a flux linkage (Λ0) to the armature by the field. A coordinate conversion unit (33) for converting into a δ-axis current (iδ) and a γ-axis current (iγ) in a rotating coordinate system having a phase (φ)
A first term obtained by multiplying the sum of a resistance value (R0) of a resistance component of the armature winding and a preset constant (kγγ) and the γ-axis current, an angular velocity (ω1) of the AC voltage, and the chain The sum ((R0 + kγγ) ·) of the product of the primary magnetic flux command (Λδ *), which is a combination of the alternating magnetic flux (Λ0) and the armature reaction magnetic flux generated by the alternating current flowing through the armature. γ-axis voltage command generator (31) for calculating γ-axis voltage command (vγ *) for the γ-axis voltage (vγ) of the AC voltage by calculating iγ + L0, ω1, Iδ *),
A third term obtained by multiplying the resistance value (R0) and the δ-axis current (iδ), a current deviation (iδ * −iδ) between the δ-axis current command for the δ-axis current and the δ-axis current, and the first The sum (R0 · iδ + Kδ · (iδ * -iδ)) with the product of the proportional gain (Kδ) is calculated, and the δ-axis voltage command (vδ *) for the δ-axis voltage (vδ) of the AC voltage is calculated. A δ-axis voltage command generator (32) to generate,
Based on the γ-axis voltage command and the δ-axis voltage command, a control signal generation unit (30) that generates a control signal for controlling the application of the AC voltage by the drive device ,
A control device for an electric motor drive device that employs a value at which a sum of the resistance value and the constant (kγγ) becomes substantially zero as the constant (kγγ) . The γ-axis voltage command generation unit (31) includes the first term, the second term, and an integral term based on a first integral gain (K (ω1)) with respect to the current deviation (iδ * −iδ). The sum ((R0 + kγγ) · iγ + L0 · ω1 · Iδ * + K (ω1) · 1 / s · (iδ * −iδ)) is calculated to generate the γ-axis voltage command, and the first polarity of the integral gain, the process proceeds to the direction and the rotation direction of the synchronous motor is positive and when the same is negative when the flow proceeds to the direction and the rotation direction are opposite to each other, according to claim 1 Control device for motor drive device. The γ-axis voltage command generation unit (31) is configured such that a value (Iδ *) obtained by dividing the command (Λδ *) of the primary magnetic flux by an inductance (L0) of the armature winding is the armature by the field. Using the relationship (Iδ * = I0 + iδ *) that is equal to the sum of the value (I0) obtained by dividing the interlinkage magnetic flux (Λ0) by the inductance and the δ-axis current command (iδ *), generating a γ-axis voltage command (v? *), the control device of the electric motor drive device according to claim 1 or 2. The command (Λδ *) of the primary magnetic flux when the absolute value of the angular velocity (ω1) is smaller than a predetermined value is set larger than a value when the absolute value of the angular velocity is equal to a predetermined value. 4. The control device for an electric motor drive device according to any one of items 1 to 3 . A control device (3) for an electric motor drive device according to any one of claims 1 to 4 ,
The driving device (1);
An electric motor drive system comprising the synchronous motor (2). The motor drive system according to claim 5 , wherein the field (23) includes a core and a permanent magnet attached to a surface of the core.

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