JP4781933B2 - Electric motor control device - Google Patents

Description

  The present invention relates to a motor control device that obtains a constant of a permanent magnet field type rotary motor and performs energization control of the rotary motor using the constant.

  In a permanent magnet field type rotary electric motor, the coil resistance of the armature changes depending on the temperature of the electric motor. Therefore, in order to perform good torque control, it is necessary to compensate for the change in the resistance of the armature when controlling the energization amount of the motor.

Therefore, conventionally, when the motor is stopped or when the motor is decelerated and stopped, a DC voltage is applied so that a constant DC current flows through the armature, and the coil of the armature is calculated from the ratio of the DC voltage and the DC current. There has been proposed a method of calculating resistance and performing energization control of an electric motor based on the calculated coil resistance (see, for example, Patent Document 1).
JP 2000-312497 A

  In addition to when the motor is stopped or decelerated and stopped, for example, even when the motor is operating at a high speed, the accuracy of energization control can be further improved by calculating the coil resistance of the armature of the motor. it can.

  Therefore, an object of the present invention is to provide a motor control device that can increase the accuracy of energization control by expanding the range of calculating the coil resistance of the armature during operation of the motor.

  The present invention has been made to achieve the above object, and is a motor control device that supplies multiphase AC power for generating a rotating magnetic field to the armature of each phase of a permanent magnet type rotary motor. A first switching element for switching connection and disconnection between the armature and the output section on the high potential side of the DC power supply, which is individually provided for the armature of each phase, and the DC power supply A second switching element for switching between connection and disconnection between the low-potential-side output section and the armature, and the first and second switching elements to turn on and off the polyphase AC power. The present invention relates to an improvement in an electric motor control device that includes an energization control unit that generates and controls energization of the electric motor.

And when the current detection means for detecting the current flowing through the armature of the electric motor and the electric current supply control of the electric motor by the electric current supply control means are being executed and the electric motor is rotating at least at a predetermined rotational speed or more the first of all the switching elements and the second switching element after setting the dead time to the OFF state, the first a switching element to a total in the oN state and the second all OFF the switching element A change in the detected current of the current detection means in the all-phase short-circuit state in which all the first switching elements are in the OFF state and all the second switching elements are in the ON state. Resistance calculation means for calculating the coil resistance of the armature of the electric motor based on the rate, and the energization control means comprises the resistance calculation means Based on the coil resistance more calculated, and performing energization control of the electric motor.

  According to the present invention, as will be described in detail later, even when the electric motor is rotating at a high speed, the resistance calculating means calculates the rate of change of the current flowing through the armature of the electric motor as the all-phase short-circuit state. By detecting, the coil resistance of the armature can be calculated. As a result, the operating range of the electric motor that can calculate the coil resistance of the armature is expanded, so that the accuracy of the energization control based on the coil resistance of the armature by the energization control means can be improved.

  Although the details will be described later, the resistance calculation means calculates the coil resistance of the armature of the electric motor based on the ratio of the change rate of the detected current of the current detection means at different time points in the all-phase short-circuit state. can do.

  And a coil temperature calculating means for calculating a coil temperature of the armature based on a ratio between a coil resistance of the armature at a preset reference temperature and a coil resistance calculated by the resistance calculating means, The energization control means performs energization control of the electric motor based on the coil temperature calculated by the coil temperature calculation means.

  According to the present invention, since the coil resistance of the armature of the electric motor changes in accordance with the coil temperature of the armature, the coil temperature calculation means is calculated by the coil resistance at the reference temperature and the resistance calculation means. The coil temperature of the armature at the time of calculating the coil resistance can be calculated based on the ratio to the coil resistance that has been made. And the said electricity supply control means can improve the precision of the said electricity supply control by performing the said electricity supply control based on the coil temperature of an armature.

  Further, the angular velocity detecting means for detecting the angular velocity of the electric motor, the angular velocity of the electric motor detected by the angular velocity detecting means, the detected current of the current detecting means, and the coil resistance calculated by the resistance calculating means. And an energized voltage constant calculating means for calculating an induced voltage constant of the electric motor, wherein the energization control means performs energization control of the electric motor based on the induced voltage constant calculated by the induced voltage constant calculating means. It is characterized by that.

  According to the present invention, although the details will be described later, the induced voltage constant calculating means calculates the induced voltage constant of the motor based on the rotation speed of the motor, the current flowing through the armature, and the resistance of the armature. can do. Since the characteristics of the electric motor change depending on the magnitude of the induced voltage constant, the energization control means improves the accuracy of the energization control by performing the energization control based on the induced voltage constant of the motor. Can be made.

  Further, the magnet temperature for calculating the field magnet temperature of the electric motor based on the ratio of the induced voltage constant of the electric motor at a preset reference temperature and the induced voltage constant calculated by the induced voltage constant calculating means. Computation means is provided, wherein the energization control means performs energization control of the electric motor based on the magnet temperature of the field of the electric motor calculated by the magnet temperature calculation means.

  According to the present invention, since the induced voltage constant of the electric motor changes according to the field magnet temperature, the magnet temperature calculating means is calculated by the induced voltage constant at the reference temperature and the induced voltage constant calculating means. Based on the ratio with the induced voltage constant, the field magnet temperature at the time of calculating the induced voltage constant can be calculated. And the said electricity supply control means can improve the precision of the said electricity supply control by performing the said electricity supply control based on the magnet temperature of a field.

  Further, the electric motor is a double rotor type electric motor in which a first rotor and a second rotor having a plurality of fields by permanent magnets are arranged concentrically around a rotating shaft, and the first rotor and the second rotor Rotor phase difference calculating means for calculating the rotor phase difference using the rotor phase difference changing means for changing the rotor phase difference that is a relative angle between the two rotors and the induced voltage constant calculated by the induced voltage constant calculating means. And the energization control unit performs energization control of the electric motor based on the rotor phase difference calculated by the rotor phase difference calculation unit.

  According to the present invention, when the rotor phase difference is changed by the rotor phase difference changing means, the induced voltage constant of the electric motor changes accordingly. Therefore, the energization control means performs the energization control based on the current rotor phase difference calculated by the rotor phase difference calculation means, so that the accuracy of the electric current control means with high accuracy in consideration of changes in the induced voltage constant of the motor Energization control can be performed.

  The energization control means determines a level of a voltage output to the armature of the motor so as to reduce a deviation between a target current set according to a predetermined target torque and a detection current of the current detection means. And a correction for increasing the level of the voltage output to the armature of the motor determined by the current feedback control unit when the phase short circuit state is released and the energization control of the motor is resumed. Output voltage correcting means for performing the above.

  In the present invention, since the energization control cannot be performed during the all-phase short-circuit state, the output torque of the electric motor decreases from the target torque. Therefore, when the all-phase short-circuit state is canceled and the energization control is resumed, the level of the voltage output to the armature of the motor determined by the current feedback control unit is increased by the output voltage correction means. By performing the correction, the output torque of the electric motor can be quickly returned to the target torque.

  In addition, a rotation speed detection means for detecting the rotation speed of the electric motor is provided, and the resistance calculation means is a vector sum of currents flowing through the armatures of the respective phases of the electric motor when the electric motor is in the all-phase short-circuit state. The coil resistance of the armature is calculated as the all-phase short-circuit state when the phase current is rotating at a rotation speed that is assumed to be equal to or greater than a predetermined value.

  According to the present invention, when the all-phase short circuit state is established, the rate of change of the current flowing through the armature of each phase of the motor is stabilized, so that the calculation accuracy of the coil resistance by the resistance calculating means can be improved. it can.

  Further, the energization control means controls an energization amount to the armature of the electric motor so that a predetermined target torque is obtained in the energization control, and the resistance calculation means has the electric motor in the all-phase short-circuit state. When the phase current is greater than or equal to the predetermined value and the motor is rotating at a rotational speed that is assumed to have a difference between the output torque of the motor and the target torque within a predetermined range, the all-phase short-circuit state The coil resistance of the electric motor is calculated as follows.

  According to the present invention, the accuracy of calculating the coil resistance by the resistance calculating means can be increased, and the degree to which the output torque of the motor deviates from the target torque when shifting to the all-phase short-circuit state is reduced. be able to.

  Further, the energization control means controls an energization amount to the armature of the electric motor so that a predetermined target torque is obtained in the energization control, and the resistance calculation means has the electric motor in the all-phase short-circuit state. When the motor is rotating at a rotational speed that is assumed to have a difference between the output torque of the motor and the target torque within a predetermined range, the armature coil resistance of the motor is calculated as the all-phase short-circuit state. It is characterized by doing.

  According to the present invention, the degree to which the output torque of the motor deviates from the target torque when the state is shifted to the all-phase short-circuit state can be reduced.

  Embodiments of the present invention will be described with reference to FIGS. FIG. 1 is an overall configuration diagram of an electric motor control apparatus according to the present invention, FIG. 2 is a configuration diagram of a DC brushless motor including the double rotor shown in FIG. 1, and FIGS. 3 and 4 are phase differences between an outer rotor and an inner rotor. FIG. 5 is a configuration diagram of the switching circuit of the PWM arithmetic unit shown in FIG. 1, and FIG. 6 is a setting circuit for the dead time and three-phase short circuit state of the PWM arithmetic unit shown in FIG. FIG. 7 is a timing chart of the circuit shown in FIG. 6, FIGS. 8 and 9 are flowcharts of processing for calculating a constant of the motor as a three-phase short circuit state, and FIG. 10 is an explanatory diagram of execution conditions of the three-phase short circuit. 11 is a configuration diagram of a trigger signal output circuit and a sample-and-hold circuit for performing three-point current measurement, FIG. 12 is an explanatory diagram of timing for performing three-point current measurement, and FIG. Of the process to calculate Akirazu, 14 is an explanatory view of a process for calculating the resistance of the armature, FIG. 15 is a voltage vector diagram in the dq coordinate system.

  Referring to FIG. 1, a motor control device (hereinafter referred to as a motor control device) of the present invention controls the operation of a motor 1 which is a DC brushless motor having a double rotor. First, the configuration of the electric motor 1 will be described with reference to FIGS.

  As shown in FIG. 2, the electric motor 1 includes an inner rotor 11 in which the fields of the permanent magnets 11a and 11b are arranged at equal intervals along the circumferential direction, and the fields of the permanent magnets 12a and 12b along the circumferential direction. And a stator 10 having an armature 10a for generating a rotating magnetic field with respect to the inner rotor 11 and the outer rotor 13. One of the inner rotor 11 and the outer rotor 12 corresponds to the first rotor of the present invention, and the other corresponds to the second rotor of the present invention.

  The inner rotor 11 and the outer rotor 12 are both arranged concentrically so that the rotating shaft is coaxial with the rotating shaft 2 of the electric motor 1. And in the inner side rotor 11, the permanent magnet 11a which makes S pole the rotation axis 2 side, and the permanent magnet 11b which makes N pole the rotation axis 2 side are arrange | positioned alternately. Similarly, in the outer rotor 12, permanent magnets 12 a having the S pole as the rotating shaft 2 side and permanent magnets 12 b having the N pole as the rotating shaft 2 side are alternately arranged.

  The electric motor 1 includes a relative rotation mechanism (not shown) such as a planetary gear mechanism in order to change a rotor phase difference that is a phase difference between the outer rotor 12 and the inner rotor 11. By operating with 25 (see FIG. 1), the rotor phase difference can be changed. As the actuator 25, for example, an electric motor or hydraulic pressure can be used.

  The phase difference between the outer rotor 12 and the inner rotor 11 can be changed to the advance side or the retard side within a range of at least 180 electrical degrees. The state of the electric motor 1 is the permanent magnet 12a of the outer rotor 12. , 12b and the permanent magnets 11a and 11b of the inner rotor 11 are arranged so that the same poles face each other, and the permanent magnets 12a and 12b of the outer rotor 12 and the permanent magnets 11a and 11b of the inner rotor 11 are different. It can be set as appropriate between the field-strengthened state in which the poles are arranged to face each other.

  FIG. 3 (a) shows a field strengthening state. Since the directions of the magnetic flux Q2 of the permanent magnets 12a and 12b of the outer rotor 12 and the magnetic flux Q1 of the permanent magnets 11a and 11b of the inner rotor 11 are the same, they are synthesized. The magnetic flux Q3 increases. On the other hand, FIG. 3B shows a field weakening state, and the directions of the magnetic flux Q2 of the permanent magnets 12a and 12b of the outer rotor 12 and the magnetic flux Q1 of the permanent magnets 11a and 11b of the inner rotor 11 are opposite. The synthesized magnetic flux Q3 becomes smaller.

  FIG. 4 is a graph comparing the induced voltages generated in the armature of the stator 10 when the motor 1 is operated at a predetermined rotational speed in the state of FIG. 3A and the state of FIG. The axis is set to the induced voltage (V), and the horizontal axis is set to the electrical angle (degrees). In the figure, a is the state of FIG. 3A (field strengthening state), and b is the state of FIG. 3B (field weakening state). From FIG. 4, it can be seen that the level of the induced voltage is significantly changed by changing the phase difference between the outer rotor 12 and the inner rotor 11.

  Thus, the induced voltage constant Ke of the electric motor 1 can be changed by changing the phase difference between the outer rotor 12 and the inner rotor 11 to increase or decrease the magnetic flux of the field. Thereby, compared with the case where the induced voltage constant Ke is constant, the operable region for the output and the rotational speed of the electric motor 1 can be expanded. Moreover, since the loss of the electric motor 1 is reduced by the dq coordinate conversion compared to the case where the field weakening control is performed by energizing the armature on the d-axis (field axis) side, the efficiency of the electric motor can be increased. .

  Next, the configuration of the motor control device will be described with reference to FIG. The motor controller treats the motor 1 by converting it into an equivalent circuit using a two-phase DC rotating coordinate system in which the field direction is the d-axis and the direction orthogonal to the d-axis is the q-axis, and a torque command value Tr_c given from the outside. The amount of current supplied to the electric motor 1 is controlled so that a torque corresponding to the output from the electric motor 1 is output.

  The motor control device is an electronic unit including a CPU, a memory, and the like, and is based on the torque command value Tr_c, the rotational speed ω of the motor 1, and the detected value θd_e of the rotor phase difference of the motor 1. The command value Id_c of the energization amount (hereinafter referred to as d-axis armature) (hereinafter referred to as d-axis armature) and the energization amount (hereinafter referred to as q-axis current) of the q-axis side armature (hereinafter referred to as q-axis armature). ) (Current command calculation unit 50 for calculating the command value Iq_c) and current sensors 60 and 61 (corresponding to the current detection means of the present invention), and the BP (bandpass filter) 57 removes unnecessary components. Based on the detection signal and the rotor angle θm of the outer rotor 12 detected by the resolver 62, the three-phase / dq current detection value Id_s and the q-axis current detection value Iq_s are calculated by three-phase / dq conversion. dq converter 56, electric motor 1 Power control unit 91 for calculating a q-axis compensation current Iq_a for compensating for a decrease in output torque due to an increase in coil temperature Tc and field magnet temperature Tm of the armature, phase voltage of motor 1 (armature of each phase The first field control unit 92 for calculating the d-axis compensation current Id_a for flowing the d-axis current so that the combined vector of the inter-terminal voltages) falls within the voltage circle set according to the power supply voltage Vdc, the rotor position A phase difference follow-up determination unit 80 is provided for determining a d-axis correction current ΔId_vol_2 for compensating for a field weakening deficiency due to a deviation between the phase difference command value θd_c and the detected value θd_e.

  In addition, the motor control means adds the d-axis compensation current Id_a and the d-axis correction current ΔId_vol_2 to the deviation between the d-axis current command value Id_c and the detected value Id_s, and calculates an ΔId, and a q-axis current command. The adder / subtractor 51 for calculating ΔIq by adding the q-axis compensation current Iq_a to the deviation between the value Iq_c and the detected value Iq_s, the voltage applied to the d-axis armature based on ΔId and ΔIq (hereinafter referred to as d-axis voltage) The current FB (feedback) control unit 53 (the current feedback control unit and the output voltage correction unit of the present invention) determines the command value Vd_c and the command value Vq_c of the voltage applied to the q-axis armature (hereinafter referred to as q-axis voltage). Dq / 3-phase converter 54 for converting the d-axis command voltage Vd_c and the q-axis command voltage Vq_c into command values Vu_c, Vv_c, Vw_c of three-phase (U, V, W) AC voltages, Vu_c, Vv_c , Vw_c to generate three-phase AC voltage Vu, Vv, Vw And, a PWM operation unit 55 to be output to the electric motor 1.

  In addition, the current command calculation unit 50, the adder / subtractor 51, the adder / subtractor 52, the current FB control unit 53, the dq / 3 phase conversion unit 54, the PWM calculation unit 55, the BP filter 57, the three phase / dq conversion unit 56, and the power control unit 91 and the first field control unit 92 constitute the energization control means of the present invention.

Furthermore, the motor control device traces the circumference of the voltage circle c (see FIG. 15 described later) in which the phase voltage vector (the combined vector of Vd_c and Vq_c) is set according to the DC power supply voltage Vdc. A second field control unit 81 for determining a necessary field weakening current ΔId_vol, a rotor phase difference command value θd_c for producing a field weakening effect equivalent to the case where ΔId_vol is supplied to the d-axis armature The Id / θd_c substitution unit 82 that outputs to the actuator 25 and the phase difference tracking determination unit 80, and the differentiator 63 that differentiates the rotor angle θm of the motor 1 detected by the resolver 62 and outputs the angular velocity ω (the angular velocity detection of the present invention). Ld, Lq calculator 72a and Ld for calculating the d-axis armature inductance Ld and the q-axis armature inductance Lq from the d-axis current command value Id_c and the q-axis current command value Iq_c. , Lq, an inductance calculation unit 72 having an L calculation unit 72b for calculating an inductance L of the armature at the rotor angle θm, and a current detection signal Iu of the U-phase current sensor 60 via the BP filter 57, which will be described later. current I ₁ of the 3 points in 3-phase short-circuit state, I _2, the current measurement unit 71 which measures the I _3, current I ₁ measured by the current measuring section 71, I _2, I ₃ and from the inductance L of the armature R calculation unit 73 for calculating resistance R, coil temperature calculation unit 74 for calculating armature coil temperature Tc from resistance R, angular velocity ω, d-axis armature inductance Ld, resistance R, and d-axis current command value Id_c , A q-axis current command value Iq_c and a q-axis voltage command value Vq_c, an induced voltage constant calculation unit 75 that calculates an induced voltage constant Ke, a phase difference detector 77 that detects a rotor phase difference, and a phase difference detector 77 by And a magnet temperature calculating unit Tm calculating the detected value θd_e and the induced voltage constant Ke from the field of the magnet temperature Tm of over motor retardation.

  Referring to FIG. 5, PWM calculation unit 55 includes armatures 230, 231, 232 of each phase (U, V, W) of motor 1 and output unit 210 a on the high side (high potential side) of DC power supply 210. Transistors 220, 222, and 224 (corresponding to the first switching element of the present invention), the armatures 231, 231, and 232 of each phase of the electric motor 1, and the low side ( Transistors 221, 223, and 225 (corresponding to the second switching element of the present invention) for switching connection / disconnection with the output unit 210b on the low potential side) and a control signal for generating a three-phase AC voltage A PWM circuit 241 for output and a base drive circuit 240 for outputting a drive signal to the bases of the transistors 220 to 225 are provided.

  And the energization amount to the electric motor 1 is controlled by the PWM control which changes the ratio of the energization direction from the DC power supply 210 to the armatures 230, 231, 232 of each phase in each control cycle. In addition, when switching the energization direction to each armature 230, 231, 232, in order to prevent both the high-side transistor and the low-side transistor from being turned on (conducting state), an overcurrent flows through the transistor. A dead time is set in which both the high-side transistor and the low-side transistor are turned off (shut off).

  For example, for the U-phase armature 230, the output from the output terminal UH of the base drive circuit 240 to the base of the transistor 220 is set to the high level, and the output from the output terminal UL of the base drive circuit 240 to the base of the transistor 221 is output. After a dead time period in which both the transistors 220 and 221 are turned off is secured at the low level, the transistor 220 or the transistor 221 is turned on to switch the energization direction to the U-phase armature 230. The same applies to the V-phase armature 231 and the W-phase armature 232.

  The PWM calculation unit 55 sets the dead time and the three-phase armatures are all connected to the high side or the three-phase short-circuited state (corresponding to the all-phase short-circuited state of the present invention). 6 is provided with the circuit configuration shown in FIG.

  The circuit configuration shown in FIG. 6 is for a U-phase armature, a comparator 100 that compares the voltage command and a triangular wave for PWM control to determine the direction of energization to the U-phase armature, and a dead time is set. The CTC circuit 120 in which the time for the three-phase short circuit state is set, the output CMP_OUT of the comparator 100, and the Q terminal output FF1_Q of the FF (flip flop) circuit 102 are exclusive ORed. An EOR circuit 103 for determining whether or not the energization direction is switched to the U-phase armature is provided.

Next, the operation of the circuit of FIG. 6 will be described according to the timing chart shown in FIG. In the following description, the high level output of each logic circuit is “1” and the low level output is “0”. First output CMP_OUT of the comparator 100 at t ₁₀ is cut instead from 0 to 1. At this time, since the output CTC1_OUT of the CTC 120 is 0, the output of the AND circuit 101 is switched from 0 to 1. As a result, the logic of the Q terminal output FF1_Q (0) of the FF circuit 102 input to the EOR circuit 103 and the output AND1_OUT (1) of the AND circuit 101 are different from each other, so that the input to the TRG terminal of the CTC circuit 104 The output EOR_OUT of the EOR circuit 103 is switched from 0 to 1, and the output CTC2_OUT of the CTC circuit 104 becomes 1 from the next rising edge of CLK until the CLK pulse corresponding to the dead time setting value is counted.

  When CTC2_OUT is 1, since 0 is input to the AND circuits 106 and 110 via the NOT circuit 105, the output AND2_OUT of the AND circuit 106 and the output AND3_OUT of the AND circuit 110 are both 0, and the FF circuit 107. The output S1 of FF and the output S2 of the FF circuit 111 are both zero.

  Here, S1 is an ON / OFF instruction signal for the high-side transistor 220 shown in FIG. 5, and S2 is an ON / OFF instruction signal for the low-side transistor 221 shown in FIG. When S1 is 0, the transistor 220 is controlled to be in an OFF state, and when S1 is 1, the transistor 220 is controlled to be in an ON state. Similarly, when S2 is 0, the transistor 221 is controlled to be in an OFF state, and when S2 is 1, the transistor 221 is controlled to be in an ON state.

Therefore, until the set time of the dead time has elapsed, the output CTC2_OUT of the CTC circuit 104 is switched from 1 to 0, and the outputs S1 and S2 of the FF circuits 107 and 111 are switched from 1 to 0 by the next CLK (in the figure). Dt: t _{11 to} t ₁₂ ), both the high-side transistor 220 and the low-side transistor 221 are maintained in the OFF state. Then, PWM control is resumed from t _12.

Further, when the short circuit instruction signal Short_c from the short circuit determination unit 90 (see FIG. 1) is input to the TRG terminal of the CTC circuit 120 (t _{13 in the} figure), it corresponds to the set value of the short circuit time from the next rising edge of CLK. The output CTC1_OUT of the CTC circuit 120 becomes 1 until a number of CLKs are input. When CTC1_OUT is 1, since 0 is input to the AND circuit 101 via the NOT circuit 121, the output AND1_OUT of the AND circuit 101 becomes 0.

As a result, since the output EOR_OUT of the EOR circuit 103 input to the TRIG terminal of the CTC circuit 104 switches from 0 to 1, the dead time (Dt: t _{14 to} t in the figure is the same as t ₁₁ to t ₁₂ described above. ₁₅ ) is set.

When the dead time ends and the output CTC2_OUT of the CTC circuit 104 becomes 0, 1 is input to the AND circuits 106 and 110 via the NOT circuit 105, so that the Q terminal output FF1_Q of the FF circuit 102 logic is inputted to the FF circuit 107 as it is, the inverted Q terminal output FF1_Q ^- logic is inputted to the FF circuit 111 as it is.

As a result, from t ₁₅ to the dead time ends, S1 is 0, S2 becomes 1, the transistor 221 of the Low-side with the transistors 220 of the High side is controlled to the OFF state is controlled to the ON state, three-phase short-circuit state (in FIG St: t ₁₅ ~t ₁₆₎ moves to. When the timer timing of the CTC circuit 120 ends and the output CTC1_OUT switches from 1 to 0, 1 is input to the AND circuit 101 via the NOT circuit 121.

As a result, since the output AND1_OUT of the AND circuit 101 switches from 0 to 1, the output EOR_OUT of the EOR circuit 103 switches from 0 to 1. Therefore, similarly to t ₁₁ ~t ₁₂ and t ₁₄ ~t ₁₅ described above, the dead time (Dt: t ₁₆ ~t ₁₇₎ is set. Then, PWM control is resumed from t ₁₇ to the dead time has elapsed.

  Note that the circuit configuration shown in FIG. 6 is similarly provided for the V phase and the W phase. However, the output (V_Short, W_Short) is shared by the CTC 120 for setting the time for the three-phase short circuit state.

  Next, according to the flowcharts shown in FIGS. 8 to 9, calculation processing of the armature resistance R, armature coil temperature Tc, induced voltage constant Ke, and field magnet temperature Tm of the motor 1 in the motor control device. explain.

The flowchart of FIG. 8 is executed by the short-circuit determination unit 90 (see FIG. 1), and the short-circuit determination unit 90 determines whether or not the electric motor 1 is in a state that can be shifted to the three-phase short-circuit state. The short-circuit determining unit 90 determines the phase current of the motor 1 (the vector sum of the currents flowing through the armatures of the respective phases) when the rotation speed Nm (calculated from the angular velocity ω) of the motor 1 is in a three-phase short-circuit state in STEP 1. ) Is equal to or greater than I ₁₀ (corresponding to a predetermined value of the present invention).

  FIG. 10 is a graph showing the relationship between the rotational speed (Nm), the torque (Tr), and the phase current (I) when the electric motor 1 is in a three-phase short circuit state, and the horizontal axis represents the rotational speed ( Nm), the left vertical axis is set to torque (Tr), and the right vertical axis is set to the current (I) flowing through the armature of the motor 1. In FIG. 10, α indicates the relationship between the rotational speed and torque when the electric motor 1 is in a three-phase short-circuit state, and β indicates the relationship between the rotational speed and current when the motor 1 is in a three-phase short-circuit state.

Then, the rotational speed electric motor 1, the phase current when the 3-phase short circuit condition branches to STEP10 when it I ₁₀ or more to become Nm1 or more, the process proceeds to STEP2 when less than Nm1. In STEP 2, the short circuit determination unit 90 determines whether the rate of change of the torque command value Tr_c is equal to or less than a preset upper limit value. When the torque command value Tr_c is equal to or less than the upper limit value, the process branches to STEP10. On the other hand, when the torque command value Tr_c exceeds the upper limit value, the process proceeds to STEP3. In this case, the process of shifting to the three-phase short-circuit state is not executed.

  In STEP 10, the short circuit determination unit 90 calculates a torque (hereinafter referred to as a short circuit torque) TRQ1 of the electric motor 1 when the three-phase short circuit state is established according to the following equation (1).

However, TRQ1: short-circuit torque (Nm), R ₀ : preset armature resistance reference value (Ω), ω: measured electric angular velocity of the motor (erad / sec), Ke ₀ : preset Induced voltage constant (V / (erad · sec)), Ld ₀ : preset d-axis armature inductance reference value (H), Lq ₀ : preset q-axis armature inductance reference value ( H).

  Then, in subsequent STEP 11, the short circuit determination unit 90 determines whether or not the torque command value Tr_c is within the range of the short circuit torque TRC1 ± Th (TRQ1-Th ≦ Tr_c ≦ TRQ1 + Th). When the torque command value Tr_c is within the range of the short circuit torque TRC1 ± Th, the process proceeds to STEP 12, and the short circuit determination unit 90 outputs the short circuit instruction signal Short_c. On the other hand, when the torque command value Tr_c is not within the short circuit torque TRC1 ± Th in STEP11, the process branches to STEP3, and the short circuit determination unit 90 does not output the short circuit instruction signal Short_c.

  Here, the range of A in FIG. 10 indicates a range in which the rotation speed of the motor is Nm1 or more (STEP 1) and the torque command value Tr_c is within the short-circuit torque TRC1 ± Th (STEP 11). 90 is (1) when the rotational speed Nm of the electric motor 1 and the torque command value Tr_c are within the range of A in FIG. 10, and (2) the rate of change of the torque command value Tr_c is not more than the upper limit value (STEP 2). When the torque command value Tr_c is within the short circuit torque TRC1 ± Th (STEP 11), the short circuit instruction signal Short_c is output. As a result, fluctuations in the output torque of the electric motor 1 are prevented from becoming large when the three-phase short circuit state is established.

  Next, referring to FIG. 9, when a short-circuit instruction signal Short_c is input in STEP 50, a current FB control unit 53 (see FIG. 1), a PWM calculation unit 55 (see FIG. 1), and a current measurement unit 71 (see FIG. 1). ) Executes the processing from STEP 51 onward.

  STEP 51 is a process by the PWM calculation unit 55. The PWM calculation unit 55 sets all the high-side transistors 220, 222, and 224 shown in FIG. 5 to the OFF (cut-off) state by the circuit shown in FIG. The transistors 221, 223, and 225 are all turned on (conductive), and the electric motor 1 is shifted to the three-phase short circuit state.

  The subsequent STEP 52 is processing by the current measuring unit 71. The current measuring unit 71 triggers three current measurement points by a three-stage shift circuit including the FF circuits 300, 301, and 302 shown in FIG. Signals TRG1, TRG2, and TRG3 are output. Specifically, by setting the reset signal to 1 (logic high level) and the GATE signal to 0 (logic low level), the trigger signals TRG1, TRG2, and TRG3 are sequentially output in synchronization with CLK.

Then, the current measurement unit 71 converts the current detection signal Iu_s output from the current sensor 60 (see FIG. 1) into digital data and takes it in by the sample and hold circuit shown in FIG. In FIG. 11B, a sample and hold circuit comprising an operational amplifier 310, 311, a switch S1, and a capacitor 312 and an AD converter 313 are for TRG1, and when TRG1 is not output, S1 is closed and the capacitor 312 is closed. The battery is charged to the level of the current detection signal Iu_s. Then, TRG1 is kept the charge level of the output switch S1 is opened capacitor 312, the charge level of the capacitor 312 is input to the AD converter 312 via the operational amplifier 311, a detection current I ₁ during TRG1 output It is converted into digital data and imported.

Similarly, the current I ₂ at the time of output of TRG ₂ is taken in by the sample-and-hold circuit including the operational amplifiers 320 and 321, the switch S 2, and the capacitor 322 and the AD converter 323. Further, the current I ₃ at the time of output of TRG ₃ is taken in by the sample-and-hold circuit including the operational amplifiers 330 and 331, the switch S 3 and the capacitor 332 and the AD converter 333.

  FIG. 12 shows the timing of current measurement by the trigger signals TRG1, TRG2, and TRG3 in time series. The horizontal axis is set to a common time axis (t), and the vertical axis is the U-phase armature from the upper stage. Is set to ON / OFF state of high side transistors (220, 222, 224 in FIG. 5) and ON / OFF state of low side transistors (221, 223, 225 in FIG. 5). Is.

In FIG. 12, t ₂₀ to t ₂₁ and t ₂₄ to t ₂₅ are PWM control execution periods, and t ₂₂ to t ₂₃ and t ₂₆ to t ₂₇ are periods of a three-phase short circuit state. Then, during the 3-phase short-circuit state, TRG1, TRG2, TRG3 is output, a current I ₁ of the 3-point, I _2, I ₃ is measured. Here, since the current changes stably in the three-phase short-circuit state, it is suitable for obtaining the current change rate.

  The subsequent STEP 53 is processing by the inductance calculation unit 72. The Ld, Lq calculation unit 72a provided in the inductance calculation unit 72 responds by applying the Id current command value Id_c to the Ld / Id map 400 shown in FIG. Ld to be obtained is obtained, and the corresponding Lq is obtained by applying the Iq current command value Iq_c to the Lq / Iq map 401. Note that the Ld / Id map 400 and the Lq / Iq map 401 are created by experiments or computer simulation, and data of these maps are stored in a memory (not shown) in advance.

  Further, the L calculating unit 72b provided in the inductance calculating unit 72 calculates the inductance L of the electric motor 1 by the following equation (2) approximated by a sine wave curve indicated by 402 in FIG.

  Where L: inductance of motor, Ld: inductance of d-axis armature, Lq: inductance of q-axis armature, θm: rotor angle of motor.

  The subsequent STEP 55 is processing by the resistance calculation unit 73, and the resistance calculation unit 73 calculates the coil resistance R of the armature of the electric motor 1 based on the equivalent circuit of the electric motor 1 shown in FIG. In FIG. 14, 500 is a voltage applied between the terminals of the armature, 501 is the coil resistance of the armature, 502 is the inductance of the armature, and 503 is an induced voltage generated in the armature. The transient transition of the current in the circuit of FIG. 14A can be expressed by the following equation (3), where E is the instantaneous potential difference.

  Where I (t): current flowing through the armature at time t, E: instantaneous potential difference, R: coil resistance of the armature, L: inductance of the armature.

  When the above equation (3) is differentiated with respect to time t to obtain a logarithm, the following equation (4) is obtained.

As shown in FIG. 14B, when the measurement points of the currents I ₁ , I ₂ , and I _{3 in} STEP 52 are t ₁ , t ₂ , and t ₃ , the time t = (t ₁ + t ₂ ) / The left side of the above equation (4) at 2 and t = (t ₂ + t ₃ ) / 2 can be approximated by the following equations (5) and (6).

Therefore, the above equation (4) at time t = (t ₁ + t ₂ ) / 2 and t = (t ₂ + t ₃ ) / 2 can be expressed in the form of the following equations (7) and (8). it can.

  Then, when the above formulas (7) and (8) are reduced side by side, the following formula (9) is obtained, and the resistance R of the armature is calculated by the following formula (10) obtained by modifying the formula (9). be able to.

Note that I ₃ -I ₂ and I ₂ -I ₁ in the above formula (10) indicate the rate of change of current per certain time Δt (trigger signal output interval). In addition, a configuration in which a three-phase short circuit is established by the PWM calculation unit 55, a configuration in which three currents I ₁ , I ₂ and I ₃ are measured by the current measurement unit 71, and a coil resistance R is calculated by the resistance calculation unit 73. Depending on the configuration, the resistance calculation means of the present invention is configured.

Next, STEP 55 is a process by the coil temperature calculation unit 74. When the coil resistance of the armature at the reference temperature T ₀ is R ₀ and the coil temperature of the armature when the coil resistance is R is T _c , R can be expressed by the following equation (11).

Where R: armature coil temperature, T ₀ : reference temperature, T _c : armature coil temperature when resistance is R, R ₀ : armature coil resistance at reference temperature T ₀ , α _c : coil Coefficient according to the temperature characteristics of resistance.

Therefore, the coil temperature Tc of the armature when the coil resistance of the armature is R can be calculated by the above equation (11) . Note that the data of R0 and α0 are stored in advance in a memory (not shown).

  Further, the processing of STEP60 to STEP63 is executed in parallel with STEP52 to STEP55 described above. STEP 60 is a process performed by the current FB control unit 53. The current FB control unit 53 corrects the q-axis voltage command value Vq_c to increase so as to increase the torque of the motor 1 that has decreased by setting the three-phase short circuit state. I do. As a result, when the three-phase short-circuit state is canceled and the PWM control is resumed, the output torque of the electric motor 1 can be made to quickly follow the torque command value Tr_c.

  The subsequent STEP 61 is processing by the induced voltage constant calculation unit 75. The induced voltage constant calculator 75 calculates the induced voltage constant Ke by the following equation (13) obtained by modifying the following d-axis voltage equation (12) based on the dq-axis voltage vector diagram shown in FIG. FIG. 15 shows the d-axis voltage Vd and the q-axis voltage Vq with the vertical axis set to the q-axis and the horizontal axis set to the d-axis. In the figure, c corresponds to the DC power supply voltage Vdc (see FIG. 1). The voltage circle to be set.

  However, Ke: induced voltage constant, ω: angular velocity, R: armature coil resistance, Iq: q-axis current, Id: d-axis current, Vq: q-axis voltage.

  The induced voltage constant calculation unit 75 substitutes the q-axis voltage command value Vq_c for Vq in Equation 13, substitutes the d-axis current detection value Id_s for Id, and substitutes the q-axis current detection value Iq_s for Iq. The induced voltage constant Ke is calculated by substituting the inductance Ld of the d-axis armature calculated by the inductance calculating unit 72 for Ld and the coil resistance R calculated by the resistance calculating unit 73 for R.

Subsequent STEP 62 to STEP 63 are processes by the magnet temperature calculator 76. The magnet temperature calculation unit 76 acquires the detected value θd_e of the rotor phase difference by the phase difference detector 77 in STEP 62. Here, assuming that the induced voltage constant corresponding to θd_e at the reference temperature T ₀ is Ke ₀ and the temperature of the field magnet when the induced voltage constant is Ke is Tm, Ke is expressed by the following equation (14). Is done.

Where Ke is an induced voltage constant at the temperature Tm, Ke _{0 is} an induced voltage constant corresponding to θd_e at the reference temperature T ₀ , and α _m is a coefficient indicating the temperature characteristics of the field magnet.

  Therefore, the temperature Tm of the field magnet when the induced voltage constant is Ke can be calculated by the following equation (15).

In STEP 63, the magnet temperature calculation unit 76 sets the induced voltage constant Ke calculated by the induced voltage constant calculation unit 75 and the induced voltage constant Ke ₀ corresponding to θd_e at the reference temperature T _{0 to} Ke in the above equation (15). By substituting, the temperature Tm of the field magnet is calculated. Note that data of α _m and Ke ₀ and T ₀ corresponding to each rotor angle are stored in the memory in advance.

  The coil resistance R of the armature of the motor 1, the coil temperature Tc, the induced voltage constant Ke, the field magnet temperature Tm, and the rotor phase difference θd are calculated by the processing according to the flowchart of FIG. 9 described above. Referring to FIG. 1, the current command calculation unit 50 calculates a command value Id_c for the d-axis current and a command value Iq_c for the q-axis current based on the detected value θd_e of the rotor phase difference.

  Further, the power control unit 91 calculates the q-axis compensation current Iq_a based on the calculated value Tc of the coil temperature and the calculated value Tm of the field magnet temperature. The first field controller 92 calculates the d-axis compensation current Id_a based on the calculated value Ke of the induced voltage constant.

  As a result, the energization control of the electric motor 1 is executed in consideration of the effects of the coil temperature Tc, the induced voltage constant Ke, and the magnet temperature Tm, which change according to the operating state of the electric motor 1, thereby improving the accuracy of the energization control. Can be made.

In the present embodiment, the first electric motor 1a and the second electric motor 1b, which are DC brushless motors having a double rotor, are shown as electric motors of the present invention. However, a general permanent magnet having one rotor is shown. The present invention can also be applied to a rotary electric motor of a type. In this case, the phase difference detector 77 is not provided, and the induced voltage constant Ke _{0 at} the reference temperature T ₀ in the equations (14) and (15) is a fixed value.

  In the present embodiment, the calculation process of the coil resistance R by the resistance calculation unit 73, the calculation process of the coil temperature Tc by the coil temperature calculation unit 74, and the calculation process of the induced voltage constant Ke by the induced voltage constant calculation unit 75 The magnet temperature Tm calculation process by the magnet temperature calculation unit 76 is executed, but the effects of the present invention can also be obtained when only a part of these processes is executed.

  In the present embodiment, referring to FIG. 5, all of the high-side transistors 220, 222, and 224 are turned off, and the low-side transistors 221, 223, and 225 are all turned on to establish a three-phase short circuit state. However, the high-side transistors 220, 222, and 224 may all be turned on, and the low-side transistors 221, 223, and 225 may be all turned off to form a three-phase short circuit state.

  In the present embodiment, the motor control device according to the present invention is shown in which the motor 1 is converted into an equivalent circuit based on the dq coordinate system, which is the rotation coordinate of the two-phase DC, but the two-phase AC is fixed. The present invention can also be applied to the case where the circuit is converted into an equivalent circuit using the αβ coordinate system, which is the coordinate system, or the case where the three-phase alternating current is used.

The whole block diagram of the control apparatus of the electric motor of this invention. The block diagram of the DC brushless motor provided with the double rotor shown in FIG. Explanatory drawing of the effect by changing the phase difference of an outer side rotor and an inner side rotor. Explanatory drawing of the effect by changing the phase difference of an outer side rotor and an inner side rotor. The block diagram of the switching circuit of the PWM calculating part shown in FIG. The block diagram of the setting circuit of the dead time of the PWM calculating part shown in FIG. 1, and a three-phase short circuit state. 7 is a timing chart of the circuit shown in FIG. The flowchart of the process which calculates the constant of an electric motor as a three-phase short circuit state. The flowchart of the process which calculates the constant of an electric motor as a three-phase short circuit state. Explanatory drawing of the execution conditions of a three-phase short circuit. FIG. 3 is a configuration diagram of a trigger signal output circuit and a sample-and-hold circuit for performing current measurement at three points. Explanatory drawing of the timing which performs the current measurement of a point. Explanatory drawing of the process which calculates the inductance of an armature. Explanatory drawing of the process which calculates resistance of an armature. The voltage vector figure in a dq coordinate system.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Electric motor, 10 ... Stator, 11 ... Inner rotor, 11a, 11b ... Permanent magnet, 12 ... Outer rotor, 12a, 12b ... Permanent magnet, 25 ... Actuator, 55 ... PWM calculating part, 60, 61 ... Current sensor, 71 DESCRIPTION OF SYMBOLS ... Current measurement part, 72 ... Inductance calculation part, 73 ... Resistance calculation part, 74 ... Coil temperature calculation part, 75 ... Induced voltage constant calculation part, 76 ... Magnet temperature calculation part, 77 ... Phase difference detector, 220, 222, 224... High side transistor, 221, 223, 225... Low side transistor

Claims (10)

A control device for an electric motor that supplies multiphase alternating current power for generating a rotating magnetic field to an armature of each phase of a permanent magnet type rotary electric motor,
A DC power supply, a first switching element for switching between connection and disconnection between the armature and an output portion on the high potential side of the DC power supply provided individually for each phase of the armature, and a low power supply of the DC power supply The multi-phase AC power is generated by a second switching element for switching between connection and disconnection between the output unit on the potential side and the armature, and ON / OFF of the first switching element and the second switching element. And a motor control device having a power control means for performing power control of the motor,
Current detecting means for detecting a current flowing in the armature of the motor;
When the energization control of the electric motor by the energization control means is being executed and the electric motor is rotating at least at a predetermined number of revolutions or more, the first switching element and the second switching element are all turned off. and after setting the dead time, the first a switching element to a total with or first all OFF state of the switching elements are all turned OFF and the second switching element is turned oN and the Resistance calculation for calculating the coil resistance of the armature of the motor based on the rate of change of the detected current of the current detection means in the all-phase short-circuit state in which all the second switching elements are in the ON state. Means and
The motor control device according to claim 1, wherein the power supply control means performs power supply control of the motor based on the coil resistance calculated by the resistance calculation means.   2. The electric motor according to claim 1, wherein the resistance calculation unit calculates the coil resistance by using a ratio of a change rate of a detection current of the current detection unit at different time points in the all-phase short-circuit state. Control device. Coil temperature calculation means for calculating the coil temperature of the armature based on the ratio of the coil resistance of the armature at a preset reference temperature and the coil resistance calculated by the resistance calculation means,
3. The motor control device according to claim 1, wherein the energization control unit performs energization control of the electric motor based on the coil temperature calculated by the coil temperature calculation unit. Angular velocity detection means for detecting the angular velocity of the electric motor;
An induced voltage constant for calculating an induced voltage constant of the electric motor based on the angular velocity of the electric motor detected by the angular velocity detecting means, the detected current of the current detecting means, and the coil resistance calculated by the resistance calculating means. A calculation means,
The energization control means performs energization control of the electric motor based on the induced voltage constant calculated by the induced voltage constant calculation means. Electric motor control device. Magnet temperature calculation means for calculating the field magnet temperature of the electric motor based on the ratio of the induced voltage constant of the electric motor at the preset reference temperature and the induced voltage constant calculated by the induced voltage constant calculation means. With
5. The motor control device according to claim 4, wherein the energization control means performs energization control of the electric motor based on the magnet temperature of the field of the electric motor calculated by the magnet temperature calculation means. The electric motor is a double rotor type electric motor in which a first rotor and a second rotor having a plurality of fields by permanent magnets are arranged concentrically around a rotating shaft,
Rotor phase difference changing means for changing a rotor phase difference that is a relative angle between the first rotor and the second rotor, and using the induced voltage constant calculated by the induced voltage constant calculating means, the rotor phase difference Rotor phase difference calculating means for calculating
5. The motor control apparatus according to claim 4, wherein the energization control unit performs energization control of the motor based on the rotor phase difference calculated by the rotor phase difference calculation unit. The energization control unit is a current that determines a level of a voltage output to the armature of the motor so as to reduce a deviation between a target current set according to a predetermined target torque and a detection current of the current detection unit. A feedback control unit;
Output voltage correction means for performing correction to increase the level of the voltage output to the armature of the motor determined by the current feedback control unit when the phase short-circuit state is released and the energization control of the motor is resumed. The motor control device according to any one of claims 1 to 6, further comprising: A rotation speed detecting means for detecting the rotation speed of the electric motor;
The resistance calculating means rotates at a rotation speed at which a phase current, which is a vector sum of currents flowing through the armatures of the respective phases of the motor, becomes equal to or greater than a predetermined value when the motor is in an all-phase short-circuit state. The motor control device according to any one of claims 1 to 7, wherein a coil resistance of an armature is calculated as the all-phase short-circuit state when the motor is in a short-circuit state. The energization control means controls the energization amount to the armature of the electric motor so that a predetermined target torque is obtained in the energization control,
The resistance calculating means is assumed that the phase current is equal to or greater than the predetermined value when the motor is in the all-phase short-circuit state, and that the difference between the output torque of the motor and the target torque is within a predetermined range. The motor control device according to claim 8, wherein a coil resistance of the motor is calculated as the all-phase short-circuit state when the motor rotates at a rotating speed. The energization control means controls the energization amount to the armature of the electric motor so that a predetermined target torque is obtained in the energization control,
When the motor is rotating at a rotational speed at which the difference between the output torque of the motor and the target torque is assumed to be within a predetermined range when the electric motor is in the all-phase short-circuit state, The motor control device according to claim 1, wherein a coil resistance of an armature of the motor is calculated as the all-phase short-circuit state.

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