JP6012822B1 - Power converter - Google Patents

Abstract

An object of the present invention is to provide a power conversion device including an isolated resonance converter capable of discharging a residual charge of a capacitor on an input side. In a resonant converter circuit 200, semiconductor switching elements 6 and 7 connected in series are connected in parallel with a smoothing capacitor 5 of a step-down converter circuit 100 to constitute a half-bridge inverter, and the output includes The resonance capacitor 8, the resonance reactor 9, and the primary winding of the transformer 10 are connected in series. The center tap of the secondary winding of the transformer 10 is connected to the output capacitor 13 and the negative electrode of the voltage source load 52. The other secondary winding is connected to the positive electrode of the output capacitor 13 so as to be rectified by the rectifier diodes 11 and 12. Further, the control unit 20 operates the semiconductor switching elements 6 and 7 to generate a switching loss and discharge the residual charge of the smoothing capacitor 5. [Selection] Figure 1

Description

  The present invention relates to a power conversion device capable of discharging electric charges accumulated in a capacitor provided in an input stage of an insulating resonance converter.

  As a conventional power converter, for example, in the switching power supply of Patent Document 1, a non-insulated converter that boosts an input power supply voltage and outputs a DC voltage, and a DC voltage output from the non-insulated converter is output to a load. Insulated bridge converters, and those having a smoothing capacitor in the middle of each converter are disclosed. The insulated bridge converter uses an LLC (two inductors (LL) and a capacitor (C)) resonant converter system, drives the converter at a fixed on-duty ratio and at a fixed switching frequency, and outputs the output voltage of the switching power supply device. Is adjusted by PWM (Pulse Width Modulation) control of the non-insulated converter in the previous stage. In this way, the converter has a two-stage configuration, and the switching control of the preceding converter can control the input voltage fluctuation to the succeeding converter. Therefore, even if the input voltage range is set to a large value, Efficient voltage conversion is possible.

  Further, in the electric motor power supply device disclosed in Patent Document 2, a booster circuit that boosts the voltage of the low-voltage DC power supply, a first capacitor connected to the input side of the booster circuit, and the boosted high voltage are converted into AC power. A drive circuit for driving the motor and a second capacitor connected to the input side of the drive circuit, and when the low-voltage DC power supply is cut off, the booster circuit is controlled to store the accumulated charge in the first capacitor. The voltage is boosted to a predetermined voltage, the second capacitor is charged, and the d-axis current is supplied to the electric motor with the electric power stored in the second capacitor, whereby the electric charges of both capacitors are discharged. Since the charge of the first capacitor is boosted to charge the second capacitor, and the charge of the second capacitor is consumed by the drive circuit, the charge of both capacitors can be discharged without a discharge resistor, In addition, the boosting increases the switching loss of the drive circuit to promote discharge.

  In addition, the power conversion device disclosed in Patent Document 3 includes a capacitor, a power conversion unit, a gate drive unit, and a control device, and the control device responds to a collision of the vehicle detected by the collision detection unit. Thus, the gate driver is controlled so as to increase the switching loss of the switching element. Thereby, the residual electric charge accumulate | stored in the smoothing capacitor in a power converter device can be discharged quickly at the time of a vehicle collision.

JP 2013-258860 A JP 2007-195352 A International Publication No. 2010-131353

However, in the conventional switching power supply of Patent Document 1, when it is desired to discharge the residual charge of the smoothing capacitor when output control is stopped, for example, a battery is connected as a load or the load is disconnected due to a failure or the like. If the power of the smoothing capacitor is to be transmitted to the load side, the voltage is applied to the output terminal, so that the power cannot be transmitted, and the electric charge remains in the smoothing capacitor. As described above, even if the residual charge of the input side capacitor of the LLC resonant converter is immediately discharged, the normal switching control cannot cope with it, and there is a problem that another discharge method needs to be prepared.

  Moreover, in the electric power supply device of the electric motor of patent document 2, and the power converter device of patent document 3, the conduction loss and the switching loss are increased by controlling on / off of the switching element of the power converter device, and the residual stored in the capacitor Although the electric charge is discharged, a method for discharging the electric charge is not shown in the power conversion device including the LLC resonant converter. When a battery is connected as a load of an LLC resonant converter with a capacitor connected to the input side, even if power is transmitted to the load side by the on / off control of the LLC resonant converter, a certain amount of charge is applied to the input capacitor. In this state, even if the on / off control for power transmission is continued, only the exciting current flows, and the conduction loss becomes very low. Furthermore, since the LLC resonant converter is designed to perform soft switching by resonance in normal operation, even if the configuration in which the gate resistance for driving the switching element is switched as shown in Patent Document 3, switching loss is adopted. Hardly occurs. As described above, the LLC resonant converter has a problem that the conduction loss and the switching loss are low only by simple on / off control, and the residual charge of the capacitor cannot be discharged in a short time.

  The present invention has been made to solve the above-described problems, and in a power conversion device including an LLC resonant converter in which a capacitor is connected to the input side, it is not necessary to add a new component, and the input side An object of the present invention is to provide a power converter capable of discharging the residual charge of the capacitor.

  In order to solve the above problems, an electric power converter according to the present invention includes an isolated resonant converter that converts a DC input voltage into a DC output voltage of a different voltage, and a capacitor connected to the input side of the isolated resonant converter. A control unit that controls the operation of the isolated resonant converter, wherein the isolated resonant converter is disposed in parallel with the capacitor and is connected in series to each other and the second switching element. A switching element, a transformer having a primary winding and a secondary winding, a resonant capacitor and a resonant reactor connected in series to the second switching element and the primary winding of the transformer and forming a closed circuit And a rectifier circuit connected to the secondary side of the transformer, and the control unit includes the first switching element and the second switching element. The DC output voltage is controlled by alternately turning on / off the element with an on-duty ratio or switching frequency with a dead time in between, and when the output of the DC output voltage is stopped, the DC output voltage is controlled. The capacitor is discharged by performing on / off control alternately with a dead time, an on-duty ratio or a switching frequency different from the above.

  According to the power conversion device of the present invention, in the power conversion device including the LLC resonant converter having a capacitor connected to the input side, a switching loss is generated by on / off control of the switching element of the LLC resonant converter, thereby providing a new There is an effect that the residual charge of the capacitor can be discharged in a short time without adding any parts.

1 is a configuration diagram of a power conversion device according to a first embodiment. FIG. 4 is a waveform diagram of each voltage and current during operation of the resonant converter circuit according to the first embodiment. 3 is a diagram illustrating a current path of a resonant converter circuit according to Embodiment 1. FIG. 6 is a diagram illustrating a relationship between a normalized frequency and a gain of the resonant converter circuit according to Embodiment 1. FIG. It is a figure which shows each voltage and electric current waveform at the time of making the switching frequency of the resonant converter circuit in Embodiment 1 lower than a series resonant frequency. It is a figure which shows each voltage and electric current waveform at the time of making the switching frequency of the resonant converter circuit in Embodiment 1 higher than a series resonant frequency. FIG. 3 is a diagram illustrating each voltage and current waveform during residual charge discharge of the smoothing capacitor in the first embodiment. FIG. 3 is a diagram illustrating a time transition waveform of a voltage and a control parameter when discharging a residual charge of a smoothing capacitor in the first embodiment. It is a figure which shows the time transition waveform of the voltage at the time of the residual charge discharge of the smoothing capacitor in Embodiment 2, and a control parameter.

  Hereinafter, the details of the configuration and operation of the power conversion device according to the embodiment of the present invention will be described with reference to FIGS. 1 to 9.

Embodiment 1 FIG.
<Device configuration>
FIG. 1 is a configuration diagram of the power conversion device according to the first embodiment, FIG. 2 is a waveform diagram of each voltage and current during operation of the resonant converter circuit, and FIG. 3 shows a current path of the resonant converter circuit. FIG. 4 is a diagram showing the relationship between the normalized frequency and the gain, and FIG. 5 shows each voltage and current waveform when the switching frequency of the resonant converter circuit is made lower than the series resonant frequency. FIG. 6 is a diagram illustrating voltage and current waveforms when the switching frequency of the resonant converter circuit is higher than the series resonant frequency. Further, FIG. 7 is a diagram showing the voltage and current waveforms during the residual charge discharge of the smoothing capacitor, and FIG. 8 is a diagram showing the time transition waveforms of the voltage and control parameters during the residual charge discharge of the smoothing capacitor. .

  First, the configuration of the power conversion device according to Embodiment 1 will be described with reference to FIG. The power conversion apparatus 1000 includes a step-down converter circuit 100 that is a DC converter and a resonance converter circuit 200 that is an insulated LLC resonance converter connected in series. Furthermore, the power conversion apparatus 1000 includes a control unit 20 for controlling the step-down converter circuit 100 and the resonant converter circuit 200.

Next, the details of the circuits and devices constituting the power conversion apparatus 1000 will be described.
First, the power converter 1000 converts the input voltage Vi from the DC power source 51 into an arbitrary DC voltage by the step-down converter circuit 100, and outputs the output voltage Vo from the resonance converter circuit 200 to the voltage source load 52 such as a battery.

  The step-down converter circuit 100 includes an input capacitor 1, a semiconductor switching element 2 such as a MOSFET (Metal Oxide Field Effect Transistor), a diode 3, a step-down reactor 4, and a smoothing capacitor 5. The input capacitor 1 is connected in parallel with the DC power supply 51, and the semiconductor switching element 2 has a drain terminal connected to the positive terminal of the DC power supply 51 and a source terminal connected to the cathode terminal of the diode 3 and one end of the step-down reactor 4. Connected with. The other end of step-down reactor 4 is connected to the positive electrode of smoothing capacitor 5. The anode terminal of the diode 3 and the negative electrode of the smoothing capacitor 5 are connected to the negative electrode side terminal of the DC power supply 51.

  The resonant converter circuit 200 includes semiconductor switching elements 6 and 7 such as MOSFETs, a resonant capacitor 8, a resonant reactor 9, a transformer 10, rectifier diodes 11 and 12 that are rectifier circuits, and an output capacitor 13. ing. The semiconductor switching elements 6 and 7 connected in series are connected in parallel with the step-down converter circuit 100 to form a half-bridge inverter. The output of the half-bridge inverter includes a resonant capacitor 8, a resonant reactor 9, and a transformer 10. Primary windings are connected in series. The secondary winding of the transformer 10 has a center tap. The center tap is connected to the negative electrode of the output capacitor 13 and the negative terminal of the voltage source load 52. The other secondary winding of the transformer 10 is The output is connected to the positive electrode of the output capacitor 13 so as to be rectified by the rectifier diodes 11 and 12.

  In addition, the power conversion apparatus 1000 uses the input voltage measurement circuit 21 that measures the input voltage Vi (= the voltage Vc of the input capacitor 1) input from the DC power supply 51 to the step-down converter circuit 100, and the DC voltage Vdc of the smoothing capacitor 5. A smoothing capacitor voltage measuring circuit 22 for measuring and an output voltage measuring circuit 23 for measuring the output voltage Vo output from the resonant LLC converter circuit are provided.

  The control unit 20 acquires the detected voltage values in the input voltage measurement circuit 21, the smoothing capacitor voltage measurement circuit 22, and the output voltage measurement circuit 23 via the signal lines 41, 42, and 43, and the control lines 31, 32, On / off control is performed on the semiconductor switching elements 2, 6, and 7 via 33.

  A diode is connected in antiparallel between the source and drain of each of the semiconductor switching elements 2, 6, 7. These diodes are built in the semiconductor switching elements 2, 6, 7, respectively. It may be. Further, as the semiconductor switching elements 2, 6, and 7, an IGBT (Insulated Gate Bipolar Transistor) may be used in addition to the MOSFET. The resonant reactor 9 may be formed by a leakage inductance of the transformer 10.

  In the power conversion device shown in FIG. 1, the resonance capacitor 8, the resonance reactor 9, and the transformer 10 are connected in this order from the connection point of the semiconductor switching elements 6 and 7. The resonance reactor 9 and the transformer 10 may be connected in series.

<Operation>
Next, the operation of the power conversion apparatus 1000 of the present embodiment configured as described above will be described. First, the operating principles of the step-down converter circuit 100 and the resonant converter circuit 200 will be described.

  In the operation of the step-down converter circuit 100, the control unit 20 controls the input current and the DC voltage Vdc of the smoothing capacitor 5 by PWM controlling the semiconductor switching element 2. As a specific PWM operation, the smoothing capacitor 5 is charged while exciting the step-down reactor 4 by the differential voltage between the input voltage Vi and the DC voltage Vdc of the smoothing capacitor 5. Further, by turning off the semiconductor switching element 2, the smoothing capacitor 5 is charged while the step-down reactor 4 is demagnetized by the DC voltage Vdc of the smoothing capacitor 5. Since the operation is a step-down operation, the input voltage Vi <the DC voltage Vdc.

  Subsequently, in the operation of the resonant converter circuit 200, the control unit 20 switches the operation of the semiconductor switching elements 6 and 7 of the resonant converter circuit 200 with the on-duty ratios D1 and D2 of approximately 50% across the dead time Td. Then, output control is performed by on / off control.

  FIG. 2 shows voltage and current waveform diagrams during the switching operation of the resonant converter circuit 200. Here, in the case where the operation is performed at the switching frequency fsw in which the time obtained by adding the dead time td to the half period of the series resonance frequency fsr of the series resonance circuit composed of the resonance capacitor 8 and the resonance reactor 9 is a half period. The waveform is shown.

  In FIG. 2, Vgs6 and Vgs7 are gate voltages applied between the gates and sources of the semiconductor switching elements 6 and 7, and Vds6 and Vds7 are voltages applied between the drains and sources of the semiconductor switching elements 6 and 7, respectively. , Vtr1 is a voltage applied to the primary side of the transformer 10, ILr is a current flowing through the resonance capacitor 8 and the resonance reactor 9 (hereinafter referred to as “resonance current”), and ILm is an excitation of the transformer 10. ID11 and ID12 show the excitation current flowing through the inductance Lm, and the waveforms of the current flowing through the rectifier diodes 11 and 12 on the secondary side of the transformer 10, respectively.

  In FIG. 2, t2 and t6 indicate times when the semiconductor switching element 6 is turned on, and t3 and t7 indicate times when the semiconductor switching element 6 is turned off. In addition, t1 and t5 indicate times when the semiconductor switching element 7 is turned off, and t4 and t8 indicate times when the semiconductor switching element 7 is turned on. A dead time td is provided between when one of the semiconductor switching elements 6 and 7 is turned off and when the other is turned on. Further, the direction of current flowing from the primary side to the transformer 10 from the resonance capacitor 8 is positive.

  3A to 3D are diagrams showing current paths when the semiconductor switching elements 6 and 7 of the resonant converter circuit 200 are on / off corresponding to each time shown in FIG. is there.

  From time t1 to time t2 shown in FIG. 3A, immediately after the semiconductor switching element 7 is turned off, the resonance current ILr flows in the order of the transformer 10 → resonance reactor 9 → resonance capacitor 8 → semiconductor switching element 6. At this time, the charge of the parasitic capacitance (not shown) of the semiconductor switching element 6 is discharged, and the parasitic capacitance (not shown) of the semiconductor switching element 7 is charged by commutation. After the drain-source voltage Vds6 of the semiconductor switching element 6 becomes 0V, a current flows through the body diode of the semiconductor switching element 6.

  Further, at time t2 to t3 shown in FIG. 3B, since the current flows through the body diode of the semiconductor switching element 6 until just before, the drain-source voltage Vds6 is 0V. Therefore, when the semiconductor switching element 6 is turned on, ZVS (Zero Voltage Switching) is established. After the semiconductor switching element 6 is turned on, the resonance current ILr flows in the order of the semiconductor switching element 6 → resonance capacitor 8 → resonance reactor 9 → transformer 10. Note that the solid line in the figure is the current path at times t2 to t3, and the dotted line is the current path of the resonance current ILr that has flowed until just before.

  In addition, at times t3 to t4 shown in FIG. 3C, immediately after the semiconductor switching element 6 is turned off, the resonance current ILr is the resonance capacitor 8 → resonance reactor 9 → transformer 10 → the body diode of the semiconductor switching element 7. It flows in the order of the route. At this time, the charge of the parasitic capacitance (not shown) of the semiconductor switching element 7 is discharged, and the parasitic capacitance (not shown) of the semiconductor switching element 6 is charged by commutation. After the drain-source voltage Vds7 of the semiconductor switching element 7 becomes 0V, a current flows through the body diode of the semiconductor switching element 7.

  Also, from time t4 to t5 shown in FIG. 3D, since the current flows through the body diode of the semiconductor switching element 7 until just before, the drain-source voltage Vds7 is 0V. Therefore, ZVS is established when the semiconductor switching element 7 is turned on. After the semiconductor switching element 7 is turned on, the resonance current ILr flows in the order of the semiconductor switching element 7 → the transformer 10 → the resonance reactor 9 → the resonance capacitor 8. Note that the solid line in the figure is the current path at times t4 to t5, and the dotted line is the current path of the resonance current ILr that has flowed until just before.

  Generally, in the above operation, the dead time Td, the switching frequency fsw, the exciting inductance Lm of the transformer 10 and the like are designed so that ZVS is established.

  Although not shown in FIG. 3, when the differential current between the resonance current ILr and the excitation current ILm flows to the secondary side of the transformer 10 and ILr> ILm, the current ID11 flows to the rectifier diode 11. , ILm> ILr, current ID12 flows through rectifier diode 12.

  Next, the relationship between the frequency and gain of the LLC resonant converter will be described. Here, the gain refers to the input / output voltage ratio of the resonant converter circuit 200 which is an LLC resonant converter.

  FIG. 4 is a diagram illustrating the relationship between the normalized frequency fn (normalized frequency) and the gain G in the resonant converter circuit 200. Here, the normalized frequency fn is expressed by switching frequency fsw / series resonance frequency fsr.

  As is clear from FIG. 4, the gain characteristic with respect to the switching frequency fsw varies depending on the inductance ratio Ln (= Lm / Lr) determined by the ratio of the exciting inductance Lm of the transformer 10 and the inductance Lr of the resonant reactor 9. In FIG. 4, by designing the inductance ratio Ln to be small, the variation of the gain G with respect to the variation of the switching frequency fsw becomes large. By utilizing this characteristic, the LLC resonant converter generally adjusts the gain G by controlling the switching frequency fsw and controls the output voltage Vo.

  Further, it can be seen from FIG. 4 that the gain G changes less as the inductance ratio Ln increases. Normally, the gain characteristic shown in FIG. 4 also changes depending on the size of the load, but if the inductance ratio Ln is sufficiently large, the influence on the gain characteristic due to the load fluctuation is small. That is, if the inductance ratio Ln is large, it is possible to obtain a stable output voltage Vo against sudden load fluctuations.

  In the present embodiment, the inductance ratio Ln of the resonant converter circuit 200 is made sufficiently large, and the excitation inductance Lm and the inductance Lr of the resonant reactor 9 are determined so that the gain fluctuation with respect to the frequency fluctuation is reduced.

<Control method>
Below, the output control in the steady state of the power converter device 1000 is demonstrated. First, a method for controlling the resonant converter circuit 200 will be described.

  As shown in FIG. 2, the control unit 20 has a series resonance of a series resonance circuit having an on-duty ratio D1 and D2 of approximately 50% and a resonance capacitor 8 and a resonance reactor 9 with a dead time td interposed therebetween. The semiconductor switching elements 6 and 7 are alternately turned on at a switching frequency fsw whose half cycle is the time obtained by adding the dead time td to the half cycle of the frequency fsr. That is, the resonant converter circuit 200 is operated at the fixed on-duty ratios D1 and D2 and the fixed switching frequency fsw.

In general, it is recommended that the LLC resonant converter be controlled so that the switching frequency fsw is equal to the series resonant frequency fsr in order to increase the efficiency by ZCS (Zero Current Switching). Since the on-time of the semiconductor switching element is shortened by the dead time Td, turn-off loss occurs. Therefore, as represented by the following equation (1), the switching frequency fsw is determined so that the half cycle of the switching frequency fsw is equal to the time obtained by adding the dead time Td to the half cycle of the series resonance frequency fsr. To do.

  For comparison, FIG. 5 shows voltage and current waveforms when the switching frequency fsw of the semiconductor switching elements 6 and 7 is sufficiently lower than the series resonance frequency fsr. FIG. 6 shows the switching frequency fsw from the series resonance frequency fsr. Each voltage-current waveform when sufficiently high is also shown.

  Next, a method for controlling the step-down converter circuit 100 will be described. As described above, since the control unit 20 controls the resonant converter circuit 200 with the fixed on-duty ratios D1 and D2 and the fixed switching frequency fsw, the gain G as the LLC resonant converter is constant. Therefore, the control unit 20 performs PWM control on the semiconductor switching element 2 of the step-down converter circuit 100 and adjusts the DC voltage Vdc of the smoothing capacitor 5 to control the output voltage Vo so as to approach the target value.

Assuming that the turns ratio of the transformer 10 of the resonant converter circuit 200 is N: 1: 1 (the primary side is N), the following expression (2) is established as a relational expression between the DC voltage Vdc of the smoothing capacitor 5 and the output voltage Vo.

  From equation (2), it can be seen that the DC voltage Vdc and the output voltage Vo are expressed as a simple proportional relationship. The control unit 20 obtains a difference between the target value of the output voltage Vo or the target value of the DC voltage Vdc converted from the target value and the voltage value obtained from the output voltage measuring circuit 23 or the smoothing capacitor voltage measuring circuit 22, The semiconductor switching element 2 of the step-down converter circuit 100 is subjected to PWM control by feedback control by PI control or the like, thereby performing follow-up control of the output voltage Vo.

<Discharge method of residual charge>
Hereinafter, a method for discharging the residual charge of smoothing capacitor 5 of power conversion device 1000 according to the present embodiment will be described. Since the voltage source load 52 is connected to the power conversion device 1000, when the DC voltage Vdc is lower than 2N × Vo shown in the equation (2), power cannot be transmitted to the secondary side of the transformer 10, In normal output control, charges remain in the smoothing capacitor 5. FIG. 7 shows each voltage and current waveform in the switching operation of the resonant converter circuit 200 when the residual charge is discharged under the condition of Vdc <2N × Vo.

  In FIG. 7, the semiconductor switching elements 6 and 7 are on / off controlled with on-duty ratios D1 and D2 of approximately 50% across the dead time td / 2, as in the normal output control. At this time, the dead time td / 2 (time t3 to t4 after the semiconductor switching element 6 is turned off) is set by setting td / 2, which is a half of the dead time td during normal output control, as the dead time Td. , T7 to t8), the semiconductor switching element 7 is turned on before the drain-source voltage Vds7 of the semiconductor switching element 7 is reduced to 0 V, and ZVS is not established, resulting in a switching loss. Similarly, for the dead time td / 2 (time t1 to t2, t5 to t6) after the semiconductor switching element 7 is turned off, the semiconductor switching element 6 is turned on before the drain-source voltage Vds6 decreases to 0V. ZVS is not established.

  Further, the switching frequency fsw of the semiconductor switching elements 6 and 7 is set to 2 fsr, which is approximately twice the series resonance frequency fsr when the switching frequency fsw during normal output control is set. As a result, the peak value of the excitation current ILm decreases compared to the normal time, that is, the resonance current ILr for decreasing the drain-source voltages Vds6 and Vds7 decreases during the dead time, and ZVS is not established. The operation causes switching loss.

  As described above, in the discharging operation, by switching the semiconductor switching elements 6 and 7 by shortening the dead time Td and increasing the switching frequency fsw, a switching loss occurs, and the charge of the smoothing capacitor 5 can be discharged. Note that either one of the shortening of the dead time Td and the increase of the switching frequency fsw have the effect of increasing the switching loss. By combining both, the switching loss can be increased more effectively.

  Next, the sequence of the discharge process will be described. 8 shows the output voltage Vdc when discharging the residual charge of the smoothing capacitor 5 when the output control is stopped, the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7, the on-duty ratio D3 of the semiconductor switching element 2, and the semiconductor switching element. The time transition waveforms of the dead time Td of 6 and 7 and the switching frequency fsw are shown.

  In FIG. 8, the period from time t10 to t12 shows the discharge period of the smoothing capacitor 5 in the present embodiment, and the control unit 20 enters the load side from the resonant converter circuit 200 at time t10 to t11. The current suppression process is performed, and the discharge process of the residual charge of the smoothing capacitor 5 by the resonant converter circuit 200 is performed at times t11 to t12. When the output is stopped, the semiconductor switching element 2 keeps turning off to disconnect the DC power source 51 and the smoothing capacitor 5 and efficiently discharge only the charge of the smoothing capacitor 5 by residual charge discharge at time t10 to t12. Therefore, the on-duty ratio D3 of the semiconductor switching element 2 is set to 0%.

In the period from time t10 to t11, the control unit 20 reduces the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 of the resonant converter circuit 200 to an on-duty ratio of approximately 50% obtained by subtracting the dead time td / 2 minutes. Increase gradually. Note that the on-duty ratio of about 50% is simply indicated as 0.5 in FIG. In addition, as described above, dead time T
It is assumed that d is td / 2 and the switching frequency fsw is 2 fsr.

  When the DC voltage Vdc is higher than 2N × Vo shown in the equation (2) and the semiconductor switching elements 6 and 7 are operated at the on-duty ratios D1 and D2 of about 50%, power is rapidly transmitted, and the load Since an inrush current is generated on the side, the inrush current can be suppressed by gradually increasing the on-duty ratios D1 and D2 before starting the discharge process. In FIG. 8, during the period from time t10 until the DC voltage Vdc matches 2N × Vo, Vdc decreases due to the transmission of power to the load side, and thereafter, charge discharge is caused by switching loss and Vdc decreases. It becomes the operation to do.

  During the period from time t11 to t12, the control unit 20 sets the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 of the resonant converter circuit 200 to an on-duty ratio of approximately 50% obtained by subtracting the dead time Td, and the dead time Td is set to td / 2, and the switching frequency fsw is set to 2 fsr. Accordingly, as described above, the operation can be performed in a state where the switching loss is increased as compared with the normal output control, and the residual charge of the smoothing capacitor 5 can be discharged.

  As described above, according to the power conversion device according to the first embodiment, in normal output control, resonant converter circuit 200 that operates by soft switching in which ZCS and ZVS are established is switched by a method different from normal output control. As a result, a switching loss is generated, and the residual charge of the capacitor on the input side of the resonant converter circuit 200 can be discharged in a short time. Further, it is possible to discharge residual charges without adding new parts.

  Further, the semiconductor switching elements 6 and 7 are alternately turned on / off with the dead time Td interposed therebetween, whereby the drain-source capacitance of the semiconductor switching elements 6 and 7 is charged and discharged, resulting in switching loss. This makes it possible to discharge residual charges.

  Further, by adjusting parameters relating to the switching operation such as the dead time Td, the switching frequency fsw, and the on-duty ratios D1 and D2 to values different from the output control, it is possible to more effectively adjust the operation so that ZVS is not established. .

  In addition, the dead time Td of the semiconductor switching elements 6 and 7 is set so that ZVS is established in order to improve efficiency in output control, but by reducing the dead time Td during discharge, It is possible to adjust in a direction in which ZVS is difficult to be established, and an operation that effectively generates a switching loss can be achieved. In the present embodiment, the dead time Td during discharge is set to td / 2, but is not limited to this value.

  In addition, as shown in FIG. 2, in the output control of the resonant converter circuit 200, the switching frequency fsw is set to a switching frequency fsw in which the time obtained by adding the dead time Td to the half cycle of the series resonance frequency fsr is approximated to ZCS. Therefore, in general, by designing such that ZVS is established at such a switching frequency fsw, ZCS and ZVS can be established at the same time to achieve high efficiency. Therefore, at the time of discharging, by increasing the switching frequency fsw, the resonance current ILr for decreasing the drain-source voltages Vds6 and Vds7 can be decreased at the dead time, and the adjustment can be made in a direction in which ZVS is difficult to be established. This is possible and can be an operation that effectively generates a switching loss. In the present embodiment, the switching frequency fsw is 2 fsr, but is not limited to this value.

  Further, by setting the switching frequency fsw of the semiconductor switching elements 6 and 7 to a fixed value at the time of discharging the residual charge, it is possible to generate the switching loss most effectively and continue the discharge under the condition that the smoothing capacitor 5 can be discharged. Can do.

  In addition, when the residual charge is discharged, the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 are gradually increased to provide a period for suppressing the inrush current to the load side. Stress can be suppressed and failure can be prevented.

  Further, even if the step-down converter circuit 100 is provided in the previous stage of the resonant converter circuit 200, the residual charge of the smoothing capacitor 5 can be discharged by the on / off control of the semiconductor switching elements 6 and 7. Further, by turning off the semiconductor switching element 2 of the step-down converter circuit 100, the DC power source 51 and the smoothing capacitor 5 can be disconnected, and only the electric charge of the smoothing capacitor 5 can be discharged efficiently.

  In the first embodiment, the case where the dead time Td at the time of discharge is shortened from that at the time of output control is described. However, the present invention is not limited to this and may be extended in reverse. During the dead time of the semiconductor switching elements 6 and 7, the drain-source voltages Vds6 and Vds7 are reduced by the energy of the resonant reactor 9 and the excitation inductance Lm. However, when the dead time T continues, the drain-source voltage Vds6 is caused by resonance. , Vds7 increases. As a result, by setting the dead time Td so as to be switched at a timing when Vds6 and Vds7 increase, the switching loss is effectively generated, and the residual charge of the smoothing capacitor 5 can be discharged.

  Further, in the first embodiment, the semiconductor switching elements 6 and 7 have the on-duty ratios D1 and D2 of approximately 50% obtained by subtracting the dead time Td, and the on-duty ratios D1 and D2 at the time of output control and discharging are set. Although it is the same, switching loss can be effectively generated also by increasing and decreasing the on-duty ratios D1 and D2 at the time of discharge as compared with those at the time of output control. Decreasing the on-duty ratios D1 and D2 is equivalent to extending the dead time Td, and increasing the on-duty ratios D1 and D2 is equivalent to reducing the dead time Td. The effect of generating a loss is obtained. Further, when the on-duty ratios D1 and D2 are close to 50% or exceed 50%, the semiconductor switching elements 6 and 7 are short-circuited. However, the residual charge of the smoothing capacitor 5 can also be discharged. is there.

  In the first embodiment, the semiconductor switching elements 6 and 7 have the on-duty ratios D1 and D2 of approximately 50% obtained by subtracting the dead time Td. However, the semiconductor switching elements 6 and 7 have a semiconductor switching function higher than the on-duty ratio D3 of the semiconductor switching element 2. By increasing the on-duty ratio D1 of the element 6, switching loss can be effectively generated. When the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 are set to 50%, the voltage of the resonant capacitor 8 is a DC voltage Vdc × 0.5. For example, the on-duty ratio D1 of the semiconductor switching element 6 is Assuming that 80% and the on-duty ratio D2 of the semiconductor switching element 7 is 20%, the voltage of the resonant capacitor 8 is the DC voltage Vdc × 0.8. When the voltage of the resonance capacitor 8 is increased, the excitation current ILm and the resonance current ILr generated by the on / off control of the semiconductor switching elements 6 and 7 are decreased, so that the drain-source voltages Vds6 and Vds7 are decreased in the dead time T. The operation becomes slow, and it becomes difficult to establish ZVS and causes a switching loss.

  In the first embodiment, in the output control of the resonant converter circuit 200, the switching frequency fsw is set to a half period obtained by adding the dead time Td to the half period of the series resonance frequency fsr. Although the method for generating the switching loss with the high switching frequency fsw has been described, the loss can be effectively generated even with the switching frequency fsw lower than this. If the switching frequency fsw is set low, the excitation current ILm and the resonance current ILr become high when the semiconductor switching elements 6 and 7 are turned on, so that the switching loss and the conduction loss can be increased.

  In the first embodiment, the inrush current at the time of discharging is suppressed by gradually increasing the on-duty ratios D1 and D2. However, a method of suppressing the inrush current by gradually decreasing the dead time Td may be used. By gradually decreasing the period corresponding to the ON period in which the on-duty ratios D1 and D2 are 50% as the initial value of the dead time Td, the switching operation becomes equivalent to the gradual increase of the on-duty ratios D1 and D2, and the inrush current at the start of discharge Can be suppressed, and device failure can be prevented.

  Moreover, the rush current at the time of discharge can also be suppressed by gradually decreasing the switching frequency fsw. In the first embodiment, when the inductance ratio Ln (= Lm / Lr) is sufficiently large, it is assumed from FIG. 4 that the gain G of the resonant converter circuit 200 is ideally not changed by the switching frequency fsw. However, in practice, the gain G tends to decrease as the normalized frequency fn becomes larger than 1. Therefore, the gain G can be gradually increased by gradually decreasing from a sufficiently high switching frequency fsw, and thus discharge can be started while preventing an inrush current to the output side.

  In the first embodiment, the secondary side of the transformer 10 of the resonant converter circuit 200 is rectified by a diode. However, the configuration of the synchronous rectification system in which the rectifier diodes 11 and 12 are replaced with semiconductor switching elements is shown. At the time of discharging, by turning off all the secondary-side semiconductor switching elements that perform synchronous rectification, the residual charge of the smoothing capacitor 5 can be reduced while preventing the backflow of power from the voltage source load 52 to the smoothing capacitor 5. Can be discharged.

  In the case of a synchronous rectification configuration in which the rectifier diodes 11 and 12 of the resonant converter circuit 200 are replaced with semiconductor switching elements, if the voltage source load 52 is not connected (load open or a load other than the voltage source). For example, during discharge, the secondary-side semiconductor switching element that performs synchronous rectification in synchronization with the semiconductor switching elements 6 and 7 is turned on / off, thereby causing the load-side power to flow backward to the smoothing capacitor 5, and the output capacitor 13. It is possible to simultaneously discharge the load-side capacitance component including the residual charge of the smoothing capacitor 5. In addition, when the inrush current prevention period is provided by gradually increasing the on-duty ratios D1, D2 of the semiconductor switching elements 6, 7, the synchronous rectifying element is operated in synchronization with the semiconductor switching elements 6, 7. Inrush current from the load side to the smoothing capacitor 5 can also be suppressed.

  Note that the step-down converter circuit 100 is not limited to the configuration shown in FIG. 1. If the DC power supply 51 and the smoothing capacitor 5 are separated from each other by a semiconductor switching element such as a step-up or step-down converter, the residual charge is similarly reduced. Discharge is possible.

  In the first embodiment, the configuration including the step-down converter circuit 100 has been described. However, a configuration in which a converter is not provided in the previous stage of the resonant converter circuit 200 may be used. The residual charge of the smoothing capacitor 5 can be discharged by the off control.

In the first embodiment, the configuration including the voltage source load 52 such as a battery has been described. However, the present invention is not limited to this, and when other loads are connected or the loads are open, Similarly, the residual charge of the smoothing capacitor 5 can be discharged.

  In the first embodiment, the case where the switching frequency fsw of the semiconductor switching elements 6 and 7 is fixed at the time of output control and discharge is described. However, the present invention is not limited to this, and the switching frequency fsw is controlled to be variable. It may be a thing.

  In the first embodiment, the resonant converter circuit 200 has a half-bridge configuration using two semiconductor switching elements on the primary side of the transformer 10. However, it is not limited to this, For example, a full bridge structure may be sufficient.

  In the first embodiment, the secondary rectifier circuit of the transformer 10 has a center tap configuration using two semiconductor switching elements. However, it is not limited to this, For example, a full bridge structure may be sufficient.

Embodiment 2. FIG.
FIG. 9 is a diagram showing a time transition waveform of the voltage and the control parameter at the time of residual charge discharge of the smoothing capacitor in the second embodiment. In the first embodiment, only the smoothing capacitor 5 is discharged, but in the second embodiment, the residual charge of the input capacitor 1 of the step-down converter circuit 100 can also be discharged. Since the apparatus configuration and the output control method are the same as those in the first embodiment, description thereof will be omitted.

<Discharge method of residual charge>
Hereinafter, a method for discharging the residual charge of input capacitor 1 of power conversion apparatus 1000 according to the present embodiment will be described.
The on / off control method of the semiconductor switching elements 6 and 7 for discharging the residual charge of the smoothing capacitor 5 is the same as that shown in FIG. The sequence of the discharge process is shown in FIG. In FIG. 9, the voltage Vc of the input capacitor 1 and the DC voltage Vdc when the residual charge is discharged when the output control is stopped, the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7, the on-duty ratio D3 of the semiconductor switching element 2, The time transition waveform of the dead time Td and switching frequency fsw of the semiconductor switching elements 6 and 7 is shown. Note that it is assumed that the DC power supply 51 is disconnected by a relay or the like at the time of discharging according to the present embodiment.

  9 indicates the discharge period of the input capacitor 1 and the smoothing capacitor 5 in the present embodiment, and the control unit 20 controls the smoothing capacitor 5 from the input capacitor 1 to the time t20 to t21. The inrush current is suppressed from the resonant converter circuit 200 to the load side from time t21 to t22, and the residual charge of the smoothing capacitor 5 by the resonance converter circuit 200 is from time t22 to t23. The discharge process is performed.

  During the period from time t20 to t21, the control unit 20 fixes the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 to 0% and fixes them off, and the on-duty ratio D3 of the semiconductor switching elements 2 from 0% to 100%. Increase gradually. By discharging the residual charge after turning on the semiconductor switching element 2, it becomes possible to discharge the residual charge of the input capacitor 1 and the smoothing capacitor 5 at the same time. However, a potential difference occurs between the input capacitor 1 and the smoothing capacitor 5. When the semiconductor switching element 2 is switched from the off state to the fixed on state, an inrush current is generated. Therefore, the on-duty ratio D3 of the semiconductor switching element 2 is gradually increased to suppress the inrush current.

  Except for the fact that the semiconductor switching element 2 is on, the time t21 to t22 and the time t22 to t23 are the same as the time t10 to t11 and t11 to t12 in FIG. The description in is omitted.

  As described above, according to the power conversion device according to the second embodiment, the configuration including the step-down converter circuit 100 in the previous stage of the resonant converter circuit 200 has the effects of the first embodiment and the semiconductor switching element 2. By turning on, the residual charges of the input capacitor 1 and the smoothing capacitor 5 can be discharged simultaneously.

  In addition, when the residual charge is discharged, the on-duty ratio D3 of the semiconductor switching element 2 is gradually increased to provide a period for suppressing the inrush current generated between the input capacitor 1 and the smoothing capacitor 5, so that the semiconductor switching element and the capacitor due to the inrush current are provided. It is possible to prevent the failure by suppressing the stress on the.

  In the second embodiment, the method of discharging the residual charge of the input capacitor 1 and the smoothing capacitor 5 in the configuration including the step-down converter circuit 100 in the previous stage of the resonant converter circuit 200 has been described. In the case where the boost converter circuit is provided, the residual charges of the input capacitor 1 and the smoothing capacitor 5 can be similarly discharged. The boost converter circuit assumes a system in which the DC power source 51 and the smoothing capacitor 5 are connected in series via a reactor and a diode, and the DC power source 51 is short-circuited via the reactor when the semiconductor switching element is turned on. The input capacitor 1 and the smoothing capacitor 5 can be discharged simultaneously by turning off the semiconductor switching element.

  In the second embodiment, in FIG. 9, the on-duty ratio D3 of the semiconductor switching element 2 is gradually increased and the on-duty ratios D1 and D2 of the semiconductor switching elements 6 and 7 are gradually increased. However, the present invention is not limited to this, and the periods may overlap.

  Further, only the discharge of the residual charge of the input capacitor 1 has been described, but the discharge of the capacitive component held by another device connected to the input side of the power converter 1000 can be similarly discharged.

  Note that the present invention is not limited to the above-described embodiment, and the embodiment can be appropriately modified and omitted within the scope of the invention.

  Moreover, in the figure, the same code | symbol shows the same or an equivalent part.

    1 Input Capacitor, 2 Semiconductor Switching Element, 3 Diode, 4 Step-down Reactor, 5 Smoothing Capacitor, 20 Control Unit, 6, 7 Semiconductor Switching Element, 8 Resonance Capacitor, 9 Resonance Reactor, 10 Transformer, 11, 12 Rectifier Diode, 13 Output Capacitor, 21 Input voltage measurement circuit, 22 Smoothing capacitor voltage measurement circuit, 23 Output voltage measurement circuit, 31, 32, 33 Control line, 41, 42, 43 Signal line, 51 DC power supply, 52 Voltage source load, 100 Step-down converter circuit 200 Resonant converter circuit, 1000 Power converter

Claims (21)

An isolated resonant converter that converts a DC input voltage into a DC output voltage of a different voltage;
A capacitor connected to the input side of the isolated resonant converter;
A control unit for controlling the operation of the isolated resonant converter,
The isolated resonant converter is
A first switching element and a second switching element arranged in parallel with the capacitor and connected in series with each other;
A transformer having a primary winding and a secondary winding;
A resonant capacitor and a resonant reactor connected in series with the second switching element and the primary winding of the transformer, and forming a closed circuit;
A rectifier circuit connected to the secondary side of the transformer,
The controller is
The first switching element and the second switching element are alternately turned on / off at an on-duty ratio or a switching frequency with a dead time therebetween, thereby controlling the DC output voltage,
When the output of the DC output voltage is stopped, the charge of the capacitor is discharged by alternately performing on / off control with a dead time, an on-duty ratio or a switching frequency different from that at the time of controlling the DC output voltage. Power converter.   The power conversion device according to claim 1, wherein the dead time when the output of the DC output voltage is stopped is shorter than the dead time when the DC output voltage is controlled.   The power converter according to claim 1, wherein the dead time when the output of the DC output voltage is stopped is longer than the dead time when the DC output voltage is controlled.   The on-duty ratios of the first switching element and the second switching element when the output of the DC output voltage is stopped are the first switching element and the second switching element when controlling the DC output voltage. 4. The power conversion device according to claim 1, wherein the power conversion device is higher than the on-duty ratio.   The on-duty ratios of the first switching element and the second switching element when the output of the DC output voltage is stopped are the first switching element and the second switching element when the DC output voltage is controlled. 4. The power conversion device according to claim 1, wherein the power conversion device is lower than the on-duty ratio.   6. The power converter according to claim 1, wherein the on-duty ratio of the first switching element is higher than the on-duty ratio of the second switching element.   At the switching frequency higher than a frequency having a period which is twice the time obtained by adding the dead time to a half period of a series resonance frequency of a series resonance circuit of the resonance capacitor and the resonance reactor The power conversion device according to any one of claims 1 to 6, wherein a switching element and the second switching element are on / off controlled. At the switching frequency lower than a frequency having a period which is twice the time obtained by adding the dead time to a half period of a series resonance frequency of a series resonance circuit constituted by the resonance capacitor and the resonance reactor, The power conversion device according to any one of claims 1 to 6, wherein a switching element and the second switching element are on / off controlled.   9. The power converter according to claim 1, wherein when the output of the DC output voltage is stopped, the switching frequency is set to a fixed value. 10.   The inrush current suppression period for suppressing an inrush current to the output side of the isolated resonant converter when the output of the DC output voltage is stopped is provided. Power converter.   The power converter according to claim 10, wherein the on-duty ratio of the first switching element and the second switching element is gradually increased in the inrush current suppression period.   The power conversion device according to claim 10, wherein the dead time is gradually decreased in the inrush current suppression period.   The power converter according to claim 10, wherein the switching frequency is gradually decreased in the inrush current suppression period. The rectifier circuit is a synchronous rectifier circuit configured by a rectifier switching element,
At the time of controlling the DC output voltage, the rectifying switching element is ON / OFF controlled,
The power converter according to any one of claims 1 to 13, wherein the rectifying switching element is not controlled to be turned on / off when the output of the DC output voltage is stopped. The rectifier circuit is a synchronous rectifier circuit configured by a rectifier switching element,
The rectifying switching element is on / off controlled when the DC output voltage is controlled, and the rectifying switching element is on / off controlled when the output of the DC output voltage is stopped. 14. The power conversion device according to any one of items 13.   The power converter according to any one of claims 1 to 15, further comprising a DC converter that outputs the DC input voltage to the capacitor.   The power converter according to claim 16, wherein the DC converter is a step-up type.   The power converter according to claim 16, wherein the DC converter is a step-down type provided with an adjustment switching element capable of adjusting the supply of power from the input side.   The power conversion device according to claim 18, wherein when the output of the DC output voltage is stopped, the adjustment switching element is turned off.   The power conversion device according to claim 18, wherein when the output of the DC output voltage is stopped, the adjustment switching element is turned on.   21. The power conversion device according to claim 20, wherein when the output of the DC output voltage is stopped, an on-duty ratio of the adjustment switching element is gradually increased.

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