JP5739836B2 - Switching power supply - Google Patents

Description

  The present invention relates to a full-bridge type switching power supply device.

  Conventionally, there is a full-bridge type switching power supply device that stabilizes an output voltage by phase shift control, as disclosed in the prior art of Patent Document 1. This type of switching power supply device is suitable for a device that outputs a relatively large power, for example, exceeding 1 kW, and is advantageous in terms of improving the efficiency of the power supply circuit and fixing the switching frequency.

  The conventional switching power supply 10 shown in FIG. 13 is also a full bridge type switching power supply that stabilizes the output voltage by phase shift control, and operates in the same manner as the switching power supply described above during normal operation. Here, in order to further improve the efficiency, a rectifying element for synchronous rectification is used in the rectification unit of the output side rectification smoothing circuit. Hereinafter, the configuration of the switching power supply device 10 and the operation during normal operation will be described in detail.

  The switching power supply 10 includes a first arm 16 that is a series circuit of a first main switching element 12 on the high side and a second main switching element 14 on the low side, a third main switching element 18 on the high side, and a low side. And a second arm 22 that is a series circuit of the fourth main switching element 20. Both ends of the first and second arms 16 and 22 are connected to a pair of input terminals 24a and 24b, and the input voltage Vi of the DC input power supply 26 is applied thereto. As the first to fourth main switching elements 12, 14, 18, and 20, general N-channel MOS type FETs are selected.

  The main transformer 28 includes a primary winding 30 and secondary windings 32 a and 32 b connected in series. The primary winding 30 is between the midpoint of the first arm 16 and the midpoint of the second arm 22. It is connected to the. When each main switching element is turned on and off, an AC voltage obtained by intermittently connecting the input voltage Vi is input to the primary winding 30, and is transformed and output from the secondary windings 32a and 32b. The dots attached to each winding represent the polarity.

  A resonant inductor 34 that contributes to a soft switching operation described later is provided between the midpoint of the first arm 16 and the midpoint of the second arm 22 and in a position in series with the primary winding 30.

  The secondary windings 32a and 32b of the main transformer 28 are rectified and smoothed by a rectifying unit 36 for rectifying the AC voltage generated in the secondary windings 32a and 32b and a smoothing unit 38 for smoothing the rectified voltage. A circuit 40 is connected. The rectifying unit 36 includes first and second rectifying elements 42a and 42b, which are N-channel MOS type FETs, and a synchronous rectification driving circuit 44 that drives the rectifying elements 42a and 42b. The first rectifying element 42a has a drain terminal connected to one end of the secondary winding 32a on the dot side, and the second rectifying element 42b has a drain terminal connected to one end on the opposite side of the dot of the secondary winding 32b, The source terminals of the rectifying elements 42a and 42b are connected to each other. The rectifying unit 36 outputs a rectified voltage between the midpoint of the secondary windings 32a and 32b and the source terminals of the rectifying elements 42a and 42b.

  The synchronous rectification drive circuit 44 detects the voltage of the secondary windings 32a and 32b or a voltage corresponding thereto (for example, the voltage of another winding not shown), and turns on or off the rectifier elements 42a and 42b. Specifically, when “dot-side positive voltage” is generated at both ends of the secondary windings 32a and 32b, the first rectifier element 42a is turned off, the second rectifier element 42b is turned on, and “dot and The first rectifying element 42a is turned on and the second rectifying element 42b is turned off when a "positive voltage is generated on the opposite side". When no voltage is generated, the first rectifying element 42a and the second rectifying element 42 are turned on. Both 42b are turned on.

  The smoothing unit 38 is a low-pass filter that smoothes the rectified voltage output between the midpoint of the secondary windings 32a and 32b and the source terminals of the rectifying elements 42a and 42b. It consists of Both ends of the smoothing capacitor 48 are connected to a pair of output terminals 50a and 50b.

  As described above, the rectifying / smoothing circuit 40 is a so-called center tap type rectifying / smoothing circuit. The rectifying voltage of the rectifying unit 36 is smoothed by the smoothing unit 38 to generate a DC output voltage Vo, which is output to the load 52. Supply voltage Vo and output current Io.

  On / off of each main switching element 12, 14, 18, 20 is controlled by the switching control circuit 56 based on the output of the output voltage detection circuit 54. The output voltage detection circuit 54 detects the output voltage Vo generated at the output terminals 50a and 50b and outputs an output voltage signal Vo1.

  The switching control circuit 56 is a circuit for turning on and off the main switching elements 12, 14, 18, and 20 so that the output voltage signal Vo1 approaches the target value Vr. As shown in FIG. A shift modulation circuit 60 and first and second arm drive circuits 62 and 64 are included. The error amplification circuit 58 is an analog inverting amplifier or the like, and amplifies a difference between the output voltage signal Vo1 and the target voltage Vr and outputs an error amplification signal Vo2.

  The phase shift modulation circuit 60 includes a pulse width modulation means 66, a pulse distribution circuit 68, and four delay circuits 70a to 70d. The pulse width modulation means 66 modulates the error amplification signal Vo2 from the error amplification circuit 58 and outputs a modulation pulse Vh. The pulse distribution circuit 68 distributes the modulation pulse Vh for phase shift control, and outputs four modulation pulses Vh1, Vh2, Vh3, and Vh4 through delay circuits 70a to 70d, respectively. The modulation pulses Vh1, Vh2, Vh3, and Vh4 are all pulse voltages having a high-level time ratio fixed at about 50%, and the phases at which the logic is inverted are shifted from each other.

  The first arm drive circuit 62 includes two buffer circuits 72a and 72b and a first drive transformer 78 having an input winding 74 and output windings 76a and 76b. The buffer circuits 72a and 72b have their outputs connected to one end on the dot side of the input winding 74 and one end on the opposite side, respectively, and in accordance with the logic of the modulation pulses Vh1 and Vh2 of the pulse distribution circuit 68, both ends of the input winding 74 are made high. The pulse voltage in the positive or negative direction is generated in the input winding 74 by changing to the level or the low level. The first drive transformer 78 has output windings 76a and 76b connected between the gate and source terminals of the first and second main switching elements 12 and 14, respectively. The dots attached to the windings represent the polarity. When a positive voltage is generated on the dot side of the output windings 76a and 76b, the first main switching element 12 is turned on, and a positive voltage is generated on the opposite side of the dots. The second main switching element 14 is turned on.

  Similar to the first arm drive circuit 62, the second arm drive circuit 64 includes two buffer circuits 80a and 80b, and a second drive transformer 86 having an input winding 82 and output windings 84a and 84b. Yes. The buffer circuits 80a and 80b have their outputs connected to one end on the dot side of the input winding 82 and one end on the opposite side, respectively, and both ends of the input winding 82 are set high according to the logic of the modulation pulses Vh3 and Vh4 of the pulse distribution circuit 68. The pulse voltage in the positive or negative direction is generated in the input winding 82 by changing to the level or the low level. The second drive transformer 86 has output windings 84a and 84b connected between the gate and source terminals of the third and fourth main switching elements 18 and 20, respectively. The dots attached to the windings indicate the polarity. When a positive voltage is generated on the dot side of the output windings 84a and 84b, the third main switching element 18 is turned on, and a positive voltage is generated on the opposite side of the dots. The fourth main switching element 20 is turned on.

  As described above, the first arm drive circuit 62 applies the first and second drive pulses Vg12 and Vg14 between the gate and source terminals of the first and second main switching elements 12 and 14 based on the modulation pulses Vh1 and Vh2. Output each. The second arm drive circuit 64 sends the third and fourth drive pulses Vg18 and Vg20 between the gate and source terminals of the third and fourth main switching elements 18 and 20 based on the modulation pulses Vh3 and Vh4, respectively. Output.

  Next, the operation | movement at the time of normal driving | operation of the switching power supply device 10 is demonstrated based on FIGS. Here, the length (period tsw) of one cycle of switching of the switching power supply device 10 is constant. As shown in the time chart of FIG. 15, the operation of one cycle can be described by being divided into periods T51 to T58. Each main switching element 12, 14, 18, 20 and each rectifying element 42a, The state of 42b and the path through which the main current flows are expressed as the equivalent circuits of FIGS. 16 and 17, an N-channel MOS FET is connected to a drain-source terminal switch (12a, 14a, 18a, 20a), a drain-source terminal parasitic capacitor (12b, 14b, 18b, 20b), And parasitic diodes (12c, 14c, 18c, 20c) in the direction of flowing current from the source terminal to the drain terminal.

  The operation of the switching power supply 10 shown in FIGS. 15 to 17 is substantially the same as that of the switching power supply disclosed in Patent Document 1 as the prior art. The switching power supply device 10 is different from the switching power supply device of Patent Document 1 in which the rectifying elements 42a and 42b of the rectifying and smoothing circuit 40 are MOS type FETs and using a diode, but when any rectifying element is used, The on / off timing during normal operation and the direction of the flowing current are the same, and there is no significant difference in the time chart during normal operation.

  As can be seen from FIGS. 15 to 17, the switching power supply 10 applies the voltage Vi across the primary winding 30 of the main transformer 28 when the first and fourth main switching elements 12 and 20 are turned on during the period T51. Similarly, when the second and third main switching elements 14 and 18 are turned on in the period T55, the reverse voltage Vi is applied to both ends of the primary winding 30. Here, since the length tsw of one cycle is constant, the output voltage Vo is approximately proportional to the time during which the voltage Vi is applied to both ends of the primary winding 30, that is, the lengths of the periods T51 and T55. For example, when the error amplification circuit 58 determines that “the output voltage signal Vo1 is lower (higher) than the target value Vr”, the modulation pulses Vh1 to Vh4 output from the phase shift modulation circuit 60 change, and each drive pulse is accordingly accompanied. The phase relationship between Vg12 to Vg20 changes. As a result, the periods T51 and T55 become longer (shorter), and the output voltage signal Vo1 rises (decreases) to become equal to the target value Vr.

  In addition, due to the resonance operation of the resonance inductor 34 in the periods T52, T54, T56, and T58, the main switching elements 12, 14, 18, and 20 have the voltage between the drain and source terminals at the time of turning off from Vi to substantially zero. It turns on at the timing. Therefore, by this soft switching operation, the switching loss of each of the main switching elements 12, 14, 18, 20 is reduced, and the occurrence of switching noise is also suppressed.

JP 2010-207014 A

  Since both the switching power supply device disclosed in Patent Document 1 as the prior art and the conventional switching power supply device 10 perform phase shift control, there is a problem that the switching control circuit becomes complicated. For example, in the switching control circuit 56 of the switching power supply device 10, as described above, the pulse distribution circuit 68 distributes the modulation pulse Vh for phase shift control, and the modulation pulses Vh1, Vh2, Vh3, and so on through the delay circuits 70a to 70d, respectively. It has a function to output Vh4. In order to realize the function of “generating four pulse voltages with fixed high-level duty ratio fixed to about 50% and changing the phase of logic inversion,” which is indispensable for phase shift control, a large number of gates Circuits, switch circuits, flip-flops, delay circuits, etc. must be combined in a complex manner. In the case of the switching power supply device of Patent Document 1, the internal configuration of the switching control circuit (switching control unit) is not described in detail, but it is considered that there is a similar problem.

  In the case of the switching power supply device 10, synchronous rectification MOS type FETs are used as the first and second rectifying elements 42 a and 42 b, and there is an advantage that the loss of the rectifying unit 36 is small. There is a problem that it is not suitable for applications in which 10 outputs are connected to each other to perform parallel operation.

  When two identical switching power supply devices 10 are prepared, the output voltage set value Vo (output voltage Vo1 when the output voltage Vo1 matches the target value Vr) slightly differs due to individual differences in the characteristics of the internal elements. When the two outputs are connected to each other and operated in parallel, the switching power supply 10 having the lower output voltage set value Vor has a slightly lower output voltage set value Vor between its own output terminals 50a and 50b. A high voltage Voh is externally applied. Then, the current flows backward through the rectifying elements 42a and 42b, which are MOS FETs capable of conducting in both directions, the smoothing inductor 46 is demagnetized, and eventually an excessive current flows due to magnetic saturation, and the rectifying elements 42a and 42b and the like flow. There is a possibility of destruction.

  Hereinafter, the backflow operation that occurs when the external power supply 88 is connected to the output terminals 50a and 50b of the switching power supply apparatus 10 and a high voltage Voh is applied will be described in detail. Here, the switching power supply 10 has a minimum time so that the time during which the voltage Vi is applied to both ends of the primary winding 30 (periods T51 and T55 in the normal operation in FIG. 15) does not become zero. It is assumed that the circuit 56 performs phase shift control in a range exceeding the minimum time.

  The operation of one period (period tsw) of switching of the switching power supply apparatus 10 can be described by being divided into periods T61 to T68 as shown in the time chart of FIG. 18, and the main switching elements 12 and 14 in each period. , 18, 20 and the states of the rectifying elements 42a and 42b and the path through which the main current flows are expressed as in the equivalent circuits of FIGS.

  First, in the period T61, the drive pulses Vg12, Vg14, Vg18, Vg20 are at a high level, a low level, a low level, and a high level, the first and fourth switches 12a, 20a are turned on, and the second and third switches 14a are turned on. , 18a are off. Therefore, “a positive voltage Vi on the opposite side of the dot” is applied to both ends of the primary winding 30 of the main transformer 28, and “a positive voltage on the opposite side of the dot” is generated on both ends of the secondary windings 32a and 32b. The rectifying element 42a is turned on and the rectifying element 42b is turned off.

  The smoothing inductor 46 has accumulated the excitation energy before the start of the period T61, and starts the operation of releasing the excitation energy when the period T61 starts. As shown in FIG. 19A, the current that the smoothing inductor 46 releases the excitation energy (energy emission current) is the path of the smoothing inductor 46, the secondary winding 32a, the rectifying element 42a, the external power supply 88, and the smoothing inductor 46. And further flows through the main transformer 28 to the path of the primary winding 30, the resonant inductor 34, the first switch 12a, the DC input power supply 26, the fourth switch 20a, and the primary winding 30, and the excitation energy of the smoothing inductor 46 Is regenerated toward the DC input power supply 26. Accordingly, as shown in FIG. 18, the current I46 of the smoothing inductor 46 has a waveform that linearly decreases from a negative value toward zero. At this time, excitation energy is accumulated in the resonant inductor 34 by the energy emission current flowing through the resonant inductor 34.

  The switching control circuit 56 receives the output voltage signal Vo1 output from the output voltage detection circuit 54, that is, the output voltage signal Vo1 corresponding to the voltage Voh between the output terminals 50a and 50b, and “the output voltage Vo1 is lower than the target value Vr. In order to reduce the output voltage signal Vo1, the phase shift control is performed to shorten the length of the period T61 (that is, the time during which the voltage Vi is applied to the primary winding 30) to the specified minimum time. With this control, the fourth drive pulse Vg20 is inverted to the low level, and the period T61 ends.

  In the period T62, the fourth drive pulse Vg20 becomes low level, so the fourth switch 20a is turned off. The other drive pulses Vg12, Vg14, and Vg18 continue to be high level, low level, and low level, and the first, second, and third switches 12a, 14a, and 18a continue to be on, off, and off. When the fourth switch 20a turns off, the energy emission current path from the smoothing inductor 46 is switched from the fourth switch 20a to the fourth parasitic diode 20c as shown in FIG. There is no change in operation, and the same state as in the period T61 is continued, and as shown in FIG. 18, the current I46 of the smoothing inductor decreases linearly toward zero. Thereafter, the third drive pulse Vg18 is inverted to a high level by phase shift control that shortens a period T65 described later to the minimum time, and the period T62 ends.

  When the period T63 is entered, the third drive pulse Vg18 becomes high level, so the third switch 18a is turned on. The other drive pulses Vg12, Vg14, Vg20 continue to be at a high level, a low level, and a low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be on, off, and off. Therefore, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the first and third switches 12a and 18a, and no voltage is generated at both ends of the primary winding 30. Then, since no voltage is generated in the secondary windings 32a and 32b, not only the rectifying element 42a but also the rectifying element 42b is turned on.

  Since the smoothing inductor 46 cannot flow the energy emission current toward the DC input power supply 26, the operation of the external power supply 88 storing the excitation energy in the smoothing inductor 46 starts when the period T63 starts. As shown in FIG. 19C, the current that the external power supply 88 stores the excitation energy in the smoothing inductor 46 (energy storage current) is the external power supply 88, the smoothing inductor 46, the secondary windings 32a and 32b, and the rectifying element 42a. , 42b and the external power supply 88. Therefore, as shown in FIG. 18, the current I46 of the smoothing inductor 46 has a waveform that increases linearly in the negative direction. At this time, since the resonance inductor 34 performs an operation of releasing the excitation energy accumulated before the start of the period T63, the resonance inductor 34, the first switch 12a, the third switch 18a, the primary winding 30, the resonance inductor 34 The energy emission current flows through the path. Thereafter, the first drive pulse Vg12 is inverted to a low level, and the period T63 ends.

  When the period T64 starts, the first drive pulse Vg12 goes to a low level, so the first switch 12a is turned off. The other drive pulses Vg14, Vg18, Vg20 continue to be at the low level, the high level, and the low level, and the second, third, and fourth switches 14a, 18a, 20a continue to be off, on, and off. When the first switch 12a is turned off, the energy emission current path from the resonant inductor 34 is switched from the first switch 12a to the first parasitic diode 12c as shown in FIG. There is no change in operation, and the same state as in the period T63 is continued. Further, since the rectifiers 42a and 42b are kept on as in the period T63, the current I46 of the smoothing inductor 46 increases linearly in the negative direction as shown in FIG. Thereafter, the second drive pulse Vg14 is inverted to a high level at the midpoint of the cycle tsw, and the period T64 ends.

  In the period T65, the second drive pulse Vg14 becomes high level, so that the switch 14a is turned on. The other drive pulses Vg12, Vg18, Vg20 continue to be at low level, high level, and low level, and the first, third, and fourth switches 12a, 18a, 20a continue to be off, on, and off. Accordingly, “dot-side positive voltage Vi” is applied to both ends of the primary winding 30 of the main transformer 28, and “dot-side positive voltage” is generated at both ends of the secondary windings 32a and 32b. Is turned off, and the rectifying element 42b is turned on.

  The smoothing inductor 46 has accumulated excitation energy before the start of the period T65, and starts the operation of releasing the excitation energy when the period transitions to the period T65. The energy emission current of the smoothing inductor 46 flows through the path of the smoothing inductor 46, the secondary winding 32b, the rectifying element 42b, the external power supply 88, and the smoothing inductor 46 as shown in FIG. Through the primary winding 30, the third switch 18 a, the DC input power supply 26, the second switch 14 a, the resonant inductor 34, and the primary winding 30. The excitation energy of the smoothing inductor 46 is directed toward the DC input power supply 26. It is regenerated. Accordingly, as shown in FIG. 18, the current I46 of the smoothing inductor 46 has a waveform that linearly decreases from a negative value toward zero. At this time, excitation energy is accumulated in the resonant inductor 34 by the energy emission current flowing through the resonant inductor 34.

  As described in the period T61, the switching control circuit 56 sets the length of the period T65 (that is, the time during which the voltage Vi is applied to the primary winding 30) to the specified minimum in order to reduce the output voltage signal Vo1. Perform phase shift control to shorten the time. By this control, the third drive pulse Vg18 is inverted to the low level, and the period T65 ends.

  In the period T66, the third drive pulse Vg18 becomes low level, so the third switch 18a is turned off. The other drive pulses Vg12, Vg14, Vg20 continue to be at the low level, the high level, and the low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be off, on, and off. When the third switch 18a turns off, the energy emission current path of the smoothing inductor 46 is switched from the third switch 18a to the third parasitic diode 18c as shown in FIG. There is no change above, and the same state as in the period T65 is continued, and as shown in FIG. 18, the current I46 of the smoothing inductor decreases linearly toward zero. After that, the fourth drive pulse Vg20 is inverted to a high level by phase shift control that shortens the period T61 to the minimum time, and the period T66 ends.

  In the period T67, the fourth drive pulse Vg20 becomes high level, so the fourth switch 20a is turned on. The other drive pulses Vg12, Vg14, and Vg18 continue to be at low level, high level, and low level, and the first, second, and third switches 12a, 14a, and 18a continue to be off, on, and off. Therefore, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the second and fourth switches 14a and 20a, and no voltage is generated at both ends of the primary winding 30. Then, no voltage is generated in the secondary windings 32a and 32b, so that not only the rectifying element 42b but also the rectifying element 42a is turned on.

  Since the smoothing inductor 46 cannot flow the energy emission current toward the DC input power supply 26, the operation of the external power supply 88 storing the excitation energy in the smoothing inductor 46 starts when the period T67 starts. The energy storage current from the external power supply 88 flows through the path of the external power supply 88, the smoothing inductor 46, the secondary windings 32a and 32b, the rectifier elements 42a and 42b, and the external power supply 88, as shown in FIG. Therefore, as shown in FIG. 18, the current I46 of the smoothing inductor 46 has a waveform that increases linearly in the negative direction. At this time, the resonance inductor 34 performs an operation of releasing the excitation energy accumulated before the start of the period T67. Therefore, the resonance inductor 34, the primary winding 30, the fourth switch 20a, the second switch 14a, and the resonance inductor 34. The energy emission current flows through the path. Thereafter, the second drive pulse Vg14 is inverted to the low level, and the period T67 ends.

  In the period T68, the second drive pulse Vg14 becomes low level, so the second switch 14a is turned off. The other drive pulses Vg12, Vg18, and Vg20 continue to be at the low level, the low level, and the high level, and the first, third, and fourth switches 14a, 18a, and 20a continue to be off, off, and on. When the second switch 12a turns off, the energy emission current path of the resonant inductor 34 is switched from the second switch 14a to the second parasitic diode 14c as shown in FIG. There is no change above and the state similar to the period T67 is continued. Further, since the rectifying elements 42a and 42b are kept on as in the period T67, the current I46 of the smoothing inductor linearly increases in the negative direction as shown in FIG. Thereafter, the second drive pulse Vg14 is inverted to a high level at the end of the cycle tsw, and the period T68 ends. When the period T68 ends, the process returns to the period T61, and the operations in the above-described periods T61 to T68 are repeated.

  Here, the current I46 of the smoothing inductor 46 becomes a problem. In the first half period T61 to T64 of the period tsw, the excitation energy is released from the smoothing inductor 46 in the periods T61 and T62. In particular, the period T61 is controlled to the specified minimum time and becomes very short. At the end of T62, the energy emission current does not become zero, and excitation energy remains. Then, the period T63 starts with the excitation energy remaining, and the excitation energy is accumulated cumulatively during the periods T63 and T64. As a result, the negative direction value of the current I46 of the smoothing inductor 46 becomes larger at the end of the period T64 than at the start of the period T61, and the magnetization of the smoothing inductor 46 proceeds. Similarly, in the period T65 to T68 in the latter half of the period tsw, the negative direction value of the current I46 of the smoothing inductor 46 is larger at the end of the period T68 than at the start of the period T65, and the smoothing inductor 46 is biased. The magnetism further proceeds. Accordingly, the smoothing inductor 46 becomes magnetically saturated while repeating the operations in the periods T61 to T68, and a large reverse current easily flows from the external power supply 88 in the direction of the smoothing inductor 46, and circuit elements such as the rectifying elements 42a and 42b break down. The possibility increases.

  In the case of the switching power supply device disclosed as the prior art of Patent Document 1, since the rectifying element of the rectifying / smoothing circuit is a diode, current is prevented from flowing from the smoothing inductor 46 to the secondary windings 32a and 32b. Therefore, even when an external voltage is applied, the operations shown in FIGS. 19 and 20 do not occur.

  The present invention has been made in view of the above-described background art, and provides a full-bridge type switching power supply device in which the switching control circuit can be simply configured and the main switching element can be subjected to the soot switching. Objective.

The present invention comprises a series circuit of a first main switching element on the high side and a second main switching element on the low side, a first arm having a DC input power supply connected to both ends thereof, and a third circuit on the high side. A second arm composed of a series circuit of a main switching element and a fourth main switching element on the low side, and a DC input power supply connected to both ends thereof;
A primary transformer having a primary winding and a secondary winding, wherein the primary winding is provided between a midpoint of the first arm and a midpoint of the second arm; and a midpoint of the first arm And a resonant inductor provided in a position in series with the primary winding between the second arm and the middle point of the second arm;
A rectifying and smoothing circuit connected to the secondary winding and configured to rectify the AC voltage generated in the secondary winding, and a smoothing unit that smoothes the rectified voltage and outputs a DC output voltage;
An output voltage detection circuit that detects an output voltage or a voltage corresponding to the output voltage and outputs an output voltage signal, and outputs first to fourth drive pulses to each of the main switching elements, and the output voltage signal is a target value. A switching control circuit for turning on and off each of the main switching elements so as to approach
The switching control circuit includes a first period in which an input voltage is applied to the primary winding in a positive direction, a second period in which no voltage is applied, a third period in which an input voltage is applied in a negative direction, and a voltage is applied again. The first to fourth drive pulses, wherein the fourth period that is not performed is one cycle, and the total time of the first and second periods is equal to the total time of the third and fourth periods, the first main switching element oN, the second main switching element off, the third main switching element off, by turning on the fourth main switching element, enters into the second period, the fourth main Reversing the switching element off, then reversing the third main switching element on, then reversing off the first main switching element before the start of the third period, When entering, the second main switching element is inverted to ON, and when entering the fourth period, the second main switching element is inverted to OFF, and then the first main switching element is inverted to ON, Thereafter, the first to fourth drive pulses for inverting the third main switching element before the first period starts and inverting the fourth main switching element when entering the first period again. Is a switching power supply device that outputs.

  The rectifying unit of the rectifying / smoothing circuit detects a voltage of a synchronous rectifying element and a voltage of the secondary winding of the main transformer or a voltage corresponding thereto to turn on and off the rectifying element, and the secondary winding When a voltage is generated on the wire, the generated voltage is rectified in the positive direction so as to be input to the input terminal of the smoothing unit, and when no voltage is generated in the secondary winding, A synchronous rectification drive circuit is configured so that both ends of the secondary winding are short-circuited to the input end of the smoothing unit. For example, each of the first to fourth main switching elements and the synchronous rectifying element is composed of an N-channel MOS FET.

The switching control circuit distributes an error amplification circuit that compares the output voltage signal with a target voltage and outputs an error amplification signal obtained by amplifying the difference, and a modulation pulse obtained by pulse-modulating the error amplification signal. A pulse width modulation circuit for outputting a first modulation pulse for one arm and a second modulation pulse for a second arm; and generating the first and second drive pulses based on the first modulation pulse, A first arm driving circuit for outputting between the gate and source terminals of the first and second main switching elements; and generating the third and fourth driving pulses based on the second modulation pulse, respectively. A second arm drive circuit that outputs between the gate and source terminals of the main switching element,
The first arm drive circuit receives the first modulation pulse, and after the first modulation pulse is inverted to a high level, the first arm drive circuit outputs the second drive pulse that is inverted to a high level after a predetermined time has elapsed. By observing the potential of the middle point of the delay circuit and the first arm and detecting that the potential has increased, charging between the gate and source terminals of the first main switching element allows the first A first gate charging circuit for inverting the drive pulse to a high level; and discharging the first drive pulse between the gate and source terminals of the first main switching element when the first modulation pulse is at a high level. It consists of a first gate discharge circuit that inverts to low level,
The second arm driving circuit outputs the fourth driving pulse that is input to the second modulation pulse, and after the second modulation pulse is inverted to a high level, and then inverted to a high level after a lapse of a predetermined time. By observing the potential of the middle point of the delay circuit and the second arm and detecting that the potential has risen, charging between the gate and source terminals of the third main switching element allows the third A second gate charging circuit for inverting the drive pulse to a high level, and discharging the third drive pulse between the gate and source terminals of the third main switching element when the second modulation pulse is at a high level. The second gate discharge circuit is inverted to a low level.

The first gate charging circuit includes a first rectifier diode that rectifies a DC voltage for a control circuit and outputs the rectified voltage to a cathode terminal, a cathode terminal of the first rectifier diode, and a source terminal of the first main switching element. A first smoothing capacitor connected between the first rectifier diode and a first transistor which is a PNP transistor whose emitter terminal is connected to the cathode terminal of the first rectifier diode and whose collector terminal is connected to the gate terminal of the first main switching element And a first diode whose anode terminal is connected to the base terminal side of the first transistor and whose cathode terminal is connected to the drain terminal side of the first main switching element, and is inserted in series at the connection point of the first diode And a first resistor that limits a current flowing through the first diode,
The second gate charging circuit rectifies the DC voltage for the control circuit and outputs the rectified voltage to the cathode terminal, and between the cathode terminal of the second rectifier diode and the source terminal of the third main switching element. A second smoothing capacitor connected; a second transistor which is a PNP transistor having an emitter terminal connected to the cathode terminal of the second rectifier diode and a collector terminal connected to the gate terminal of the third main switching element; The terminal is connected to the base terminal side of the second transistor, the cathode terminal is connected to the drain terminal side of the third main switching element, and inserted in series at the connection point of the second diode, It is comprised with the 2nd resistance which restrict | limits the electric current which flows into a 2nd diode.

Further, the first gate discharge circuit is connected between a first drive transformer for transmitting the first modulation pulse and a gate-source terminal of the first main switching element, according to an output of the first drive transformer. A first extraction transistor that is turned on and off, and when the first modulation pulse is at a high level, the first extraction transistor is turned on to discharge between the gate and source terminals of the first main switching element, and The two-gate discharge circuit includes a second drive transformer that transmits the second modulation pulse and a second drive transformer that is connected between the gate and source terminals of the third main switching element and that is turned on / off according to the output of the second drive transformer. An extraction transistor, and when the second modulation pulse is at a high level, the second extraction transistor is turned on and the third extraction pulse is turned on. It may be configured to discharge the gate-source terminals of the switching element. In this case, the first gate discharge circuit is provided with a first narrow pulse generation circuit that outputs a first narrow pulse that becomes a high level for a short time when the input first modulation pulse is inverted to a high level. The first driving transformer transmits timing information of the first narrow pulse, the first modulation pulse being inverted to a high level, and the second gate discharge circuit receives the second modulation input. A second narrow pulse generation circuit is provided that outputs a second narrow pulse that becomes a high level for a short time when the pulse is inverted to a high level, and the second drive transformer is the second narrow pulse, It may be configured to transmit timing information at which the two modulation pulses are inverted to a high level. Further, the resonant inductor may utilize a leakage inductance inside the main transformer.

  In addition, the above-mentioned notation of drain, source, and gate corresponds to transistor elements such as MOS type FETs and IGBTs, but in the case of bipolar transistors, they are replaced with collectors, emitters, and bases, respectively. is there.

  The switching power supply device of the present invention can simplify the configuration of the switching control circuit as compared with the case of phase shift control, and contributes to downsizing, energy saving, and cost reduction of the device. Further, there is an advantage that a general full-bridge control IC that is not used for phase shift control can be used as the pulse width modulation circuit in the switching control circuit. There are few types of control ICs for phase shift control on the market, but in the case of general full-bridge type control ICs, there are multiple types of commercial products with their respective characteristics. A user-friendly one can be selected according to the application. On the other hand, a general full-bridge type control IC does not consider soft switching of the main switching element, but in the case of this switching power supply device, first and second arm drive circuits and a resonant inductor having a simple configuration are provided. As a result, soft switching of the main switching element becomes possible, and generation of switching loss and switching noise can be easily suppressed.

1 is a block diagram showing a first embodiment of a switching power supply device of the present invention. It is a circuit diagram which shows the structure of the switching control circuit of the switching power supply device of 1st embodiment. It is a time chart which shows the operation | movement at the time of normal driving | operation of the switching power supply device of 1st embodiment. 3 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T11 to T14 of the time chart of FIG. . 3 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T15 to T18 of the time chart of FIG. . It is a block diagram which shows 2nd embodiment of the switching power supply device of this invention. It is a circuit diagram which shows the structure of the switching control circuit of the switching power supply device of 2nd embodiment. It is a circuit diagram which shows the structure of the synchronous rectification drive circuit of the switching power supply device of 2nd embodiment. It is a time chart which shows the operation | movement at the time of the external voltage application of the switching power supply device of 2nd embodiment. 9 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T21 to T24 of the time chart of FIG. 9 and the path through which the main current flows. . FIG. 10 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T25 to T28 of the time chart of FIG. . It is a circuit diagram (a) and (b) which shows the 1st and 2nd modification of a rectification smoothing circuit. It is a block diagram which shows the switching power supply device of the conventional phase shift control. It is a circuit diagram which shows the structure of the switching control circuit of the conventional switching power supply device. It is a time chart which shows the operation | movement at the time of normal driving | operation of the conventional switching power supply device. 15 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T51 to T54 of the time chart of FIG. . 15 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element in the period T55 to T58 of the time chart of FIG. . It is a time chart which shows the operation | movement at the time of the external voltage application of the conventional switching power supply device. 18 is an equivalent circuit (a), (b), (c), (d) showing the state of each main switching element and each rectifying element and the path through which the main current flows in the periods T61 to T64 in the time chart of FIG. . 18 is equivalent circuits (a), (b), (c), and (d) showing the states of the main switching elements and the rectifying elements in the period T65 to T68 in the time chart of FIG. .

  Hereinafter, a first embodiment of a switching power supply device of the present invention will be described with reference to FIGS. Here, the same components as those of the conventional switching power supply device 10 are described with the same reference numerals.

  As shown in FIG. 1, the switching power supply 90 according to the first embodiment is the same as the switching power supply 10 in the configuration including the first arm 16, the second arm 22, the main transformer 28, and the resonant inductor 34. As the first to fourth main switching elements 12, 14, 18, and 20, for example, a general N-channel MOS type FET is selected. Further, the resonant inductor 34 may be an independent inductance element whose inductance is sufficiently smaller than that of the primary winding 30 of the main transformer 28, or the leakage inductance in the main transformer 28 may be used to reduce the number of components. Good.

  The secondary windings 32a and 32b of the main transformer 28 include a rectifying / smoothing circuit 94 including a rectifying unit 92 that rectifies an AC voltage generated in the secondary windings 32a and 32b and a smoothing unit 38 similar to the above. Is connected. The rectifying unit 92 includes first and second rectifying elements 92a and 92b which are high-speed switching diodes. The first rectifying element 92a has a cathode terminal connected to one end on the dot side of the secondary winding 32a. The second rectifying element 92b has a cathode terminal connected to one end of the secondary winding 32b opposite to the dot, and anode terminals connected to each other. The rectifier 92 outputs a rectified voltage between the midpoint of the secondary windings 32a and 32b and the anode terminals of the rectifier elements 92a and 92b.

  Thus, the rectifying / smoothing circuit 94 is a so-called center tap type rectifying / smoothing circuit, and each rectifying element 92a, 92b becomes conductive or non-conductive according to the voltage change of the secondary windings 32a, 32b, and the secondary When a voltage is generated in the windings 32a and 32b, the generated voltage is rectified in the positive direction and input to the input terminal of the smoothing unit 38. When no voltage is generated in the secondary windings 32a and 32b, An operation of short-circuiting both ends of the secondary windings 32 a and 32 b to the input end of the smoothing unit 38 is performed. Then, the smoothing unit 38 smoothes the rectified voltage of the rectifying unit 36 and outputs a predetermined output voltage Vo, and supplies the output voltage Vo and the output current Io toward the load 52.

  On / off of each main switching element 12, 14, 18, 20 is controlled by the switching control circuit 96 based on the output of the output voltage detection circuit 54. The output voltage detection circuit 54 detects the output voltage Vo generated at the output terminals 50a and 50b or a voltage corresponding thereto, and outputs an output voltage signal Vo1. “Detecting a voltage corresponding to this” refers to indirectly detecting the output voltage Vo. For example, by providing another winding in the main transformer 28 and rectifying and smoothing the voltage of the other winding, the output voltage Vo is detected. A circuit that detects a voltage substantially proportional to Vo is conceivable.

  The switching control circuit 96 is a circuit for turning on and off the main switching elements 12, 14, 18, and 20 so that the output voltage signal Vo1 approaches the target value Vr. As shown in FIG. A width modulation circuit 98 and first and second arm drive circuits 100 and 102 are included. The error amplifying circuit 58 is, for example, an analog inverting amplifier, and amplifies the difference between the output voltage signal Vo1 and the target voltage Vr and outputs an error amplified signal Vo2.

  The pulse width modulation circuit 98 includes pulse width modulation means 66 and a pulse distribution circuit 104. The pulse width modulation means 66 modulates the error amplification signal Vo2 from the error amplification circuit 58 and outputs a modulation pulse Vh. The pulse distribution circuit 104 distributes the modulation pulse Vh for controlling the first arm 16 and the second arm 22, and outputs two modulation pulses Vh1 and Vh2. The first and second modulation pulses Vh1 and Vh2 are pulse voltages having high time ratios equal to each other, the phase in which the logic is inverted is shifted by 180 degrees, and each time ratio changes according to the error amplification signal Vo2. .

  The first arm driving circuit 100 includes a first delay circuit 106, a first gate charging circuit 108, and a first gate discharging circuit 110. The first delay circuit 106 includes, for example, an integration circuit and a buffer circuit, and a second drive pulse Vg14 obtained by delaying the first modulation pulse Vh1 of the pulse distribution circuit 104 by a time tk1 is used as the gate of the second main switching element 14. • Output between source terminals.

  The first gate charging circuit 108 rectifies the DC voltage of the DC power source 112 for the control circuit and outputs it to the cathode terminal, the cathode terminal of the first rectifier diode 114 and the first main switching element 12. A first smoothing capacitor 116 connected between the source terminal, a first charge switch 118 connected to the cathode terminal of the first rectifier diode 114 and the gate terminal of the first main switching element 12, and a first charge switch 118. And a first switch control circuit 120 for controlling. The first switch control circuit 120 observes the second drive pulse Vg14 of the second main switching element 14, and after the time tk2 has elapsed after the second drive pulse Vg14 is inverted from the high level to the low level, The switch 118 is turned on for a predetermined short time. When the first charging switch 118 is turned on, the gate-source terminal of the first main switching element 12 is charged, and the first main switching element 12 is turned on.

  The first gate discharge circuit 110 includes a first drive transformer 122 having an input winding 122a and an output winding 122b, a first switching transistor 124 connected in series to the input winding 122a, and a gate of the first main switching element 12. The source terminal is connected to the first winding transistor 126 so as to be short-circuited and releasable, and is driven by the voltage of the output winding 122b. A direct current is connected to both ends of the series circuit of the input winding 122a and the first switching transistor 124. A power supply 112 is connected. The dots of the windings 122a and 122b of the first drive transformer 122 represent the polarity.

  The first switching transistor 124 is turned on when the first modulation pulse Vh1 of the pulse distribution circuit 104 is at a high level, thereby generating a pulse voltage having the same phase as the first modulation pulse Vh1 in the input winding 122a. This pulse voltage is transmitted between the gate and source terminals of the first extraction transistor 126 from the output winding 122b through the first drive transformer 122, and the first extraction transistor 126 is turned on when the first modulation pulse Vh1 is at a high level. . When the first extraction transistor 126 is turned on, the gate-source terminal of the first main switching element 12 is discharged, and the first main switching element 12 is turned off. Thus, the first driving pulse Vg12 for driving the first main switching element 12 is generated by the first gate charging circuit 108 and the first gate discharging circuit 110.

  Similar to the first arm driving circuit 100, the second arm driving circuit 102 includes a second delay circuit 128, a second gate charging circuit 130, and a second gate discharging circuit 132. The second delay circuit 128 includes, for example, an integration circuit and a buffer circuit. The second drive pulse Vg20 obtained by delaying the second modulation pulse Vh2 of the pulse distribution circuit 104 by a time tk1 is used as the gate of the fourth main switching element 20. • Output between source terminals.

  The second gate charging circuit 130 rectifies the DC voltage of the DC power source 112 for the control circuit and outputs it to the cathode terminal, the cathode terminal of the second rectifier diode 134 and the third main switching element 18. A second smoothing capacitor 136 connected between the source terminal, a second charge switch 138 connected to the cathode terminal of the second rectifier diode 134 and the gate terminal of the third main switching element 18, and a second charge switch 138. And a second switch control circuit 140 for controlling. The second switch control circuit 140 observes the fourth drive pulse Vg20 of the fourth main switching element 20, and after the time tk2 has elapsed after the fourth drive pulse Vg20 is inverted from the high level to the low level, The switch 138 is turned on for a predetermined short time. When the second charging switch 138 is turned on, the gate-source terminal of the third main switching element 18 is charged, and the third main switching element 18 is turned on.

  The second gate discharge circuit 132 includes a second drive transformer 142 having an input winding 142a and an output winding 142b, a second switching transistor 144 connected in series to the input winding 142a, and a gate of the third main switching element 18. The source terminal is connected so as to be short-circuited and releasable, and is constituted by a second extraction transistor 146 that is driven by the voltage of the output winding 142b, and a direct current is connected to both ends of the series circuit of the input winding 142a and the second switching transistor 144. A power supply 112 is connected. The dots of the windings 142a and 142b of the second drive transformer 142 represent the polarity. The second switching transistor 144 is turned on when the second modulation pulse Vh2 of the pulse distribution circuit 104 is at a high level, thereby generating a pulse voltage having the same phase as the first modulation pulse Vh1 in the input winding 142a. This pulse voltage is transmitted between the gate and source terminals of the second extraction transistor 146 from the output winding 142b through the second drive transformer 142, and the second extraction transistor 146 is turned on when the second modulation pulse Vh2 is at a high level. . When the second extraction transistor 146 is turned on, the gate-source terminal of the third main switching element 18 is discharged, and the third main switching element 18 is turned off. As described above, the second driving pulse Vg18 for driving the third main switching element 18 is generated by the second gate charging circuit 130 and the second gate discharging circuit 132.

  With the above configuration, the switching control circuit 96 outputs the first to fourth drive pulses Vg12, Vg14, Vg18, and Vg20 to turn on and off the first to fourth main switching elements 12, 14, 18, and 20. A first period (period T11 to be described later) in which the input voltage Vi is applied in the positive direction to the primary winding 30 of the main transformer 28, a second period (period T12 to T14 to be described later) in which no voltage is applied, and the input voltage Vi A first period is a third period (period T15 to be described later) and a fourth period (period T16 to T18 to be described later) in which no voltage is applied again is one cycle, and the total time of the first and second periods Switching control is performed to make the total time of the third and fourth periods equal. The switching cycle tsw (the length of one cycle) is constant.

  Next, the operation during normal operation of the switching power supply device 90 will be described with reference to FIGS. The operation of one cycle of the switching power supply device 90 can be described by being divided into periods T11 to T18 as shown in the time chart of FIG. 3, and each of the main switching elements 12, 14, 18, and 20 in each period. The state and the path through which the main current flows are expressed as in the equivalent circuits of FIGS. 4 and 5, an N-channel MOS type FET is connected to a drain-source terminal switch (12a, 14a, 18a, 20a), a drain-source terminal parasitic capacitor (12b, 14b, 18b, 20b), And parasitic diodes (12c, 14c, 18c, 20c) in the direction of flowing current from the source terminal to the drain terminal.

  First, during the period T11, the drive pulses Vg12, Vg14, Vg18, Vg20 are at a high level, a low level, a low level, and a high level, the first and fourth switches 12a, 20a are turned on, and the second and third switches 14a. , 18a are off. Therefore, “a positive voltage Vi on the opposite side of the dot” is applied to both ends of the primary winding 30 of the main transformer 28, and “a positive voltage on the opposite side of the dot” is generated on both ends of the secondary windings 32a and 32b. The rectifying element 92a becomes conductive and the rectifying element 92b becomes non-conductive.

  In this state, as shown in FIG. 4A, a current flows through the path of the DC input power supply 26, the first switch 12a, the resonant inductor 34, the primary winding 30, the fourth switch 20a, and the input DC power supply 26. A current also flows through the main transformer 28 to the secondary winding 32a, the smoothing inductor 46, the smoothing capacitor 48, the load 52, the rectifying element 92a, and the secondary winding 32a. At this time, excitation energy is accumulated in the resonant inductor 34 and the smoothing inductor 46 by the current.

  The switching control circuit 96 receives the output voltage signal Vo1 output from the output voltage detection circuit 54, and the length of the period T11 (that is, the voltage Vi is applied to the primary winding 30 so that the output voltage Vo1 becomes equal to the target value Vr. Determine the length of the first period applied). By this control, the fourth drive pulse Vg20 is inverted to the low level, and the period T11 ends.

  When the period transitions to the period T12, the fourth drive pulse Vg20 goes to a low level, so the fourth switch 20a is turned off. The other drive pulses Vg12, Vg14, and Vg18 continue to be high level, low level, and low level, and the first, second, and third switches 12a, 14a, and 18a continue to be on, off, and off. As a result, no voltage is generated in the primary winding 30 and the secondary windings 32a and 32b.

  When the fourth switch 20a turns off, the resonance inductor 34 starts to emit excitation energy. As shown in FIG. 4B, the current that the resonant inductor 34 releases the excitation energy (energy emission current) is the resonant inductor 34, the primary winding 30, the third parasitic capacitor 18b, the first switch 12a, and the resonant inductor 34. , The charge of the third parasitic capacitor 18b is discharged, and the voltage Vd18 between the drain and source terminals of the third main switching element 18 decreases toward zero. At the same time, a current for charging the fourth parasitic capacitor 20 b flows through the path of the resonant inductor 34, the primary winding 30, the fourth parasitic capacitor 20 b, the DC input power supply 26, the first switch 12 a, and the resonant inductor 34. The voltage Vd20 between the drain and source terminals of the main switching element 20 increases toward the voltage Vi.

  In addition, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 starts an operation of releasing excitation energy. The current (energy emission current) that the smoothing inductor 46 releases the excitation energy flows through the smoothing inductor 46, the smoothing capacitor 48 and the load 52, the rectifying elements 92 a and 92 b, the secondary windings 32 a and 32 b, and the smoothing inductor 46.

  Thereafter, when the time tk2 has elapsed since the fourth drive pulse Vg20 is inverted to the low level, the third drive pulse Vg18 is inverted to the high level, and the period T12 ends. Here, the time tk2 is set to a time required for the voltage Vd18 between the drain and source terminals of the third main switching element 18 to drop from Vi to substantially zero. The rate at which the voltage Vd18 decreases can be approximated by the resonant frequency of the resonant inductor 34 and the corresponding parasitic capacitor.

  In the period T13, the third drive pulse Vg18 goes high, so the third switch 18a is turned on. The other drive pulses Vg12, Vg14, Vg20 continue to be at a high level, a low level, and a low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be on, off, and off. Accordingly, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the first and third switches 12a and 18a, and a voltage is generated in the primary winding 30 and the secondary windings 32a and 32b. do not do.

  When the third switch 18a turns on, the voltage Vd18 has dropped to substantially zero, so that ideal soft switching is performed. When the third switch 18a turns on, as shown in FIG. 4C, the path of the energy emission current of the resonant inductor 34 is the resonant inductor 34, the primary winding 30, the third switch 18a, the first switch 12a, It becomes a path of the resonant inductor 34. In addition, since the voltage Vd20 between the drain and source terminals of the fourth main switching element 20 has reached Vi, the current for charging the fourth parasitic capacitor 20b does not flow.

  Further, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 continues the operation of releasing the excitation energy, and the energy discharge current of the smoothing inductor 46 is changed to the smoothing inductor 46, the smoothing capacitor 48, the load 52, It flows through the path of the rectifying elements 92a and 92b, the secondary windings 32a and 32b, and the smoothing inductor 46.

  Thereafter, the first drive pulse Vg12 is inverted to a low level at a timing before time tk before the intermediate point of the period tsw, and the period T13 ends.

  In the period T14, the first drive pulse Vg12 becomes low level, so the first switch 12a is turned off. The other drive pulses Vg14, Vg18, Vg20 continue to be at the low level, the high level, and the low level, and the second, third, and fourth switches 14a, 18a, 20a continue to be off, on, and off. As a result, no voltage is generated in the primary winding 30 and the secondary windings 32a and 32b.

  When the first switch 12a is turned off, as shown in FIG. 4D, the energy emission current of the resonant inductor 34 is changed to the resonant inductor 34, the primary winding 30, the third switch 18a, the DC input power supply 26, the second The second capacitor 14b flows through the path of the parasitic capacitor 14b and the resonant inductor 34, the electric charge of the second parasitic capacitor 14b is discharged, and the voltage Vd14 between the drain and source terminals of the second main switching element 14 decreases toward zero. At the same time, a current for charging the first parasitic capacitor 12 b flows through the path of the resonant inductor 34, the primary winding 30, the third switch 18 a, the first parasitic capacitor 12 b, and the resonant inductor 34. The voltage Vd12 between the drain and source terminals increases toward the voltage Vi.

  Further, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 continues the operation of releasing the excitation energy, and the energy discharge current of the smoothing inductor 46 is changed to the smoothing inductor 46, the smoothing capacitor 48, the load 52, It flows through the path of the rectifying elements 92a and 92b, the secondary windings 32a and 32b, and the smoothing inductor 46.

  Thereafter, the second drive pulse Vg14 is inverted to a high level at an intermediate point of the constant period tsw, and the period T14 ends. Here, the time tk1 that is the length of the period T14 is set to the time required for the voltage Vd14 between the drain and source terminals of the second main switching element 14 to drop from Vi to substantially zero. The speed at which the voltage Vd14 decreases is the same as described above.

  In the period T15, the second drive pulse Vg14 becomes high level, so the second switch 14a is turned on. The other drive pulses Vg12, Vg18, Vg20 continue to be at low level, high level, and low level, and the first, third, and fourth switches 12a, 18a, 20a continue to be off, on, and off. Accordingly, “dot-side positive voltage Vi” is applied to both ends of the primary winding 30 of the main transformer 28, and “dot-side positive voltage” is generated at both ends of the secondary windings 32a and 32b. Is turned off, and the rectifying element 92b is turned on.

  In this state, as shown in FIG. 5A, a current flows through the path of the DC input power supply 26, the third switch 18a, the primary winding 30, the resonant inductor 34, the second switch 14a, and the input DC power supply 26, A current also flows through the main transformer 28 to the secondary winding 32b, the smoothing inductor 46, the smoothing capacitor 48, the load 52, the rectifying element 92b, and the secondary winding 32b. At this time, excitation energy is accumulated in the resonant inductor 34 and the smoothing inductor 46 by the current.

  The switching control circuit 96 receives the output voltage signal Vo1 output from the output voltage detection circuit 54, and the length of the period T15 (that is, the voltage Vi is applied to the primary winding 30 so that the output voltage Vo1 becomes equal to the target value Vr. Determine the length of the third period to be applied). By this control, the second drive pulse Vg14 is inverted to the low level, and the period T15 ends.

  When the period T16 is entered, the second drive pulse Vg14 becomes low level, so the second switch 14a is turned off. The other drive pulses Vg12, Vg18, Vg20 continue to be at low level, high level, and low level, and the first, third, and fourth switches 12a, 18a, 20a continue to be off, on, and off. As a result, no voltage is generated in the primary winding 30 and the secondary windings 32a and 32b.

  When the second switch 14a is turned off, the resonance inductor 34 starts to emit excitation energy. As shown in FIG. 5B, the energy discharge current of the resonant inductor 34 flows through the path of the resonant inductor 34, the first parasitic capacitor 12b, the third switch 18a, the primary winding 30, and the resonant inductor 34, and the first parasitic current is generated. The electric charge of the capacitor 12b is discharged, and the voltage Vd12 between the drain and source terminals of the first main switching element 12 decreases toward zero. At the same time, a current for charging the second parasitic capacitor 14 b flows through the path of the resonant inductor 34, the second parasitic capacitor 14 b, the DC input power supply 26, the third switch 18 a, the primary winding 30, and the resonant inductor 34. The voltage Vd14 between the drain and source terminals of the main switching element 14 increases toward the voltage Vi.

  In addition, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 starts an operation of releasing excitation energy. The energy emission current of the smoothing inductor 46 flows through the path of the smoothing inductor 46, the smoothing capacitor 48 and the load 52, the rectifying elements 92a and 92b, the secondary windings 32a and 32b, and the smoothing inductor 46.

  Thereafter, when the time tk2 has elapsed after the second drive pulse Vg14 is inverted to the low level, the first drive pulse Vg12 is inverted to the high level, and the period T16 ends. Here, the time tk2 is set to the time required for the voltage Vd12 between the drain and source terminals of the first main switching element 12 to drop from Vi to substantially zero. The speed at which the voltage Vd12 decreases is the same as described above.

  In the period T17, the first drive pulse Vg12 becomes high level, so the first switch 12a is turned on. The other drive pulses Vg14, Vg18, Vg20 continue to be at the low level, the high level, and the low level, and the second, third, and fourth switches 14a, 18a, 20a continue to be off, on, and off. Accordingly, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the first and third switches 12a and 18a, and a voltage is generated in the primary winding 30 and the secondary windings 32a and 32b. do not do.

  When the first switch 12a is turned on, the voltage Vd12 has dropped to substantially zero, so that ideal soft switching is performed. When the first switch 12a is turned on, as shown in FIG. 5C, the path of the energy emission current of the resonant inductor 34 is the resonant inductor 34, the first switch 12a, the third switch 18a, the primary winding 30, It becomes a path of the resonant inductor 34. In addition, since the voltage Vd14 between the drain and source terminals of the second main switching element 14 has reached Vi, the current for charging the second parasitic capacitor 14b does not flow.

  Further, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 continues the operation of releasing the excitation energy, and the energy discharge current of the smoothing inductor 46 is changed to the smoothing inductor 46, the smoothing capacitor 48, the load 52, It flows through the path of the rectifying elements 92a and 92b, the secondary windings 32a and 32b, and the smoothing inductor 46.

  Thereafter, the third drive pulse Vg18 is inverted to the low level at the timing before the end of the cycle tsw by the time tk1, and the period T17 ends.

  When the period T18 starts, the third drive pulse Vg18 goes to a low level, so the third switch 18a is turned off. The other drive pulses Vg12, Vg14, Vg20 continue to be at a high level, a low level, and a low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be on, off, and off. As a result, no voltage is generated in the primary winding 30 and the secondary windings 32a and 32b.

  When the third switch 18a turns off, as shown in FIG. 5D, the energy emission current of the resonant inductor 34 is changed to the resonant inductor 34, the first switch 12a, the DC input power supply 26, the fourth parasitic capacitor 20b, the primary The current flows through the winding 30 and the path of the resonant inductor 34, and the charge of the fourth parasitic capacitor 20b is discharged. The voltage Vd20 between the drain and source terminals of the fourth main switching element 20 decreases toward zero. At the same time, a current for charging the third parasitic capacitor 18 b flows through the path of the resonant inductor 34, the first switch 12 a, the third parasitic capacitor 18 b, the primary winding 30, and the resonant inductor 34. The drain-source voltage Vd18 increases toward the voltage Vi.

  Further, since no voltage is generated in the secondary windings 32a and 32b, the smoothing inductor 46 continues the operation of releasing the excitation energy, and the energy discharge current of the smoothing inductor 46 is changed to the smoothing inductor 46, the smoothing capacitor 48, the load 52, It flows through the path of the rectifying elements 92a and 92b, the secondary windings 32a and 32b, and the smoothing inductor 46.

  Thereafter, the fourth drive pulse Vg20 is inverted to a high level at the end point of the constant cycle tsw, and the period T18 ends. Here, the time tk, which is the length of the period T18, is set to the time required for the voltage Vd20 between the drain and source terminals of the fourth main switching element 20 to drop from Vi to substantially zero. The speed at which the voltage Vd20 decreases is the same as described above. When the period T18 ends, the process returns to the period T11, the soft switching of the fourth main switching element is performed, and the operations in the above-described periods T11 to T18 are repeated.

  In the switching power supply device 90, since the first and rectifying elements 92a and 92b of the rectifying and smoothing circuit 94 are diodes, the backflow operation as shown in FIGS. 19 to 22 does not occur when an external voltage is applied.

  As described above, the switching power supply device 90 of the first embodiment makes the configuration of the switching control circuit 96 simpler than the conventional phase shift control switching control circuit (for example, the switching control circuit 56 of FIG. 14). be able to. Further, as the pulse width modulation circuit 98, a control IC that outputs two modulation pulses Vh1 and Vh2 for the first and second arms 16 and 22, that is, a general full-bridge control IC that is not for phase shift control, is used. There is an advantage that it can be used. There are few types of control ICs for phase shift control on the market, but in the case of general full-bridge type control ICs, there are multiple types of commercial products with their own characteristics (for example, overcurrent protection and other attached functions) Therefore, it is possible to select an easy-to-use device according to the application of the switching power supply device 90. On the other hand, a general full-bridge type control IC does not consider soft switching of the main switching element, but in the case of the switching power supply device 90, the first and second arm drive circuits 100 and 102 having a simple configuration and By providing the resonant inductor 34, the first to fourth main switching elements 12, 14, 18, and 20 can be soft-switched, and switching loss and switching noise can be easily suppressed.

  Next, a second embodiment of the switching power supply device according to the present invention will be described with reference to FIGS. Here, the same configurations as those of the switching power supply devices 10 and 90 are denoted by the same reference numerals, and description thereof is omitted.

  As shown in FIGS. 6 and 7, the switching power supply 148 of the second embodiment includes a rectifying unit 36 that performs synchronous rectification and a new switching control circuit 150 as in the conventional switching power supply 10. Other configurations are the same as those of the switching power supply device 90. Hereinafter, the configuration of the rectifier 36 and the switching control circuit 150 will be described.

  The rectifying unit 36 includes first and second rectifying elements 42a and 42b, which are N-channel MOS type FETs, and a synchronous rectification driving circuit 44 that drives the first and second rectifying elements 42a and 42b. For example, as shown in FIG. 8, the synchronous rectification drive circuit 44 includes an auxiliary winding 152 provided in the main transformer 28 and two first and second auxiliary diodes 154a and 154b. The auxiliary winding 152 has one end on the dot side connected to the gate terminal of the second rectifying element 42b, and one end on the side opposite to the dot connected to the gate terminal of the first rectifying element 42a. The first auxiliary diode 154a is connected between the gate and source terminals of the first rectifying element 42a with the cathode terminal as the gate terminal side. The second auxiliary diode 154b is connected between the gate and source terminals of the second rectifying element 42b with the cathode terminal as the gate terminal side. Accordingly, when “dot-side positive voltage” is generated at both ends of the secondary windings 32a and 32b and the auxiliary winding 152, the first rectifying element 42a is turned off, the second rectifying element 42b is turned on, and the dot When a positive voltage is generated on the opposite side, the first rectifying element 42a is turned on and the second rectifying element 42b is turned off. When no voltage is generated, the first rectifying element 42a and the second rectifying element 42 Both 42b can be turned on. In other words, the synchronous rectification drive circuit 44 detects the voltage corresponding to the secondary windings 32a and 32b to turn on and off the rectifier elements 42a and 42b, and when the voltage is generated in the secondary winding 32a or 32b. The generated voltage is rectified in the positive direction and input to the input terminal of the smoothing unit 38. When no voltage is generated in the secondary winding 32a or 32b, both ends of the secondary windings 32a and 32b Is short-circuited to the input end of the smoothing unit 38. The configuration of the synchronous rectification drive circuit 44 can be replaced with various configurations as disclosed in, for example, Japanese Patent Laid-Open No. 2003-189608.

  The switching control circuit 150 is a circuit for turning on and off the main switching elements 12, 14, 18, and 20 so that the output voltage signal Vo1 approaches the target value Vr. The error control circuit 58, the pulse width modulation circuit 98, the first And second arm drive circuits 156 and 158. As described above, the error amplification circuit 58 is, for example, an analog inverting amplifier, and amplifies the difference between the output voltage signal Vo1 and the target voltage Vr and outputs an error amplification signal Vo2. The pulse width modulation circuit 98 is also composed of the pulse width modulation means 66 and the pulse distribution circuit 104 as described above. That is, the pulse width modulation unit 66 modulates the error amplification signal Vo2 of the error amplification circuit 58 and outputs the modulation pulse Vh, and the pulse distribution circuit 104 controls the modulation pulse Vh for the first arm 16 and the second arm 22. The two modulation pulses Vh1 and Vh2 are output. The first and second modulation pulses Vh1 and Vh2 are pulse voltages having high time ratios equal to each other, the phase in which the logic is inverted is shifted by 180 degrees, and each time ratio changes according to the error amplification signal Vo2. .

  The first arm driving circuit 156 includes a first delay circuit 106, a first gate charging circuit 160, and a first gate discharging circuit 162. As described above, the first delay circuit 106 outputs the second drive pulse Vg14 obtained by delaying the first modulation pulse Vh1 of the pulse distribution circuit 104 by the time tk1 between the gate and source terminals of the second main switching element 14. To do.

  The first gate charging circuit 160 rectifies the DC voltage of the DC power source 112 for the control circuit and outputs it to the cathode terminal, the cathode terminal of the first rectifier diode 114 and the first main switching element 12. And a first smoothing capacitor 116 connected between the source terminals. Further, a first transistor 164 which is a PNP transistor whose emitter terminal is connected to the cathode terminal of the first rectifier diode 114 and whose collector terminal is connected to the gate terminal of the first main switching element 12, and whose anode terminal is the first transistor 164. Are connected in series to the connection point of the first diode 166 and the first diode 166 whose cathode terminal is connected to the drain terminal of the first main switching element, and limits the current of the first diode 166. And a first resistor 168. The first transistor 164 observes the voltage Vd14 between the drain and source terminals of the second main switching element 14, and the voltage Vd14 is changed from the input voltage Vi (the voltage at the drain terminal of the first main switching element 12) to the DC power supply 112. When the voltage obtained by subtracting the voltage (the voltage of the first smoothing capacitor) is exceeded, the base current flows and turns on (assuming that the saturation voltage between the base and emitter of the first transistor 164 is zero). When the first transistor 164 is turned on, the gate-source terminal of the first main switching element 12 is charged, and the first main switching element 12 is turned on.

  The first gate discharge circuit 162 includes a first drive transformer 122 having an input winding 122a and an output winding 122b, a first switching transistor 124 connected in series to the input winding 122a, and a gate of the first main switching element 12. A first pull-out transistor 126 is connected so that the source terminals can be short-circuited and opened and driven by the voltage of the output winding 122b, and a direct current is connected to both ends of the series circuit of the input winding 122a and the first switching transistor 124. A power supply 112 is connected. The dots of the windings 122a and 122b of the first drive transformer 122 represent the polarity. In addition, the first narrow pulse Vhs1 indicating the high level for a short time (for example, about 50 to 300 nsec) is received at the timing when the first modulation pulse Vh1 is inverted to the high level upon receiving the first modulation pulse Vh1 of the pulse distribution circuit 104. A first narrow pulse generation circuit 170 for generating is provided, and an output thereof is connected to a base terminal of the first switching transistor 124.

  The first switching transistor 124 is turned on for a short time during which the first narrow pulse Vhs1 is at a high level, thereby generating a pulse voltage having the same phase as the first narrow pulse Vhs1 at both ends of the input winding 122a. This narrow pulse voltage is transmitted between the gate and source terminals of the first extraction transistor 126 from the output winding 122b through the first drive transformer 122. When the first narrow pulse Vhs1 is at a high level, the first extraction transistor 126 is turned on. To do. When the first extraction transistor 126 is turned on, the gate-source terminal of the first main switching element 12 is discharged, and the first main switching element 12 is turned off. After that, the first narrow pulse Vhs1 quickly becomes low level and the first extraction transistor 126 is turned off, but the low level between the gate and source terminals is maintained until charging by the first charging circuit 160, The first main switching element 12 is kept off. Thus, the first driving pulse Vg12 for driving the first main switching element 12 is generated by the first gate charging circuit 160 and the first gate discharging circuit 162.

  Similar to the first arm driving circuit 156, the second arm driving circuit 158 includes a second delay circuit 128, a second gate charging circuit 172, and a second gate discharging circuit 174. As described above, the second delay circuit 128 outputs the fourth drive pulse Vg20 obtained by delaying the second modulation pulse Vh2 of the pulse distribution circuit 104 by the time tk1 between the gate and source terminals of the fourth main switching element 20. To do.

  The second gate charging circuit 172 rectifies the DC voltage of the DC power source 112 for the control circuit and outputs it to the cathode terminal, the cathode terminal of the second rectifier diode 134 and the third main switching element 18. And a second smoothing capacitor 136 connected to the source terminal. Furthermore, the second transistor 176 is a PNP transistor whose emitter terminal is connected to the cathode terminal of the second rectifier diode 134 and whose collector terminal is connected to the gate terminal of the third main switching element 18, and whose anode terminal is the second transistor 176. Is connected in series to the connection point of the second diode 178 and the second diode 178, the cathode terminal of which is connected to the drain terminal of the third main switching element, and the current of the second diode 178 is limited. A second resistor 180. The second transistor 176 observes the voltage Vd20 between the drain and source terminals of the fourth main switching element 20, and the voltage Vd20 is changed from the input voltage Vi (the voltage at the drain terminal of the third main switching element 12) to the DC power source 112. When the voltage obtained by subtracting the voltage (the voltage of the first smoothing capacitor) is exceeded, the base current flows and turns on (assuming that the saturation voltage between the base and emitter of the second transistor 176 is zero). When the second transistor 176 is turned on, the gate-source terminal of the third main switching element 18 is charged, and the third main switching element 18 is turned on.

  The second gate discharge circuit 174 includes a second drive transformer 142 having an input winding 142a and an output winding 142b, a second switching transistor 144 connected in series to the input winding 142a, and gates of the third main switching element 18. A second extraction transistor 146 connected between the source terminals so as to be short-circuited and opened and driven by the voltage of the output winding 142b; and a DC power source at both ends of the series circuit of the input winding 142a and the second switching transistor 144 112 is connected. The dots of the windings 142a and 142b of the second drive transformer 142 represent the polarity. Also, the second modulation pulse Vh2 of the pulse distribution circuit 104 is received, and a second narrow pulse Vhs2 indicating a high level is generated at a timing when the second modulation pulse Vh2 is inverted to a high level (for example, a time of about 50 to 300 nsec). The second narrow pulse generation circuit 182 is provided, and its output terminal is connected to the base terminal of the second switching transistor 144.

  The second switching transistor 144 generates a pulse voltage having the same phase as the second narrow pulse Vhs2 at both ends of the input winding 142a by being turned on for a short time when the second narrow pulse Vhs2 is at a high level. This narrow pulse voltage is transmitted through the second drive transformer 142 from the output winding 142b between the gate and source terminals of the second extraction transistor 146, and when the narrow pulse Vhs2 is at the high level, the second extraction transistor 146 is turned on. When the second extraction transistor 146 is turned on, the gate-source terminal of the third main switching element 18 is discharged, and the third main switching element 18 is turned off. After that, the second narrow pulse Vhs2 quickly becomes low level and the second extraction transistor 146 is turned off, but the low level between the gate and source terminals is maintained until charging by the second charging circuit 172, The third main switching element 18 continues to be turned off. Thus, the third drive pulse Vg18 for driving the third main switching element 18 is generated by the second gate charging circuit 172 and the second gate discharging circuit 174.

  With the above configuration, the switching control circuit 150 outputs the first to fourth drive pulses Vg12, Vg14, Vg18, and Vg20 to turn on and off the first to four main switching elements 12, 14, 18, and 20. The input voltage Vi is applied to the primary winding 30 of the main transformer 28 in the positive direction (period T11), the second period (period T12 to T14) in which no voltage is applied, and the input voltage Vi is applied in the negative direction. The third period to be applied (period T15) and the fourth period in which no voltage is applied again (period T16 to T18) are defined as one cycle, and the total time of the first and second periods and the total time of the third and fourth periods The switching control is performed so as to be equal. The switching cycle tsw (the length of one cycle) is constant.

  Next, the operation during normal operation of the switching power supply device 148 will be described. The time chart and the equivalent circuit during normal operation are the same as in the case of the switching power supply 90 described above, and are represented as shown in FIGS. However, the switching power supply device 148 is different from the switching power supply device 90 in the method of determining the timing when the period T12 ends and the timing when the period T16 ends.

  During the period T12, after the fourth drive pulse Vg20 is inverted to the low level, the second gate charging circuit 172 indicates that the drain-source voltage Vd20 of the fourth main switching element 20 has increased and has become close to the input voltage Vi. Is detected, and when the second transistor 176 is turned on and the third drive pulse Vg18 is inverted to the high level, the period T12 ends. Accordingly, when the third main switching element 18 is turned on after the transition to the period T13, the voltage Vd18 is reduced to substantially zero, so that ideal soft switching is performed.

  Similarly, during the period T16, after the second drive pulse Vg14 is inverted to a low level, the first gate indicates that the drain-source voltage Vd14 of the second main switching element 14 has risen and is close to the input voltage Vi. When the charging circuit 160 detects that the first transistor 164 is turned on and the first drive pulse Vg12 is inverted to a high level, the period T16 ends. Accordingly, when the first main switching element 12 is turned on after the transition to the period T17, the ideal soft switching is performed because the voltage Vd12 decreases to substantially zero.

  Here, the rate at which the voltages Vd12 and Vd18 decrease is determined by the resonant frequency of the resonant inductor 34 and the corresponding parasitic capacitor, so the length of the periods T12 and T16 is the switching control circuit 96 of the switching power supply 90 described above. Like, it will be almost time tk2.

  Next, the backflow operation when the external power supply 88 is connected to the output terminals 50a and 50b of the switching power supply device 148 and a voltage Voh higher than the output voltage set value Vor is applied will be described. Here, the switching power supply 148 has a minimum time so that the time during which the voltage Vi is applied to both ends of the primary winding 30 (periods T11 and T15 during normal operation in FIG. 3) does not become zero. It is assumed that the circuit 150 performs control within a range exceeding this minimum time.

  The operation of one period (period tsw) of switching of the switching power supply device 148 can be described by being divided into periods T21 to T28 as shown in the time chart of FIG. 9, and each main switching element 12, 14 in each period is described. , 18, 20 and the states of the rectifying elements 42a and 42b and the path through which the main current flows are expressed as in the equivalent circuits of FIGS.

  First, in the period T21, the drive pulses Vg12, Vg14, Vg18, and Vg20 are high level, low level, low level, and high level, the first and fourth switches 12a and 20a are turned on, and the second and third switches 14a. , 18a are off. Accordingly, “a positive voltage Vi on the opposite side of the dot” is applied to both ends of the primary winding 30 of the main transformer 28, and “a positive voltage on the opposite side of the dot is applied to both ends of the secondary windings 32 a and 32 b and the auxiliary winding 152. Is generated, the rectifying element 42a is turned on and the rectifying element 42b is turned off.

  The smoothing inductor 46 has accumulated the excitation energy before the start of the period T21. When the smoothing inductor 46 shifts to the period T21, the operation for releasing the excitation energy starts. As shown in FIG. 10A, the energy emission current of the smoothing inductor 46 flows through the path of the smoothing inductor 46, the secondary winding 32a, the rectifying element 42a, the external power supply 88, and the smoothing inductor 46, and further passes through the main transformer 28. Through the path of the primary winding 30, the resonant inductor 34, the first switch 12 a, the DC input power supply 26, the fourth switch 20 a, and the primary winding 30, and the excitation energy of the smoothing inductor 46 is directed toward the DC input power supply 26. It is regenerated. Therefore, as shown in FIG. 9, the current I46 of the smoothing inductor 46 has a waveform that linearly decreases from a negative value toward zero. At this time, excitation energy is accumulated in the resonant inductor 34 by the energy emission current flowing through the resonant inductor 34.

  The switching control circuit 150 receives the output voltage signal Vo1 output from the output voltage detection circuit 54, that is, the output voltage signal Vo1 corresponding to the voltage Voh between the output terminals 50a and 50b, and “the output voltage Vo1 is lower than the target value Vr. In order to make the output voltage signal Vo1 low, control is performed to shorten the length of the period T21 (that is, the time during which the voltage Vi is applied to the primary winding 30) to a specified minimum time. By this control, the fourth drive pulse Vg20 is inverted to the low level, and the period T21 ends.

  In the period T22, the fourth drive pulse Vg20 becomes low level, so the fourth switch 20a is turned off. The other drive pulses Vg12, Vg14, and Vg18 continue to be high level, low level, and low level, and the first, second, and third switches 12a, 14a, and 18a continue to be on, off, and off. When the fourth switch 20a is turned off, the path of the energy emission current from the smoothing inductor 46 is switched from the fourth switch 20a to the fourth parasitic diode 20c as shown in FIG. There is no change in operation, and the same state as in the period T21 is continued, and as shown in FIG. 9, the current I46 of the smoothing inductor 46 decreases linearly toward zero. While the fourth parasitic diode 20c is conducting due to the energy discharge current, the voltage Vd20 does not increase, so the second gate charging circuit 172 does not operate and the third switch 18a continues to be turned off. Thereafter, when the smoothing inductor 46 releases all the excitation energy and the current I46 becomes zero, the period T22 ends.

  In the period T23, since the energy emission current of the smoothing inductor 46 is lost, the fourth parasitic diode 20c is turned off, the fourth parasitic capacitor 20b is charged and the voltage Vd20 rises to the input voltage Vi, and the second gate charging circuit. 172 operates, the third drive pulse Vg18 becomes high level, and the third switch 18a is turned on. The other drive pulses Vg12, Vg14, Vg20 continue to be at a high level, a low level, and a low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be on, off, and off. Therefore, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the first and third switches 12a and 18a, and no voltage is generated at both ends of the primary winding 30. Then, since no voltage is generated in the secondary windings 32a and 32b, not only the rectifying element 42a but also the rectifying element 42b is turned on.

  In this state, the external power supply 88 starts to store the excitation energy in the smoothing inductor 46. The energy storage current from the external power supply 88 flows through the path of the external power supply 88, the smoothing inductor 46, the secondary windings 32a and 32b, the rectifier elements 42a and 42b, and the external power supply 88, as shown in FIG. Therefore, as shown in FIG. 9, the current I46 of the smoothing inductor 46 has a waveform that increases linearly in the negative direction. At this time, the resonant inductor 34 performs an operation of releasing the excitation energy accumulated before the start of the period T23, so that the resonant inductor 34, the first switch 12a, the third switch 18a, the primary winding 30, the resonant inductor 34 The energy emission current flows through the path. Thereafter, the first drive pulse Vg12 is inverted to a low level at a timing before time tk1 before the intermediate point of the period tsw, and the period T23 ends.

  In the period T24, the first drive pulse Vg12 becomes low level, so the first switch 12a is turned off. The other drive pulses Vg14, Vg18, Vg20 continue to be at the low level, the high level, and the low level, and the second, third, and fourth switches 14a, 18a, 20a continue to be off, on, and off. When the first switch 12a is turned off, the energy emission current path from the resonant inductor 34 is switched from the first switch 12a to the first parasitic diode 12c as shown in FIG. There is no change in operation, and the same state as in the period T23 is continued. Further, since the rectifier elements 42a and 42b are kept on as in the period T23, the current I46 of the smoothing inductor linearly increases in the negative direction as shown in FIG. Thereafter, the second drive pulse Vg14 is inverted to a high level at an intermediate point of the constant period tsw, and the period T24 ends.

  When the period T25 is entered, the second drive pulse Vg14 becomes high level, so that the switch 14a is turned on. The other drive pulses Vg12, Vg18, Vg20 continue to be at low level, high level, and low level, and the first, third, and fourth switches 12a, 18a, 20a continue to be off, on, and off. Accordingly, “dot-side positive voltage Vi” is applied to both ends of the primary winding 30 of the main transformer 28, and “dot-side positive voltage” is generated at both ends of the secondary windings 32a and 32b. Is turned off, and the rectifying element 42b is turned on.

  The smoothing inductor 46 has accumulated the excitation energy before the start of the period T25, and starts the operation of releasing the excitation energy when the period T25 starts. As shown in FIG. 11A, the energy emission current of the smoothing inductor 46 flows through the path of the smoothing inductor 46, the secondary winding 32b, the rectifying element 42b, the external power supply 88, and the smoothing inductor 46, and further passes through the main transformer 28. Through the primary winding 30, the third switch 18 a, the DC input power supply 26, the second switch 14 a, the resonant inductor 34, and the primary winding 30. The excitation energy of the smoothing inductor 46 is directed toward the DC input power supply 26. It is regenerated. Therefore, as shown in FIG. 9, the current I46 of the smoothing inductor 46 has a waveform that linearly decreases from a negative value toward zero. At this time, excitation energy is accumulated in the resonant inductor 34 by the energy emission current flowing through the resonant inductor 34.

  As described in the period T21, the switching control circuit 150 sets the length of the period T25 (that is, the time during which the voltage Vi is applied to the primary winding 30) to the specified minimum in order to reduce the output voltage signal Vo1. Control to shorten the time. By this control, the second drive pulse Vg14 is inverted to the low level, and the period T25 ends.

  When the period T26 is entered, the second drive pulse Vg14 becomes low level, so the second switch 14a is turned off. The other drive pulses Vg12, Vg18, Vg20 continue to be at low level, high level, and low level, and the first, third, and fourth switches 12a, 18a, 20a continue to be off, on, and off. When the second switch 14a is turned off, the energy emission current path from the smoothing inductor 46 is switched from the second switch 14a to the second parasitic diode 14c as shown in FIG. There is no change in operation, and the same state as in the period T25 is continued, and as shown in FIG. 9, the current I46 of the smoothing inductor decreases linearly toward zero. While the second parasitic diode 14c is conducting due to the energy discharge current, the voltage Vd14 does not increase, so the first gate charging circuit 160 does not operate and the first switch 12a continues to be turned off. Thereafter, when the smoothing inductor 46 releases all the excitation energy and the current I46 becomes zero, the period T26 ends.

  When the period T27 is entered, the energy discharge current of the smoothing inductor 46 disappears, so the second parasitic diode 14c is turned off, the second parasitic capacitor 14b is charged, and the voltage Vd14 rises to the input voltage Vi. 160 operates, the first drive pulse Vg12 becomes high level, and the first switch 12a is turned on. The other drive pulses Vg14, Vg18, Vg20 continue to be at the low level, the high level, and the low level, and the second, third, and fourth switches 14a, 18a, 20a continue to be off, on, and off. Therefore, both ends of the series circuit of the primary winding 30 and the resonant inductor 34 are short-circuited through the first and third switches 12a and 18a, and no voltage is generated in the voltage across the primary winding 30. Then, since no voltage is generated in the secondary windings 32a and 32b, not only the rectifying element 42b but also the rectifying element 42a is turned on.

  In this state, the external power supply 88 starts to store the excitation energy in the smoothing inductor 46. The energy storage current from the external power supply 88 flows through the path of the external power supply 88, the smoothing inductor 46, the secondary windings 32a and 32b, the rectifying elements 42a and 42b, and the external power supply 88, as shown in FIG. Therefore, as shown in FIG. 9, the current I46 of the smoothing inductor 46 has a waveform that increases linearly in the negative direction. At this time, the resonant inductor 34 performs an operation of releasing the excitation energy accumulated before the start of the period T27, and therefore, the resonant inductor 34, the primary winding 30, the third switch 18a, the first switch 12a, and the resonant inductor 34. The energy emission current flows through the path. Thereafter, the third drive pulse Vg18 is inverted to a low level at a timing earlier than the end point of the cycle tsw by time tk1, and the period T27 ends.

  In the period T28, the third drive pulse Vg18 becomes low level, so the third switch 18a is turned off. The other drive pulses Vg12, Vg14, Vg20 continue to be at a high level, a low level, and a low level, and the first, second, and fourth switches 12a, 14a, 20a continue to be on, off, and off. When the third switch 18a is turned off, the energy emission current path from the resonant inductor 34 is switched from the third switch 18a to the third parasitic diode 18c as shown in FIG. There is no change in operation, and the same state as in the period T27 is continued. Further, since the rectifying elements 42a and 42b are kept on as in the period T27, the smoothing inductor current I46 linearly increases in the negative direction as shown in FIG. Thereafter, the fourth drive pulse Vg20 is inverted to the high level at the end point of the constant cycle tsw, and the period T28 ends. When the period T28 ends, the process returns to the period T21, and the operations in the above-described periods T21 to T28 are repeated.

  Here, looking at the current I46 of the smoothing inductor 46, the excitation energy is released from the smoothing inductor 46 in the periods T21 to T24 in the first half of the cycle tsw during the periods T21 and T22. Do not exit until is zero. Accordingly, the periods T23 and T24 in which the excitation energy is accumulated after all the excitation energy is released, and the lengths of the periods T23 and T24 are shortened by the length of the periods T21 and T22, so the excitation energy is cumulative. Will not accumulate. As a result, the negative value of the current I46 of the smoothing inductor 46 becomes the same at the start of the period T21 and at the end of the period T24. Similarly, in the period T25 to T28 in the latter half of the period tsw, the value in the negative direction of the current I46 of the smoothing inductor 46 is the same at the start of the period T25 and at the end of the period T28. Therefore, even if the operations in the periods T21 to T28 are repeated, the smoothing inductor 46 is demagnetized and does not become magnetically saturated, so that a large reverse current does not flow from the external power supply 88 to the smoothing inductor 46.

  As described above, the switching power supply device 148 of the second embodiment can obtain the same effects as the switching power supply device 90 of the first embodiment, and further, the first and second gate discharge circuits 162 and 174. Since the first and second narrow pulse generation circuits 170 and 182 are provided, there is an advantage that the first and second drive transformers 122 and 142 can be reduced in size. In the case of the switching power supply device 90, the first and second drive transformers 122 and 142 of the first and second gate discharge circuits 110 and 132 have the first and second modulation pulses Vh1, Since it functions to transmit Vh2, a certain amount of inductance is required, and it is difficult to reduce the outer shape to a certain level or less. The same applies to the first and second drive transformers 78 and 86 of the conventional switching power supply device 10 of FIG. On the other hand, in the case of the switching power supply device 148, the first and second drive transformers 122 and 142 serve to transmit the first and second narrow pulses Vhs1 and Vhs2 having a relatively short high level time. The inductance can be reduced, and the outer shape of the transformer element can be reduced by reducing the number of turns of each winding or reducing the cross-sectional area of the core.

  In addition, since the switching power supply device 148 includes the rectifying unit 36 that performs synchronous rectification in the rectifying and smoothing circuit 40, conduction loss generated in the rectifying elements 42a and 42b can be reduced. On the other hand, since it is configured to perform synchronous rectification, a reverse current operation becomes a problem when a voltage Voh higher than the output voltage set value Vor is externally applied to the synchronous rectification output terminals 50a and 50b. Detects that “the voltage Vd14 has risen to near the input voltage Vi” and inverts the first drive voltage Vg12 to a high level, and the second gate charging circuit 172 has “the voltage Vd20 has risen to near the input voltage Vi” And the synchronous rectification drive circuit 44 detects the voltage change of the secondary windings 32a and 32b and turns on / off the corresponding rectification elements 42a and 42b. By the operation, it is possible to realize an operation of releasing all the excitation energy stored in the smoothing inductor 46 every half cycle of switching, so that the magnetic saturation due to the magnetization of the smoothing inductor 46 is prevented. Sealed, a large reverse current flow. Therefore, it is safe to connect the outputs of the plurality of switching power supply devices 148 to each other and perform parallel operation.

  The present invention is not limited to the above embodiment. For example, the first and second gate charging circuits 108 and 130 in FIG. 2 may be replaced with the first and second gate charging circuits 160 and 172 in FIG. Similarly, the first and second gate discharge circuits 110 and 132 in FIG. 2 may be replaced with the first and second gate discharge circuits 162 and 174 in FIG.

The switching control circuit is not limited to the configuration of the switching control circuits 96 and 150 described above. That is, a first period in which the input voltage is applied to the primary winding in the positive direction, a second period in which no voltage is applied, a third period in which the input voltage is applied in the negative direction, and a fourth period in which no voltage is applied again. 1 cycle, and the first to fourth drive pulses for equalizing the total time of the first and second periods and the total time of the third and fourth periods, wherein the first main switching element is turned on, The second main switching element is turned off, the third main switching element is turned off, the fourth main switching element is turned on, and when the second period starts, the fourth main switching element is inverted to off, and then the third main switching element is turned on. Is turned on, and then the first main switching element is turned off before the third period starts. When the third period is entered, the second main switching element is turned on and the fourth period is entered. And the second main Invert the switching element off, then invert the first main switching element, and then output first through fourth drive pulses that invert the third main switching element off before the first period begins What is necessary is just to have the function of. A switching control circuit that realizes such a function can be configured simply as compared with a circuit that performs phase shift control, and soft switching of each main switching element can be easily realized.

  Further, instead of the center tap type rectifying and smoothing circuit 40, a known full-wave rectifying type or current doubler type rectifying and smoothing circuit may be used. For example, when a synchronous rectification rectification unit 184 is provided in the full-wave rectification type rectification smoothing circuit 183, as shown in FIG. 12A, four rectification elements 186a in a bridge configuration are formed in the secondary winding 32 of the main transformer 28. To 186d are connected, and the respective rectifying elements 186a to 186d are driven by the synchronous rectification driving circuit 188. The synchronous rectification drive circuit 188 detects the voltage of the secondary winding 32 via the auxiliary winding 152 provided in the main transformer 28, and when the voltage is generated in the secondary winding 32, the generated voltage is When the rectifying elements 186a to 186d are turned on or off so that the voltage rectified in the positive direction is input to the input terminal of the smoothing unit 38, and no voltage is generated in the secondary winding 32, the secondary winding 32 The rectifying elements 186a to 186d are turned on so that both ends of the rectifier are short-circuited to the input end of the smoothing unit 38.

  Further, when the synchronous rectification 192 is provided in the current doubler type rectification smoothing circuit 190, as shown in FIG. 12B, two rectification elements 194a and 194b are connected to the secondary winding 32 of the main transformer 28. A configuration in which the rectifying elements 194a and 194b are driven by the synchronous rectification driving circuit 196 is conceivable. In this case, the smoothing unit 198 includes two smoothing inductors 46a and 46b and a smoothing capacitor 48, and the both-ends voltage of the rectifying element 194a and the both-ends voltage of the rectifying element 194b are individually input to the two input ends, and smoothed. To synthesize. The synchronous rectification drive circuit 196 detects the voltage of the secondary winding 32 via the auxiliary winding 152 provided in the main transformer 28, and when the voltage is generated in the secondary winding 32, the generated voltage is When the rectifying elements 194a and 194b are turned on or off so as to be rectified in the positive direction and input to one of the input terminals of the smoothing unit 198, and no voltage is generated in the secondary winding 32, the secondary winding The rectifying elements 194a to 194d are turned on so that both ends of the line 32 are short-circuited to both input ends of the smoothing unit 38.

  Since the rectifying / smoothing circuits 183 and 190 have synchronous rectifying rectifying units 184 and 192, a reverse flow operation as shown in FIG. 9 may occur when a voltage higher than the output voltage setting value is externally applied to the output. For example, the first and second main switching circuits 160 and 172 shown in FIG. 7 are used to detect the voltages Vd14 and Vd20 between the drain and source terminals of the second and fourth main switching elements. If the element is turned on, it is safe as described above.

10, 90, 148 Switching power supply device 12 First main switching element 14 Second main switching element 16 First arm 18 Third main switching element 20 Fourth main switching element 22 Second arm 28 Main transformer 30 Primary windings 32, 32a , 32b Secondary winding 34 Resonant inductor 36, 92, 184, 192 Rectifier 38, 198 Smoother 40, 94, 183, 190 Rectifier smoother 42a, 42b, 92a, 92b, 186a to 186d, 194a, 194b Rectifier 44, 188, 196 Synchronous rectification drive circuit 46, 46a, 46b Smoothing inductor 48 Smoothing capacitor 54 Output voltage detection circuit 56, 96, 150 Switching control circuit 58 Error amplification circuit 62, 100, 156 First arm drive circuit 64, 102, 158 Second arm drive circuit 68 104 Pulse distribution circuit 98 Pulse width modulation circuit 106 First delay circuit 108, 160 First gate charging circuit 110, 162 First gate discharge circuit 112 DC power supply 114 First rectifier diode 116 First smoothing capacitor 122 First drive transformer 122a Input Winding 122b Output winding 124 First switching transistor 126 First extraction transistor 128 Second delay circuit 130, 172 Second gate charging circuit 132, 174 Second gate discharging circuit 134 Second rectifier diode 136 Second smoothing capacitor 142 Second Drive transformer 142a Input winding 142b Output winding 144 Second switching transistor 146 Second extraction transistor 164 First transistor 166 First diode 168 First resistor 170 First narrow pulse generation circuit 176 Second transformer Register 178 Second diode 180 Second resistor 182 second narrow pulse generating circuit
Io output current
I46 Smoothing inductor current
Vg12, Vg14, Vg18, Vg20 1st to 4th drive pulses
Vh modulation pulse
Vh1, Vh2, Vh3, Vh4 1st to 4th modulation pulses
Vhs1, Vhs2 First narrow pulse, second narrow pulse
Vi input voltage
Vo output voltage
Vo1 output voltage signal
Vo2 error amplification signal
Vr target value

Claims (7)

A first arm composed of a series circuit of a first main switching element on the high side and a second main switching element on the low side, and a DC input power supply connected to both ends thereof;
A second arm composed of a series circuit of a third main switching element on the high side and a fourth main switching element on the low side, and a DC input power supply connected to both ends thereof;
A primary transformer having a primary winding and a secondary winding, wherein the primary winding is provided between a midpoint of the first arm and a midpoint of the second arm;
A resonant inductor provided between the midpoint of the first arm and the midpoint of the second arm in a position in series with the primary winding;
A rectifying and smoothing circuit connected to the secondary winding and configured to rectify the AC voltage generated in the secondary winding, and a smoothing unit that smoothes the rectified voltage and outputs a DC output voltage;
An output voltage detection circuit for detecting an output voltage or a voltage corresponding thereto and outputting an output voltage signal;
A full-bridge type switching comprising: a switching control circuit that outputs first to fourth driving pulses to each of the main switching elements and that turns on and off the main switching elements so that the output voltage signal approaches a target value. In power supply,
The switching control circuit includes:
The first period in which the input voltage is applied to the primary winding in the positive direction, the second period in which no voltage is applied, the third period in which the input voltage is applied in the negative direction, and the fourth period in which no voltage is applied again are 1 The first to fourth drive pulses having a period and equalizing a total time of the first and second periods and a total time of the third and fourth periods,
To the first period, the first main switching element ON, the second main switching element off, the third main switching element off, by turning on the fourth main switching element,
Upon entering the second period, the fourth main switching element is inverted off, then the third main switching element is inverted on, and then the first main switching element before the third period starts. Flip the element off,
When entering the third period, the second main switching element is turned on,
Upon entering the fourth period, the second main switching element is inverted off, then the first main switching element is inverted on, and then the third main switching element before the first period starts. Flip the element off,
When the first period is entered again, the first to fourth drive pulses for inverting the fourth main switching element to ON are output ,
The switching control circuit distributes an error amplification circuit that compares the output voltage signal with a target voltage and outputs an error amplification signal obtained by amplifying the difference, and a modulation pulse obtained by pulse-modulating the error amplification signal. A pulse width modulation circuit for outputting a first modulation pulse for one arm and a second modulation pulse for a second arm; and generating the first and second drive pulses based on the first modulation pulse, A first arm driving circuit for outputting between the gate and source terminals of the first and second main switching elements; and generating the third and fourth driving pulses based on the second modulation pulse, respectively. A second arm drive circuit that outputs between the gate and source terminals of the main switching element,
The first arm driving circuit includes:
A first delay circuit that outputs the second drive pulse that is input to the first modulation pulse and that is inverted to a high level after a lapse of a fixed time after the first modulation pulse is inverted to a high level;
By observing the potential at the middle point of the first arm and detecting that the potential has increased, the first drive pulse is increased by charging between the gate and source terminals of the first main switching element. A first gate charging circuit that inverts to a level;
A first gate discharge circuit that inverts the first drive pulse to a low level by discharging between the gate and source terminals of the first main switching element when the first modulation pulse is at a high level;
The second arm drive circuit is
A second delay circuit that outputs the fourth drive pulse that is input to the second modulation pulse and that is inverted to a high level after a lapse of a fixed time after the second modulation pulse is inverted to a high level;
By observing the potential at the middle point of the second arm and detecting that the potential has increased, the third drive pulse is increased by charging between the gate and source terminals of the third main switching element. A second gate charging circuit to invert to the level;
A second gate discharge circuit that inverts the third drive pulse to a low level by discharging between the gate and source terminals of the third main switching element when the second modulation pulse is at a high level. switching power supply apparatus characterized by there. The first gate charging circuit includes:
A first rectifier diode that rectifies a DC voltage for a control circuit and outputs the rectified voltage to a cathode terminal; and a first smoothing capacitor connected between the cathode terminal of the first rectifier diode and the source terminal of the first main switching element A first transistor that is a PNP transistor having an emitter terminal connected to the cathode terminal of the first rectifier diode, a collector terminal connected to the gate terminal of the first main switching element, and an anode terminal of the first transistor. A first diode connected to the base terminal side and having a cathode terminal connected to the drain terminal side of the first main switching element, and a current flowing through the first diode inserted in series at a connection point of the first diode Consisting of a first resistor to limit,
The second gate charging circuit is
A second rectifier diode that rectifies a DC voltage for the control circuit and outputs the rectified voltage to the cathode terminal; and a second smoothing capacitor connected between the cathode terminal of the second rectifier diode and the source terminal of the third main switching element A second transistor that is a PNP transistor having an emitter terminal connected to the cathode terminal of the second rectifier diode, a collector terminal connected to the gate terminal of the third main switching element, and an anode terminal of the second transistor. A second diode connected to the base terminal side and having a cathode terminal connected to the drain terminal side of the third main switching element, and a current flowing through the second diode is inserted in series at the connection point of the second diode. The switching power supply device according to claim 1, wherein the switching power supply device includes a second resistor to be limited. The first gate discharge circuit is connected between the first drive transformer for transmitting the first modulation pulse and the gate-source terminal of the first main switching element, and is turned on / off according to the output of the first drive transformer. A first extraction transistor, and when the first modulation pulse is at a high level, the first extraction transistor is turned on to discharge between the gate and source terminals of the first main switching element,
The second gate discharge circuit is connected between the second drive transformer for transmitting the second modulation pulse and the gate-source terminal of the third main switching element, and is turned on / off according to the output of the second drive transformer. A second extraction transistor is provided, and when the second modulation pulse is at a high level, the second extraction transistor is turned on to discharge between the gate and source terminals of the third main switching element. The switching power supply device described . The first gate discharge circuit is provided with a first narrow pulse generation circuit that outputs a first narrow pulse that becomes a high level for a short time when the input first modulation pulse is inverted to a high level. One driving transformer is the first narrow pulse, and transmits timing information at which the first modulation pulse is inverted to a high level,
The second gate discharge circuit includes a second narrow pulse generation circuit that outputs a second narrow pulse that becomes a high level for a short time when the input second modulation pulse is inverted to a high level. The switching power supply device according to claim 3, wherein the two-drive transformer transmits timing information that is the second narrow pulse and the second modulation pulse is inverted to a high level.   The switching power supply device according to claim 1, wherein the resonant inductor is a leakage inductance inside the main transformer. The rectifying unit of the rectifying and smoothing circuit includes a rectifying element for synchronous rectification,
When the voltage of the secondary winding of the main transformer or a voltage corresponding thereto is detected to turn on and off the rectifier element, when a voltage is generated in the secondary winding, the generated voltage is set in the positive direction. Rectifying and input to the input terminal of the smoothing unit, and when no voltage is generated in the secondary winding, both ends of the secondary winding are short-circuited to the input terminal of the smoothing unit The switching power supply device according to any one of claims 1 to 5, wherein the switching power supply device comprises a synchronous rectification driving circuit. 7. The switching power supply device according to claim 6, wherein each of the first to fourth main switching elements and the synchronous rectification element is configured by an N-channel MOS FET.

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