JP5672145B2 - Rotating machine control device - Google Patents

Description

  The present invention relates to a DC / AC converter circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, by turning on / off the switching element of the DC / AC converter circuit. The present invention relates to a control device for a rotating machine that controls a control amount having at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a magnetic flux of the rotating machine.

  As this type of control device, for example, as seen in Patent Document 1 below, the current of the three-phase motor is predicted for each case where the operation state of the inverter is set variously, and the predicted current and the command current are An apparatus has been proposed that performs so-called model predictive control for operating an inverter so as to achieve an operation state in which a deviation can be minimized. According to this, since the inverter is operated so as to optimize the behavior of the current predicted based on the operation state of the inverter, the followability to the command current at the time of transition can be improved. For this reason, model predictive control is considered to be highly useful for applications that require particularly high performance as transient tracking characteristics, such as a motor generator control device as an in-vehicle main unit.

  By the way, the inverter includes a high-potential side switching element that connects the positive electrode of the DC voltage source to the motor generator, and a low-potential side switching element that connects the negative electrode of the DC voltage source to the motor generator. When N-channel MOS field effect transistors or IGBTs are used as these switching elements, the switching elements are turned on / off by variably operating the potential of the switching control terminal with respect to the potential of the output terminal of the switching element. In this case, the potential of the output terminal of the switching element on the low potential side becomes a fixed potential, while the potential of the output terminal of the switching element on the high potential side varies depending on on / off of the switching element on the low potential side. For this reason, the potential of the power supply of the drive circuit for turning on / off the switching element on the high potential side constantly changes.

  As one of the power sources having the above-described varying potential for the driving circuit for the switching element on the high potential side, there is one constituted by a bootstrap circuit as seen in the following Patent Document 2. This is because when a switching element on the low potential side is turned on, a capacitor whose reference potential is the output terminal of the switching element on the high potential side connected in series with the switching element is charged with the power supply for the switching element on the low potential side. Circuit. According to this, the drive circuit of the switching element on the high potential side can be driven by the charging energy of the capacitor.

JP 2008-228419 A JP 2010-158117 A

  However, when model predictive control is applied to the hardware means using the bootstrap circuit, the voltage of the capacitor for the drive circuit on the high potential side decreases, so that the switching element on the high potential side is driven appropriately. There is a risk that it will not be possible. This is because, in the model predictive control, unlike the known triangular wave PWM processing, whether or not the switching element on the low potential side is turned on is left to the left. For this reason, the OFF state of the switching element on the low potential side is continued, whereby the capacitor is not charged, and the voltage of the capacitor may be excessively reduced.

  The present invention has been made in the process of solving the above-described problems, and an object thereof is a DC / AC conversion circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source. A rotating machine that controls a controlled variable having at least one of a current flowing through the rotating machine, a torque of the rotating machine, and a magnetic flux of the rotating machine by turning on / off a switching element of the DC / AC converter circuit It is to provide a new control apparatus.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

1st invention turns ON / OFF the switching element of this DC / AC conversion circuit about the DC / AC conversion circuit provided with the switching element which selectively connects the terminal of a rotary machine to each of the positive electrode and negative electrode of DC voltage source In the control device for a rotating machine that controls a control amount having at least one of the current flowing through the rotating machine, the torque of the rotating machine, and the magnetic flux of the rotating machine, the drive circuit that drives the switching element includes: The power of a DC voltage source whose reference potential is one of a pair of ends of a current flow path of one of the switching element connected to the positive electrode and the switching element connected to the negative electrode is turned on. It is provided with a bootstrap circuit that charges a capacitor that is a power source of the other drive circuit during the period, and is fixed Predicting means for predicting the control amount when the operating state of the DC-AC converter circuit expressed by a voltage vector in the standard system is temporarily set, and the prediction based on the control amount predicted by the predicting means A determination unit that evaluates an operation state corresponding to a control amount and determines a highly evaluated operation state as an operation state of the DC-AC conversion circuit, and operates the DC-AC conversion circuit so as to be in the determined operation state An operating means, and a priority means for increasing the priority of the operating state in which either one is turned on when the operating state is determined by the determining means under a situation where the charging voltage of the capacitor decreases. Features.

  Since the operation state determined by the determining means does not necessarily charge the capacitor, there is a possibility that the voltage of the capacitor is excessively reduced. In this regard, in the above invention, such a situation is avoided by providing the priority means.

According to a second invention, in the first invention, the priority means includes a period in which one of the switching elements connected to the same terminal of the rotating machine is in an on state and the other switching element in an on state. It is characterized in that a decrease in the charging voltage is grasped based on both periods and the priority processing is performed based on this.

  The period during which any one of the switching elements is in the ON state is a charging period of the corresponding capacitor, and the period during which any other switching element is in the ON state is the discharging period of the corresponding capacitor. In the above-mentioned invention, in view of this point, by referring to both the above periods, it is possible to grasp the decrease in the voltage of the capacitor with high accuracy based on the balance of the outflow amount and the inflow amount of the capacitor.

According to a third invention, in the first or second invention, the priority means includes an estimation means for estimating that a reduction amount of the charging voltage of the capacitor has reached an upper limit value based on an operation by the operation means, The priority processing is performed based on the estimation by the estimation means.

In a fourth aspect based on the third aspect , the estimating means includes a counter for each one terminal of the rotating machine, and the counter determines whether an on state of the one of the switching elements is determined by the determining means. And the value of the counter is decreased based on one of the determination means and the ON state of the other switching element determined by the determination means, and the value of the counter is increased based on the other It is characterized by that.

  The counter is a parameter having a positive or negative correlation with the capacitor voltage. For this reason, the value of the counter can be used as an estimated value of the voltage of the capacitor.

According to a fifth aspect , in the fourth aspect , the counter is a discharge counter and is increased or decreased every time the ON state of the other switching element is determined by the determining means. And the estimation means includes a charge counter that counts the number of times that the ON state of any one of the switching elements is determined by the determining means, and any one of the ON states of the any one switching element that is determined by the operating means. And updating means for updating the discharge counter based on a value calculated by map using the value of the charge counter as an input based on switching the other switching element to an ON state.

  When the capacitor is charged, the voltage of the capacitor does not increase in proportion to time, but has nonlinearity with respect to time. In the above invention, in view of this point, the value of the discharge counter is not directly used for correction of the discharge counter, but is corrected by the discharge counter based on the value calculated by mapping from the value of the charge counter. It is possible to more accurately reflect the rising speed of the capacitor voltage.

A sixth invention is characterized in that, in the third invention, the estimating means performs the estimation based on a model equation indicating a time transition of the voltage of the capacitor.

According to a seventh invention, in the first invention, the priority means has a function of acquiring a detected value of the voltage of the capacitor.

An eighth invention, the third to seventh any crab Oite the invention, the priority means, if the amount of decrease in the voltage of the capacitor is determined to have reached the maximum value, the one of the switching The determination unit is caused to determine an operation state in which an element corresponding to the capacitor reaching the upper limit value is turned on.

In a ninth aspect based on the eighth aspect , when the priority unit determines that the amount of decrease has reached an upper limit value, the priority unit is configured to switch one of the switching elements connected to each terminal of the rotating machine. The determination unit is caused to determine an operation state in which all are turned on.

  In the above invention, the processing of the priority means can be simplified. Further, the change rate of the difference between the control amount and its command value is smaller when the zero voltage vector is used than when the effective voltage vector is used in the low modulation rate region. For this reason, at least at a low modulation rate, it is possible to suppress an error in the control amount caused by forcibly using the operation state prioritized by the priority unit.

In a tenth aspect based on the ninth aspect , on the condition that the priority means does not charge the capacitor that is determined to have reached the upper limit value in the operation state evaluated by the determination means. The determination unit is configured to determine an operation state in which any one of the switching elements connected to each terminal of the rotating machine is turned on.

  According to the above condition, the influence of the priority unit on the determination unit can be reduced as much as possible.

According to an eleventh aspect , in the first or second aspect , the priority unit includes a bias unit that makes the evaluation of the operation state that does not decrease the voltage of the capacitor by the determination unit higher than the evaluation of the operation state that decreases. It is characterized by providing.

In a twelfth aspect based on the eleventh aspect , the bias means includes voltage prediction means for predicting a voltage of a capacitor generated by the operation state for each of the operation states temporarily set by the prediction means. When the voltage predicted by the prediction unit is smaller than a specified value, the evaluation of the operation state that does not decrease the voltage of the capacitor by the determination unit is made higher than the evaluation of the operation state that decreases.

1 is a system configuration diagram according to a first embodiment. FIG. The block diagram regarding the production | generation process of the operation signal concerning the embodiment. The figure which shows the voltage vector expressing the operation state of the inverter concerning the embodiment. The flowchart which shows the process sequence of the model prediction control concerning the embodiment. 6 is a flowchart showing a procedure of counter update processing according to the embodiment. The time chart which shows the effect of the embodiment. The flowchart which shows the process sequence of the model prediction control concerning 2nd Embodiment. The figure which shows the prediction process method of the voltage of the capacitor | condenser concerning the embodiment. The flowchart which shows the process sequence of the model prediction control concerning 3rd Embodiment. The flowchart which shows the process sequence of the model prediction control concerning 4th Embodiment. The block diagram regarding the production | generation process of the operation signal concerning 5th Embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a motor generator 10 mounted on an industrial robot will be described with reference to the drawings.

  FIG. 1 shows the overall configuration of a motor generator control system according to this embodiment. The motor generator 10 as an actuator for an industrial robot is a three-phase permanent magnet synchronous motor. Here, a surface magnet synchronous machine (SPMSM) is assumed as the motor generator 10.

  The motor generator 10 is connected to the MG battery 12 via the inverter INV. The inverter INV includes three sets of series connection bodies of switching elements S * p, S * n (* = u, v, w), and the connection points of these series connection bodies are the U, V, Each is connected to the W phase. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements S * # (* = u, v, w; # = p, n). In addition, a diode D * # is connected in antiparallel to each of these.

  In the present embodiment, the following is provided as detection means for detecting the state of the motor generator 10 and the inverter INV. First, a rotation angle sensor 14 for detecting the rotation angle (electrical angle θ) of the motor generator 10 is provided. Further, a current sensor 16 that detects currents iu, iv, and iw flowing through the phases of the motor generator 10 is provided. Furthermore, a voltage sensor 18 for detecting an input voltage (power supply voltage VDC) of the inverter INV is provided.

  The detection values of the various sensors are taken into the control device 20. The control device 20 generates an operation signal for operating the inverter INV based on the detection values of these various sensors, and outputs the operation signal to the drive unit DU connected to each of the switching elements S * # constituting the inverter INV. Here, the signal for operating the switching element S * # of the inverter INV is the operation signal g * #.

  The drive unit DU drives the switching element to be driven on / off by manipulating the potential difference of the open / close control terminal (gate) with respect to the output terminal (emitter) of the switching element S * # to be driven. This drive unit DU operates using the potential of the output terminal (emitter) of the switching element S * # to be driven as a reference potential. For this reason, the device that is driven by the switching element S * n of the lower arm operates based on the same potential. On the other hand, the operating potential of the switching target S * p of the upper arm varies depending on whether the switching element S * n of the lower arm of the same leg is on or off.

  In view of these points, in the present embodiment, the same drive power supply 22 is used for the drive units DU that are driven by the switching elements S * n of the lower arm. The drive power supply 22 is a DC voltage source having the emitter of the switching element S * n as a reference potential. On the other hand, for the drive unit DU that drives each of the switching elements S * p of the upper arm, capacitors C * (* = u, v, w) connected to the respective emitters of the switching elements S * p are used. Used as a DC voltage source. These capacitors C * are charged by the drive power source 22. That is, each of the capacitors C * can be charged with the electric charge of the drive power supply 22 via the resistor 26 and the diode Db *. Here, the diode Db * is a rectifying unit that prevents the positive charge from flowing from the capacitor C * to the drive power supply 22 and allows the positive charge to be charged from the drive power supply 22 to the capacitor C *. is there.

  According to such a configuration, when the switching element S * n is turned on, the negative electrode potential of the capacitor C * is lowered to the negative electrode potential of the drive power supply 22, so that the charging voltage of the capacitor C * is the terminal of the drive power supply 22. If it is lower than the voltage, the capacitor C * can be charged.

  FIG. 2 shows processing by the control device 20.

  The control device 20 operates the inverter INV so as to control the torque of the motor generator 10 to the required torque Tr. Specifically, the inverter INV is operated so that the command current for realizing the required torque Tr matches the current flowing through the motor generator 10. That is, in the present embodiment, the torque of the motor generator 10 becomes the final control amount. In order to control the torque, the current flowing through the motor generator 10 is directly controlled as a control current. To do. In particular, in the present embodiment, in order to control the current flowing through the motor generator 10 to a command current, the current of the motor generator 10 is predicted and predicted when the operation state of the inverter INV is temporarily set to each of a plurality of types. The temporarily set operation state is evaluated based on the current. Then, model predictive control is performed in which the highly evaluated one is adopted as the actual operation state of the inverter INV.

  Specifically, the phase currents iu, iv, iw detected by the current sensor 16 are converted into real currents id, iq in the rotating coordinate system by the dq converter 30. Further, the electrical angle θ detected by the rotation angle sensor 14 is input to the speed calculation unit 32, whereby the rotation speed (electrical angular speed ω) is calculated. On the other hand, the command current setting unit 34 receives the requested torque Tr and outputs command currents idr and iqr in the dq coordinate system. The command currents idr and iqr, the actual currents id and iq, and the electrical angle θ are input to the model prediction control unit 40. The model prediction control unit 40 determines a voltage vector Vi that defines the operation state of the inverter INV based on these input parameters, and inputs the voltage vector Vi to the operation unit 36. The operation unit 36 generates the operation signal g * # based on the input voltage vector Vi and outputs it to the inverter INV.

  Here, the voltage vectors expressing the operation state of the inverter INV are eight voltage vectors shown in FIG. For example, a voltage vector representing an operation state (indicated as “lower” in the drawing) in which the low-potential side switching elements Sun, Svn, Swn are turned on is the voltage vector V0, and the high-potential side switching elements Sup, A voltage vector representing an operation state (indicated as “upper” in the drawing) in which Svp and Swp are turned on is a voltage vector V7. These voltage vectors V0 and V7 are for short-circuiting all phases of the motor generator 10 and are called zero voltage vectors because the voltage applied to the motor generator 10 from the inverter INV becomes zero. On the other hand, the remaining six voltage vectors V1 to V6 are defined by an operation pattern in which switching elements that are turned on exist in both the upper arm and the lower arm, and are called effective voltage vectors. Yes. As shown in FIG. 3B, the voltage vectors V1, V3, and V5 correspond to the positive sides of the U phase, the V phase, and the W phase, respectively.

  Next, details of the processing of the model prediction control unit 40 will be described. The operation state setting unit 41 shown in FIG. 2 temporarily sets the operation state of the inverter INV. The operation state of the inverter INV temporarily set here is expressed by each of the voltage vectors V0 to V7 shown in FIG. The dq converter 42 calculates the voltage vector Vdq = (vd, vq) of the dq coordinate system by performing dq conversion on the voltage vector set by the operation state setting unit 41. In order to perform such conversion, the voltage vectors V0 to V7 in the operation state setting unit 41 are, for example, “upper” is “VDC / 2” and “lower” is “−VDC / 2” in FIG. It can be expressed as above. In this case, for example, the voltage vector V0 is (−VDC / 2, −VDC / 2, −VDC / 2), and the voltage vector V1 is (VDC / 2, −VDC / 2, −VDC / 2). .

In the prediction unit 43, the current id when the operation state of the inverter INV is set to the state set by the operation state setting unit 41 based on the voltage vector (vd, vq), the actual currents id, iq, and the electrical angular velocity ω. , Iq is predicted. Here, the voltage equation expressed by the following (c1) and (c2) is discretized from the following state equations (formulas (c3) and (c4)) obtained by solving the current differential term. Predict.
vd = (R + pLd) id−ωLqiq (c1)
vq = ωLdid + (R + pLq) iq + ωφ (c2)
pid
= − (R / Ld) id + ω (Lq / Ld) iq + vd / Ld (c3)
piq
= −ω (Ld / Lq) id− (Rd / Lq) iq + vq / Lq−ωφ / Lq (c4)
Incidentally, in the above formulas (c1) and (c2), the resistance R, the differential operator p, the d-axis inductance Ld, the q-axis inductance Lq, and the armature flux linkage constant φ are used.

  The prediction of the current is performed for each of a plurality of operation states temporarily set by the operation state setting unit 41.

  On the other hand, the operation state determination unit 44 determines the operation state of the inverter INV using the predicted currents ide and iq predicted by the prediction unit 43 and the command currents idr and iqr as inputs. Here, each of the operation states set by the operation state setting unit 41 is evaluated by the evaluation function J, and the operation state having the highest evaluation is selected. As this evaluation function J, in the present embodiment, a function whose value increases as the evaluation becomes lower is adopted. Specifically, the evaluation function J is calculated based on the inner product value of the difference between the command current vector Idqr = (idr, iqr) and the predicted current vector Idqe = (ide, iqe). This is a technique for expressing that the evaluation is lower as the value is larger in view of the fact that the deviation of each component between the command current vector Idqr and the predicted current vector Idqe can be both positive and negative values. As a result, it is possible to construct an evaluation function J in which the evaluation becomes lower as the difference between the components of the command current vector Idqr and the predicted current vector Idqe is larger.

  Based on the operation state thus determined, the operation unit 36 generates and outputs an operation signal g * #.

  By the way, according to the model predictive control, since the operation state having the highest evaluation quantified by the evaluation function J is selected, it is guaranteed that the switching element S * n on the low potential side is periodically turned on. Absent. This means that there is no guarantee that the capacitor C * will be charged regularly. If the capacitor C * is not charged and the charging voltage of the capacitor C * decreases, there is a concern that the drive unit DU cannot be driven appropriately.

  Therefore, in this embodiment, such a problem is avoided by performing model predictive control in the following manner.

  FIG. 4 shows a processing procedure of model predictive control according to the present embodiment. This process is repeatedly executed at a predetermined control cycle Tc.

  In this series of processing, first, in step S10, the electrical angle θ (n) and the actual currents id (n) and iq (n) are detected, and the voltage vector V (n) determined in the previous control cycle. Is output. In subsequent step S12, the current (ide (n + 1), iqe (n + 1)) in one control cycle ahead is predicted. This is a process of predicting what will happen to the current one control cycle ahead based on the voltage vector V (n) output in step S10. Here, the currents ide (n + 1) and iqe (n + 1) are calculated by using the model expressed by the above equations (c3) and (c4) discretized by the forward difference method with the control cycle Tc. . At this time, the actual currents id (n) and iq (n) detected in step S10 are used as the initial value of the current, and the voltage vector V (n) is used as the voltage vector on the dq axis. And dq-transformed by using an angle obtained by adding “ωTc / 2” to the electrical angle θ (n) detected in step (b).

  In the subsequent step S14, a counter updating process for quantifying the amount of decrease in the charging voltage of the capacitor C * is performed. FIG. 5 shows details of the processing in step S14. Note that the processing of FIG. 5 is actually executed separately for each of the capacitors Cu, Cv, and Cw.

  In this series of processes, first, in step S30, it is determined whether or not the state in which the capacitor C * cannot be charged continues. In other words, it is determined whether or not the previous S * n switching state determined by the voltage vector V (n−1) is OFF and the current S * n switching state determined by the voltage vector V (n) is OFF. To do. If a positive determination is made in step S30, the discharge counter is subtracted in step S32. Here, the discharge counter has a positive correlation with the charging voltage of the capacitor C *.

  On the other hand, if a negative determination is made in step S30, whether or not the capacitor C * is switched to a chargeable voltage vector in step S34 by the output of the voltage vector V (n) by the process in step S10 of FIG. Judging. In other words, it is determined whether or not the previous switching state of the switching element S * n is off and the current switching state is on. If a positive determination is made in step S34, the charge counter is reset in step S36. Here, the charge counter is a counter that measures the duration of the chargeable state after the capacitor C * can be charged.

  If a negative determination is made in step S34, the process proceeds to step S38. In step S38, it is determined whether or not the chargeable state of capacitor C * continues. In other words, it is determined whether or not the previous switching state of the switching element S * n is on and the current switching state is on. If a positive determination is made in step S38, a charge counter is added in step S40.

  On the other hand, when a negative determination is made in step S38, the process proceeds to step S42. Here, a negative determination is made in step S38 when the capacitor C * is switched to a state where charging is impossible, that is, the previous switching state of the switching element S * n is ON and the current switching state is ON. This is the case when it is off. In this embodiment, in this case, a map operation is performed on the value of the discharge counter corresponding to the charge voltage of the capacitor C * based on the charge counter. Here, the reason why the map value is used without directly using the value of the charge counter is that when the capacitor C * is charged, the relationship between the charging voltage and the charging time has nonlinearity. Specifically, the rate of increase of the charging voltage decreases as the charging time increases. Therefore, a quantitative value (map value) of the voltage of the capacitor C * corresponding to the charging time (charge counter value) is calculated by reflecting this non-linear relationship. Specifically, this map value indicates that charging is started with a voltage equal to or lower than a specified voltage used for determining whether or not a request for forcibly charging the capacitor C * is generated by a process described later as an initial value of the charging voltage. It is set assuming

  In a succeeding step S44, it is determined whether or not the map value is larger than the discharge counter. This process is a setting for determining whether or not to change the discharge counter to a map value. If a positive determination is made in step S44, the discharge counter is changed to a map value in step S46. On the other hand, when a negative determination is made in step S44, such a change is not performed. This is a setting in view of the fact that the above-described map sets the charging voltage of the capacitor C * when the capacitor C * is switched to a chargeable state as a fixed value. That is, in this case, the fixed value of the voltage of the capacitor C * that is mapped can be obtained by quantifying a voltage lower than the actual voltage of the capacitor C *. For this reason, in such a case, the discharge counter is not updated.

  In addition, when the process of said step S32, S36, S40, S46 is completed, it transfers to the process of previous step S16 of FIG.

  In step S16, it is determined whether or not all the discharge counters corresponding to the capacitors C * are larger than a predetermined value. This process is for determining whether or not the charging voltage of the capacitor C * is excessively decreased. Here, the predetermined value is a threshold required to forcibly charge the capacitor C *. In particular, the predetermined value is set to a value corresponding to a voltage value equal to or higher than the initial value of the charging voltage assumed in the map setting in the process of FIG.

  If a negative determination is made in step S16, that is, if it is determined that there is a predetermined value or less in the discharge counter corresponding to each capacitor C *, the charging voltage of the capacitor C * decreases excessively in step S17. As a result, the forced charge flag is turned on.

  When the process of step S17 is completed or when an affirmative determination is made in step S16, the evaluation function J of each voltage vector V0 to V7 is calculated in steps S18 and S20. That is, until the evaluation function J is calculated for all of the voltage vectors V0 to V7 (step S18: YES), the current of two control cycles ahead is predicted for each of the cases where the voltage vector in the next control cycle is temporarily set. Process. Here, in step S20, first, a voltage vector V (n + 1) in the next control cycle is temporarily set, and predicted currents ide (n + 2) and iqe (n + 2) are calculated in the same manner as in step S12. However, here, the predicted currents ide (n + 1) and iqe (n + 1) calculated in step S12 are used as the initial value of the current, and the voltage vector V (n + 1) is used as the voltage vector on the dq axis. What dq-transformed by the angle which added "3 (omega) Tc / 2" to the electrical angle (theta) (n) detected in step S10 is used. Then, the evaluation function J is calculated for each of these predicted currents ide (n + 2) and iqe (n + 2).

  In the subsequent step S22, if the forced charging flag is turned on and the operation state that minimizes the evaluation function J is adopted, whether or not the capacitor C * that has caused the negative determination of step S16 is charged. Judge whether or not. If a negative determination is made in step S22, the voltage vector V (n + 1) in the next control cycle is determined as a voltage vector that minimizes the evaluation function J in step S24. On the other hand, when an affirmative determination is made in step S22, in step S26, the voltage vector V (n + 1) is determined to be the voltage vector V0, and the forced charge flag is turned off.

  When the processes in steps S24 and S26 are completed, variables n, n + 1, etc. are decremented in step S28. In addition, when the process of step S28 is completed, this series of processes is once complete | finished.

  FIG. 6A shows the transition of the voltage of the capacitor Cu according to this embodiment. On the other hand, FIG. 6B shows the transition of the voltage of the capacitor Cu by the process of determining the operation state of the inverter INV only by the evaluation function J. As shown in the figure, in the present embodiment, it is possible to reliably avoid that the voltage of the capacitor Cu becomes lower than a threshold at which the operation of the drive unit DU cannot be guaranteed.

  When the voltage vector V0 is forcibly adopted as described above, this may increase the difference between the current flowing through the motor generator 10 and the command currents idr and iqr. However, since the processing for forcibly adopting the voltage vector V0 is basically performed at a low rotational speed, this deviation can be ignored. That is, at a low rotational speed, the modulation rate of the inverter INV is small. On the other hand, the change speed vector of the error edq in the dq coordinate system is expressed by the difference between the fundamental component of the output voltage of the inverter INV and the voltage vector (instantaneous voltage) currently employed. Here, when the zero voltage vector is employed, the difference becomes the fundamental wave component itself. When the modulation factor is small, the fundamental wave component is small compared to the instantaneous voltage. Therefore, the change speed of the error edq when the zero voltage vector is used is compared with the case where an effective voltage vector that can charge the capacitor C * is used. And become smaller.

  It should be noted that the process of forcibly adopting the voltage vector V0 is basically performed at the low rotation speed when the rotation speed is high and the time required for one rotation of the control cycle Tc and the electrical angle. This is because when the difference is reduced, an operation state in which the switching element S * n is turned on in a short time is easily adopted. That is, even in the model predictive control, it is difficult to make a selection that causes the switching element of the lower arm not to be continuously turned on in the period of one electrical angle cycle.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) Under the condition that the charging voltage of the capacitor C * decreases, the operation state for charging the capacitor C * is selected regardless of the evaluation function J. Thereby, the situation where the voltage of the capacitor C * is excessively reduced can be avoided.

  (2) Decreasing the charging voltage of the capacitor C * based on both the period in which the high potential side switching element S * p is in the on state and the period in which the low potential side switching element S * n is in the on state. I figured out. As a result, the voltage drop of the capacitor C * can be grasped with high accuracy.

  (3) The value of the discharge counter can be corrected by the value calculated by the map from the value of the charge counter each time the capacitor C * is switched to a state where it cannot be charged. Thereby, the voltage of the capacitor C * can be quantified with high accuracy by the discharge counter.

  (4) The voltage vector V0 was forcibly selected under the condition that the charging voltage of the capacitor C * was lowered. Thereby, the process for forcibly charging the capacitor C * can be simplified. Also, in the region where the modulation factor is low, the change rate of the error edq is smaller at the zero voltage vector than at the effective voltage vector, so that the control amount error caused by the forced charging process is suppressed. You can also.

(5) The voltage vector V0 is forcibly selected on the condition that the operation state having the highest evaluation using the evaluation function J is an operation state in which the capacitor C * that is desired to be charged cannot be charged. Thereby, the operation state that minimizes the error edq can be used as much as possible.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, the evaluation function J is changed so that the evaluation of the operation state in which the capacitor C * is charged becomes high under the situation where the charging voltage of the capacitor C * is lowered.

  FIG. 7 shows a processing procedure of model predictive control according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle. Note that, in FIG. 7, the same step numbers are attached for convenience to those corresponding to the processing shown in FIG.

  In this series of processes, in step S50, for each voltage vector temporarily set as the voltage vector V (n + 1), a process for predicting the voltage Vc * of the capacitor C * is performed. Hereinafter, the prediction process will be described with reference to FIG.

In the present embodiment, as shown in FIG. 8, the voltage Vc * is predicted using a model in which a resistor 50 having a resistance value R2 is connected in parallel to the capacitor C * as the discharge path of the capacitor C *. Here, the resistance value R1 of the resistor 26, the capacitance C of the capacitor C *, and the voltage Ed of the drive power supply 22 are used. When the switching element S * n is in the ON state, the charge of the drive power supply 22 is Since the capacitor C * is charged and discharged through the resistor 50, the following equation (c5) is established.
Vc *
= R2 · Ed · {1−exp (−t / τc)} / (R1 + R2) + Vc * 0 · exp (−t / τc)
... (c5)
However, the initial value Vc * 0 of the voltage of the capacitor C * before one control cycle Tc and the time constant τc expressed by the following equation (c6) are used.
τc = R1 · R2 · C / (R1 + R2) (c6)
On the other hand, when the switching element S * p is in the ON state, the change in the voltage of the capacitor C * is caused only by the discharge through the resistor 50, so the following equation (c7) is established.
Vc * = Vc * 0 · exp (−t / τd) (c7)
However, the time constant τd shown below is used.
τd = R2 · C (c8)
By using the above equations (c5) and (c7), it is possible to predict the voltage Vc * associated with the use of the temporarily set voltage vector V (n + 1). In particular, in step S50, the voltage at the time when one control cycle Tc has elapsed from the time when the temporarily set voltage vector V (n + 1) is adopted is estimated. This can be done by setting the voltage Vc * predicted in the previous step S50 to the initial value Vc * 0 and the parameter t to the control period Tc. However, the voltage Vc * predicted in the previous step S50 is assumed to correspond to the voltage vector V (n) output in step S10.

  In subsequent step S20a, an evaluation function J relating to the temporarily set voltage vector V (n + 1) is calculated. In this embodiment, the evaluation function J is added to the vector norm of the error edq by adding the larger of “Vref−Vc *” and zero for all phases of the u, v, and w phases. calculate. Here, “Vref−Vc *” is a setting for lowering the evaluation of the operation state in which this situation is left under the situation where the voltage of the capacitor C * is lowered. Here, the reference voltage Vref is a parameter indicating a situation in which the voltage of the capacitor C * decreases. The reference voltage Vref is set to a value larger than the upper limit value at which the drive unit DU cannot be driven normally.

  When the evaluation function J is calculated for all the voltage vectors V0 to V7, the process proceeds to step S52, and the voltage vector corresponding to the operation state that minimizes the evaluation function J is determined as the next voltage vector V (n + 1). .

  According to the embodiment described above, the following effects can be obtained.

(6) The evaluation of the operation state that does not decrease the voltage of the capacitor C * is made higher than the evaluation of the operation state that decreases. Thereby, both controllability of the control amount of the motor generator 10 and controllability of the voltage of the capacitor C * can be achieved.
<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 9 shows a processing procedure of model predictive control according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle. In FIG. 9, the same step numbers are assigned for convenience to those corresponding to the processing shown in FIG.

  In this series of processes, after the process of step S12, instead of the process of step S14, the voltage Vc * of the capacitor C * is estimated in step S60. Here, the estimation process may be a prediction process of the voltage Vc * of the capacitor C * ahead of one control cycle Tc due to the output of the voltage vector V (n), but the capacitor C at the time when the process of step S10 is performed. It may be an estimation of the voltage of *. Here, “estimation” is used as a term including “prediction”. This estimation process can be performed in the same manner as the prediction process in the second embodiment.

In a succeeding step S62, it is determined whether or not all the estimated voltages Vc * are larger than the threshold value Vth. This process corresponds to the process in step S16 of FIG. If a negative determination is made in step S62, the process proceeds to step S17. Then, when the process of step S17 is completed or when an affirmative determination is made in step S62, the process proceeds to step S18.
<Fourth Embodiment>
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 10 shows a processing procedure of model predictive control according to the present embodiment. This process is repeatedly executed, for example, at a predetermined cycle. In FIG. 10, the same step numbers are assigned for convenience to those corresponding to the processing shown in FIG. 4.

In this series of processes, after the process of step S12, instead of the process of step S14, the voltage Vc * of the capacitor C * is detected in step S70. This can be done by providing means for detecting the voltage across each of the capacitors C *. In a succeeding step S72, it is determined whether or not all the estimated voltages Vc * are larger than the threshold value Vth. This process corresponds to the process in step S16 of FIG. If a negative determination is made in step S72, the process proceeds to step S17. Then, when the process of step S17 is completed or when an affirmative determination is made in step S72, the process proceeds to step S18.
<Fifth Embodiment>
Hereinafter, a fifth embodiment will be described with reference to the drawings, focusing on differences from the first embodiment.

  In the present embodiment, torque and magnetic flux are directly controlled variables, and these command values and predicted values are input to determine the operation state of the inverter INV.

  FIG. 11 shows processing by the control device 20 according to the present embodiment. In FIG. 11, processes corresponding to the processes shown in FIG. 2 are given the same reference numerals for convenience.

  As illustrated, the torque / magnetic flux prediction unit 60 predicts the magnetic flux vector Φ and the torque T of the motor generator 10 based on the predicted currents ide and iqe. Here, the magnetic flux vector Φ = (Φd, Φq) is predicted by the following equations (c9) and (c10), and the torque T is predicted by the following equation (c11).

Φd = Ld · id + φ (c9)
Φq = Lq · iq (c10)
T = P (Φd · iq−Φq · id) (c11)
Incidentally, in the above formula (c11), the number P of pole pairs is used.

  On the other hand, in the magnetic flux map 62, the command magnetic flux vector Φr is set based on the required torque Tr. Here, the command magnetic flux vector Φr is set according to a request for realizing, for example, maximum torque control that can obtain the maximum torque with the minimum current among those satisfying the required torque Tr.

The operation state determination unit 44a quantifies the degree of deviation between the control amount and the command value based on the difference between the predicted torque Te and the required torque Tr and the difference between the components of the predicted magnetic flux vector Φe and the command magnetic flux vector Φr. A voltage vector that minimizes the degree of deviation is employed. The quantification here is determined based on the sum of values obtained by multiplying the squares of these differences by weighting coefficients a and b (a ≠ b, a ≠ 0, b ≠ 0). Here, the weighting factors a and b are taken into consideration that the magnitudes of torque and magnetic flux are different. That is, for example, when setting a unit in which the numerical value of the torque is larger, the torque deviation tends to be larger. Therefore, when the weighting factors a and b are not used, the voltage vector has a low magnetic flux controllability. However, there is a possibility that disadvantages such as that the evaluation does not become so low. For this reason, the weight coefficients a and b are used as means for compensating for the difference in absolute value of the plurality of input parameters for evaluation.
<Other embodiments>
Each of the above embodiments may be modified as follows.

"About update means"
In the first embodiment, when the value of the discharge counter is larger than the map value, the value of the discharge counter may be increased by a predetermined ratio of the map value.

About counters
In the first embodiment, the amount of change per control cycle Tc of the discharge counter or the charge counter may be variably set according to the current counter value or the like instead of a fixed value. Thereby, it can be simulated that the rate of voltage increase due to charging of the capacitor C * or the rate of voltage decrease due to discharge changes with time.

  In the first embodiment, a process of adding instead of the process of subtracting the discharge counter may be employed, and a process of subtracting may be employed instead of the process of adding the charge counter. In this case, since the discharge counter has a negative correlation with the charge voltage of the capacitor C *, the amount of decrease in the charge voltage of the capacitor C * has reached the upper limit based on the large value of the discharge counter. Just judge.

  Only a discharge counter may be provided without providing a charge counter. In this case, however, the discharge counter is reset when the forced charging process is performed.

  The present invention is not limited to one provided for each terminal of motor generator 10. For example, the discharge counter may be added by the number of phases in which the high potential side switching element S * p is turned on. Even in this case, since the value of the discharge counter and the voltage of the capacitor C * are considered to have a negative correlation, it is possible to select the voltage vector V0 based on the fact that the value of the discharge counter reaches a predetermined value. It is valid.

"About estimation means"
The model formula indicating the time transition of the voltage of the capacitor C * is not limited to that exemplified in the second embodiment. For example, a voltage drop due to charging of the gate of switching element S * # and a voltage drop due to power consumed by the circuit in drive unit DU may be modeled separately. For example, the model shown in FIG. 8 is used as a means for calculating the voltage drop due to the power consumed by the circuit in the drive unit DU, and when the switching element S * p is turned on, This can be realized by reducing the calculated voltage by a predetermined amount.

"Voltage prediction means"
The fact that the model formula for prediction is not limited to that exemplified in the second embodiment is as described in the section “About estimation means”.

  Further, the present invention is not limited to using a model formula, and a parameter having a positive or negative correlation with the voltage of the capacitor C * is calculated, for example, a discharge counter value realized by the cooperation of the discharge counter and the charge counter. It may be a means.

"About bias means"
This is not limited to adding (Vref−Ve *) to the evaluation function J on condition that the predicted voltage Ve * is smaller than the reference voltage Vref. For example, (Vref−Ve *) may be added to the evaluation function J on condition that the amount that the predicted voltage Ve * exceeds the reference voltage Vref is equal to or less than a specified amount. For example, the square of (Vref−Ve *) may be added to the evaluation function J on condition that the predicted voltage Ve * is smaller than the reference voltage Vref.

  The evaluation is not limited to one in which the evaluation is changed in multiple stages according to the degree to which the voltage Vc * with respect to the capacitor C * becomes smaller than the specified value. For example, when voltage Vc * becomes smaller than a specified value, capacitor C * is charged with a predetermined amount (> 0) smaller than the maximum value that can be taken as the norm of error edq vector when motor generator 10 is normally controlled. It may be added only to the evaluation function J of the operation state that is not performed.

  Further, the present invention is not limited to operating the evaluation at the time of determination by the determination means when the voltage Vc * is smaller than the specified value. For example, the evaluation of the operation state where the capacitor C * is not charged at all times may be lower than the evaluation of the operation state where the capacitor C * is charged.

"About preferred means"
In the first embodiment (step S22 in FIG. 4), it is not necessary to provide a condition that the operation state that minimizes the evaluation function J does not charge the capacitor C * whose discharge counter is equal to or less than a predetermined value.

  Further, for example, in the first embodiment, instead of forcibly setting the voltage vector V0, it may be forcibly changed to an operation state in which the capacitor C * whose discharge counter is equal to or less than a predetermined value is charged.

“Temporarily set operation status”
For example, the number of switching terminals in the switching state of the motor generator 10 may be “1” or less, or “2” or less.

About prediction means
It is not limited to predicting only the control amount generated by the next voltage vector V (n + 1). For example, the control amount by the operation of the inverter INV at the update timing several control cycles ahead may be sequentially predicted.

About the decision means
For example, in the first embodiment, the weighted average processing value of the difference between the predicted current ide (n + 2) and the command current idr (n + 2) and the difference between the predicted current iqe (n + 2) and the command current iqr (n + 2). May be used as a parameter for evaluation of the degree of deviation. In short, in order to quantify that the evaluation becomes lower as the degree of divergence is larger, it may be quantified by a parameter having a positive or negative correlation with the degree of divergence.

"About controlled variables"
The control amount used for determining the operation of the inverter INV based on the command value and the predicted value is not limited to any of torque, magnetic flux, and current. For example, only torque or magnetic flux may be used. For example, torque and current may be used. Here, when the control amount is other than the current, the direct detection target by the sensor may be other than the current.

  In each of the above embodiments, the ultimate control amount of the rotating machine (the control amount that is ultimately required to be a desired amount regardless of whether or not it is a prediction target) is the torque. For example, the rotational speed may be used.

"Switching elements that make up a DC / AC converter circuit"
For example, a field effect transistor such as a MOS field effect transistor may be used instead of the IGBT. Here, not only the N channel but also the P channel may be used. However, since the P-channel MOS field effect transistor is turned on / off by manipulating the potential of the open / close control terminal (gate) with respect to the input terminal (source), the switching element S * p on the high potential side The power source is a drive power source 22, and the power source of the switching element S * n on the low potential side is a capacitor C *.

"About rotating machines"
It is not limited to a three-phase rotating machine. For example, it may be a four-phase or more rotating machine such as a five-phase rotating machine in which the windings of the five terminals are connected to each other.
In each of the above embodiments, it is assumed that the stator windings connected to each terminal of the rotating machine are star-connected. However, the present invention is not limited to this, and may be delta-connected.

  The rotating machine is not limited to a surface magnet synchronous machine, and may be an arbitrary synchronous machine such as an embedded magnet synchronous machine or a field winding type synchronous machine. Furthermore, it is not limited to a synchronous machine, but may be an induction rotating machine such as an induction motor.

  The rotating machine is not limited to that mounted on an industrial robot. For example, it may be used as a main machine of a vehicle.

"others"
The DC voltage source is not limited to the MG battery 12 and may be, for example, an output terminal of a converter that boosts the voltage of the MG battery 12.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 12 ... MG battery (one embodiment of DC voltage source), 20 ... Control device (one embodiment of control device of rotating machine), 22 ... Drive power supply, Cu, Cv, Cw ... Capacitor.

Claims (3)

A DC / AC converter circuit including a switching element that selectively connects a terminal of a rotating machine to each of a positive electrode and a negative electrode of a DC voltage source, by turning on / off the switching element of the DC / AC converter circuit. In a control device for a rotating machine that controls a control amount having at least one of a flowing current, a torque of the rotating machine, and a magnetic flux of the rotating machine,
The drive circuit for driving the switching element includes a DC voltage source having a reference potential at one of a pair of ends of a current flow path of one of the switching element connected to the positive electrode and the switching element connected to the negative electrode And a bootstrap circuit that charges a capacitor serving as a power source for the other drive circuit during any one of the on periods,
Predicting means for predicting the control amount when temporarily setting the operation state of the DC-AC converter circuit represented by a voltage vector in a fixed coordinate system;
A determination unit that evaluates an operation state corresponding to the predicted control amount based on the control amount predicted by the prediction unit, and determines a highly evaluated operation state as an operation state of the DC-AC converter circuit;
Operating means for operating the DC-AC converter circuit so as to be in the determined operating state;
In a situation where the charging voltage of the capacitor is lowered, when determining the operation state by the determination unit, priority means for increasing the priority of the operation state in which either one is turned on;
Equipped with a,
The control device for a rotating machine, wherein the priority unit includes a bias unit that makes the evaluation of the operation state that does not decrease the voltage of the capacitor by the determination unit higher than the evaluation of the operation state that reduces the voltage . The bias unit includes a voltage prediction unit that predicts a voltage of a capacitor generated by the operation state for each operation state temporarily set by the prediction unit, and the voltage predicted by the voltage prediction unit is lower than a specified value. smaller case, the evaluation of the operating state not to lower the voltage of said capacitor by said determining means, the control device for a rotary machine according to claim 1, wherein the higher than rated operating conditions to reduce. The priority unit is configured to charge the battery based on both a period in which one of the switching elements connected to the same terminal of the rotating machine is in an on state and a period in which the other switching element is in an on state. 3. The control device for a rotating machine according to claim 1, wherein the lowering of the voltage is grasped and the priority processing is performed based on the decrease.