JP5619225B1 - Control device for synchronous motor - Google Patents

Abstract

In a synchronous motor having no magnetic saliency or low magnetic saliency, the control of the motor is stopped or the motor control is stopped when the estimation accuracy of the magnetic pole position deteriorates due to the influence of a disturbance or the like. The estimation is performed again so that sufficient accuracy can be obtained. A control apparatus for a synchronous motor applies a bias current to a motor 107 to temporarily estimate a magnetic pole position θ1, and passes a bias current with the magnetic pole position θ1 as an initial phase. , A second magnetic pole position estimator 1101 that estimates the magnetic pole position θ2 by converging the feature quantity related to the phase difference between the bias current phase and the magnetic pole position to a predetermined value by the convergence calculation, and the magnetic pole position θ1 and the magnetic pole position θ2 And a drive control unit 1514 that controls the drive of the motor 107 based on the difference between them. [Selection] FIG.

Description

  Embodiments described herein relate generally to a control device for a synchronous motor used in, for example, an elevator hoist.

  In general, in a control device for a synchronous motor, the rotation angle is measured using a rotation angle sensor, and the synchronous motor is driven by supplying a current synchronized with the rotation angle. On the other hand, due to problems such as cost, installation space, and reliability, a “sensorless control” technique that does not use a rotation angle sensor has been developed.

  One of the techniques is a technique for estimating the magnetic pole position based on a high-frequency current when a high-frequency voltage is applied to the synchronous motor by using the magnetic saliency of the rotor, especially when the synchronous motor is stopped and at a low speed, and There is a technique for discriminating the polarity of a magnetic pole using a magnetic saturation phenomenon.

  Here, the magnetic saliency and the magnetic saturation phenomenon will be briefly described.

  The presence / absence of magnetic saliency depends on whether or not the magnetic flux easily passes. Magnets are materials that do not easily pass magnetic flux. On the other hand, the magnetic flux easily passes through the iron core surrounding the magnet. The ease of passing the magnetic flux is expressed by the size of the inductance L. That is, it is difficult for magnetic flux to pass in a certain direction of the magnet, and the value of the inductance L is small. On the other hand, the magnetic flux easily passes in the direction in which the iron core is present, and the value of inductance L is large. Such a difference in the passage of magnetic flux depending on the direction is called “magnetic saliency”.

  In the case of “with magnetic saliency”, since there is an electrical characteristic even when the motor is rotating or at a low speed, the magnetic pole position can be estimated based on the inductance LdLq of the dq axis.

  “Magnetic saturation phenomenon” refers to a phenomenon in which magnetic flux is too dense in a magnetic material, making it difficult for the magnetic flux to flow. A magnet is originally a material that is difficult to pass magnetic flux. Therefore, in the case of an embedded magnet type motor, if the magnetic flux caused by the current flowing through the stator coil is made to go around the rotor and the ease of passing the magnetic flux is detected from the inductance L, the magnetic flux generated by the magnet and the stator coil are When the generated magnetic flux is in the same direction, the ease of passing the magnetic flux, that is, the inductance L is minimized. By utilizing this, the magnetic pole position can be estimated without a sensor. Note that “estimating (detecting) the magnetic pole position” has the same meaning as “estimating (detecting) the rotation angle”.

  There is also a control system using a PG (Pulse Generator) as a rotation angle sensor. In this control system, the rotation angle is calculated by counting pulses output from the PG in accordance with the rotation of the motor. However, only the displacement is known by counting the pulses. In order to obtain the absolute angle, it is usually necessary to detect a Z pulse that is output only once per rotation, and thus there is a problem that control is difficult.

  In order to solve such a problem, there is a technique of estimating the magnetic pole position by applying the sensorless control technique described above and setting it as the initial position of PG.

  In addition, in a permanent magnet synchronous motor, in SPMSM (Surface Permanent Magnet Synchronous Motor) in which the rotor has no magnetic saliency, in principle, the magnetic pole position is estimated using the magnetic saliency. It becomes impossible. Therefore, there is a technique that realizes estimation of the magnetic pole position even in SPMSM by utilizing the magnetic saturation phenomenon.

  This is because when a positive and negative voltage vector having a predetermined direction and magnitude is given, magnetic saturation occurs only in one of the positive and negative voltage vectors due to the current flowing through each voltage vector, and the current peak value is obtained. The magnetic pole position is estimated by utilizing the occurrence of the difference.

Japanese Patent No. 3312472 Japanese Patent No. 3401155 Japanese Patent No. 3213751 Japanese Patent No. 4211133

  However, in the above technique, since the magnetic saturation is generated only by applying the voltage vector, the current value cannot be accurately controlled. Therefore, the magnetic saturation phenomenon may not always be used accurately. For example, if the voltage vector is too large or the output time width is too long, the current increases too much, and magnetic saturation occurs in both positive and negative voltage vectors. For this reason, since a clear difference is difficult to appear in the current peak value, the magnetic pole position cannot be estimated accurately.

  Conversely, if the voltage vector is too small or the time width is too short, the current remains small, and no magnetic saturation occurs with either positive or negative voltage vector. Therefore, no clear difference appears in the current peak value as described above, and the magnetic pole position cannot be accurately estimated.

  In particular, when the motor changes, the motor constant and magnetic saturation characteristics change. For this reason, it is necessary to investigate and reset the optimum voltage vector size and output time, which is time consuming and does not necessarily result in an optimum current value due to manufacturing errors.

  The problem to be solved by the present invention is to control a motor in a synchronous motor having no magnetic saliency or having a low magnetic saliency when the estimation accuracy of the magnetic pole position deteriorates due to the influence of a disturbance or the like. It is an object of the present invention to provide a control apparatus for a synchronous motor that is stopped or re-estimated so as to obtain sufficient accuracy for controlling the motor.

  The control apparatus for a synchronous motor according to the present embodiment allows a bias current having a predetermined magnitude and a phase change from 0 to 360 degrees to flow through the synchronous motor so that a current change occurs near the current operating point of the bias current. Based on the applied high-frequency voltage and the high-frequency current flowing corresponding to the high-frequency voltage, an inductance equivalent value in the same phase direction as the bias current is obtained, and the bias current phase at which the inductance equivalent value is an extreme value is determined as the first value. A first magnetic pole position estimating means that estimates the magnetic pole position of the first magnetic pole position, and a bias current of a predetermined magnitude is passed with the first magnetic pole position estimated by the first magnetic pole position estimating means as an initial phase. Bias current phase based on the high-frequency voltage applied so that a current change occurs near the operating point and the high-frequency current flowing in response to the high-frequency voltage A feature value related to the phase difference from the magnetic pole position is calculated, a convergence calculation is performed to correct the bias current phase so that the feature value converges to a predetermined value, and the bias current phase obtained as a result of the convergence is calculated as the second magnetic pole position. Second magnetic pole position estimating means for estimating the second magnetic pole position as the magnetic pole position of the synchronous motor, the first magnetic pole position estimated by the first magnetic pole position estimating means and the second magnetic pole position Drive control means for controlling the drive of the synchronous motor based on the difference from the second magnetic pole position estimated by the second magnetic pole position estimation means.

FIG. 1 is a block diagram showing a configuration relating to a first magnetic pole position estimation unit of the control apparatus for a synchronous motor according to the first embodiment. FIG. 2 is a view for explaining the definition of the coordinate system of the synchronous motor in the embodiment. FIG. 3 is a waveform diagram of a high-frequency voltage command of the synchronous motor in the same embodiment. FIG. 4A and FIG. 4B are waveform diagrams of a bias current phase angle command for the synchronous motor in the same embodiment. FIG. 5 is a view for explaining the definition of the bias current of the synchronous motor in the embodiment. FIG. 6A and FIG. 6B are diagrams showing examples of response waveforms of the bias current of the synchronous motor in the same embodiment. FIG. 7 is a flowchart showing a first magnetic pole position estimation process performed by the synchronous motor control apparatus according to the embodiment. FIG. 8 is a flowchart showing a second magnetic pole position estimation process by the synchronous motor control apparatus according to the embodiment. FIG. 9 is a cross-sectional view schematically showing the configuration of the SPMSM as the synchronous motor in the embodiment. FIG. 10 is a diagram showing magnetic flux characteristics with respect to the bias current of the synchronous motor in the same embodiment. FIG. 11 is a diagram showing the relationship between the phase angle of the bias current and the inductance of the synchronous motor in the same embodiment. FIG. 12 is a block diagram showing a configuration relating to a second magnetic pole position estimation unit of the control apparatus for the synchronous motor in the same embodiment. FIG. 13 is a view for explaining the definition of the coordinate system of the synchronous motor in the embodiment. FIG. 14A and FIG. 14B are diagrams showing characteristics of the characteristic amount of the synchronous motor in the same embodiment. FIG. 15 is a block diagram showing the configuration of the convergence calculation by the second magnetic pole position estimation unit of the synchronous motor control apparatus according to the embodiment. FIG. 16 is a block diagram illustrating a configuration of a magnetic pole position comparison portion of the synchronous motor control device according to the embodiment. FIG. 17 is a block diagram illustrating a configuration of a repetitive mechanism of magnetic pole position estimation processing of the synchronous motor control device according to the second embodiment. FIG. 18 is a flowchart showing a first magnetic pole position estimation process performed by the synchronous motor control apparatus according to the second embodiment. FIG. 19 is a flowchart showing a second magnetic pole position estimation process performed by the synchronous motor control apparatus according to the second embodiment. FIG. 20 is a flowchart showing a first magnetic pole position estimation process performed by the synchronous motor control apparatus according to the third embodiment. FIG. 21 is a flowchart showing a second magnetic pole position estimation process performed by the synchronous motor control apparatus according to the third embodiment. FIG. 22 is a diagram for explaining the disturbance of the characteristic amount characteristic in the third embodiment.

  Hereinafter, embodiments will be described with reference to the drawings.

(First embodiment)
The control apparatus for a synchronous motor in the present embodiment includes a first magnetic pole position estimation unit and a second magnetic pole position estimation unit. Below, it explains in detail divided into (a) the configuration of the first magnetic pole position estimation unit and (b) the configuration of the second magnetic pole position estimation unit. For convenience, the configuration of the first magnetic pole position estimation unit (FIG. 1) and the configuration of the second magnetic pole position estimation unit (FIG. 12) will be described separately, but they are provided in the same control device. .

  As will be described later, the first magnetic pole position estimation unit provisionally estimates the magnetic pole position θ1 by applying a bias current to the synchronous motor. The second magnetic pole position estimation unit causes a bias current to flow with the magnetic pole position θ1 estimated by the first magnetic pole position estimation unit as an initial phase, and features related to the phase difference between the bias current phase and the magnetic pole position by convergence calculation Is converged to a predetermined value to estimate the magnetic pole position θ2. This magnetic pole position θ2 is finally detected as an accurate position.

(A) Configuration of First Magnetic Pole Position Estimation Unit First, the configuration of the first magnetic pole position estimation unit will be described.

  FIG. 1 is a block diagram showing a configuration relating to a first magnetic pole position estimation unit of the control apparatus for a synchronous motor according to the first embodiment.

  An inverter 105 is provided as motor driving means. This inverter 105 receives a gate command for driving the inverter from the triangular wave PWM modulation unit 104 as input, and switches AC / DC power to each other by switching ON / OFF of a main circuit switching element (not shown) so that the motor 107 Drive.

  The motor 107 is an electric motor such as SM (synchronous motor) or PMSM (permanent magnet synchronous motor) that excites a rotating magnetic field in the stator in synchronization with the rotation of the rotor. A rotating magnetic field is generated by a three-phase alternating current flowing in each excitation phase of the UVW of the motor 107, and rotating torque is output by magnetic interaction with the rotor.

  Hereinafter, PMSM will be described as an example. Even when SM is used, the same model can be applied because only the difference in the rotor field is made by a permanent magnet or a field coil on the model.

  Current detectors 106 a and 106 b detect a two-phase or three-phase current response value of the three-phase AC current flowing through motor 107. The example of FIG. 1 shows a configuration for detecting a two-phase current. The brake 108 fixes the rotation of the rotor of the motor 107.

  The magnetic pole position detection unit 109 detects the rotation angle of the rotor (rotor) using a rotation angle sensor such as an encoder. The coordinate conversion unit (αβ / UVW) 103a and the coordinate conversion unit (UVW / αβ) 103b perform coordinate conversion between a three-phase fixed coordinate system and a fixed coordinate system having two orthogonal axes (αβ axes).

Here, the definition of the coordinate system of the synchronous motor in this embodiment is shown in FIG.
Stator coils 201a, 201b, and 201c are U-phase, V-phase, and W-phase coils of the stator, respectively. The αβ-axis fixed coordinate system 202 is a coordinate system in which the α-axis coincides with the U-phase direction and the β-axis advances 90 degrees in phase. The dq-axis rotating coordinate system 203 is a coordinate system in which the d-axis coincides with the direction of the magnetic pole position of the rotor 204 and the q-axis advances 90 degrees in phase. The phase difference between the α axis and the d axis is the rotation angle θ _m .

In FIG. 1, a conversion process in a coordinate conversion unit (αβ / UVW) 103a that performs coordinate conversion of the voltage command value is shown in Equation 1. Expression 2 shows a conversion process in the coordinate conversion unit (UVW / αβ) 103b for converting the current response value.

The current control unit 102 compares the current response values i _α and i _β with the current command values i _αref and i _βref to determine the voltage command values v _α and v _β . A general calculation process in the current control unit 102 is expressed by Equation 3 using a PI controller.

  Here, Kp and Ki are a proportional gain and an integral gain, respectively, and s is a Laplace operator.

  Triangular wave PWM modulation unit 104 modulates a three-phase voltage command value for driving motor 107 with triangular wave PWM, and outputs a gate signal that is an ON / OFF command for each phase switching element of inverter 105.

The high frequency voltage superimposing unit 113 calculates, for example, high frequency voltage commands v _{α_hf} and v _{β_hf} having a constant amplitude waveform shown in FIG. The current control outputs v _α0 and v _β0 are added to the high frequency voltage commands v _{α_hf} and v _{β_hf} to generate new voltage commands v _α and v _β .

The high-frequency voltage commands v _{α_hf} and v _{β_hf in} FIG. 3 can be expressed as in Expression 4.

Here, ω _hf is a high-frequency angular frequency, and H _v is a high-frequency amplitude, both of which are preset values. t is time. H _v is preferably set to a value that high-frequency current flowing through the high-frequency voltage is sufficiently small amplitude as compared with the bias current command value to be described later. This is because if the bias current becomes too small due to the high frequency current, the magnetic saturation phenomenon of the motor is alleviated and the magnetic pole position cannot be estimated accurately.

The bias current command generation unit 101 receives the preset amplitude command I _bias and the phase angle command value θ _bias which is the output of the bias current phase angle command generation unit 112 as inputs, and generates current commands i _αref and i _βref by Equation 5. To do.

The bias current phase angle command generator 112 outputs a phase angle command value _θbias that changes continuously or intermittently over time. For example, as shown in FIG. 4 (a), 0 to 360 ° may be output continuously, or may be output intermittently as shown in FIG. 4 (b). At this time, the change speed of the phase angle is set to an angular frequency sufficiently lower than the angular frequency ω _hf in Equation 4. In other words, the period T _bias is set to be sufficiently longer than the period T _hf illustrated in FIG.

FIG. 5 is a diagram showing a spatial arrangement of bias current commands in the αβ axis fixed coordinate system.
The high frequency voltage generated by the high frequency voltage superimposing unit 113 generates a high frequency current whose operating point moves slightly in the vicinity of the current operating point shown in FIG.

FIG. 6 is a diagram illustrating an example of a response waveform of a bias current that is changed in phase with a predetermined amplitude and is supplied to the motor 107 by current control.
Since the high frequency voltage is superimposed, in the entire waveform shown in FIG. 6 (a), i _α and i _β are thick waveforms, but in the partially enlarged waveform diagram shown in FIG. 6 (b). The high-frequency waveform has a period T _hf . As described above, the change in the phase angle of the bias current is sufficiently long with respect to the cycle of the high frequency, so that the high frequency component is hardly affected by the rotation of the bias current.

The high-frequency current detection unit 110 detects high-frequency currents i _{α_hf} and i _{β_hf} having the same frequency as the high-frequency voltage using a general high-pass filter or band-pass filter. The cut-off frequency of the filter is set to a high frequency that can remove at least a change component of the bias current. In the case of a bandpass filter, the cutoff frequency on the high frequency side is set to a frequency that cuts detection noise in the current detection units 106a and 106b.

The first magnetic pole position estimation unit 111 calculates an inductance equivalent value L _bias in the bias current direction. That is, the first magnetic pole position estimating unit 111 calculates the component i _{hf_bias} of the bias current direction θ _bias detected by the high frequency current detecting unit 110 as _shown in Equation 6, and calculates the amplitude value I _{hf_bias} of this component. measure. For example, the amplitude value may be measured by storing the maximum value and the minimum value appearing during the period _Thhf and calculating the difference.

  Here, ψ is a phase difference with respect to the superimposed high-frequency voltage.

Subsequently, the first magnetic pole position estimation unit 111 calculates an inductance equivalent value L _bias as shown in Expression 7 from the amplitude value I _{hf_bias} .

If a high frequency voltage amplitude _{H v} and the angular frequency omega _hf a constant value, an inductance equivalent value _{L bias} is inversely proportional to the high-frequency current amplitude _{I hf_bias.} Therefore, I _{hf_bias} can be regarded as an inductance equivalent value. Similarly, if the time change rate in the vicinity of the zero crossing of the high-frequency current expressed by Equation 6 is measured or calculated, it becomes a value corresponding to I _{hf_bias} . Therefore, this time change rate may be set as an inductance equivalent value.

Further, the first magnetic pole position estimation unit 111, an inductance equivalent value _{L bias} between a bias current direction theta _bias is changed from 0 to 360 degrees and outputs a bias current direction theta _{Bias_min} as a minimum value. When I _{hf_bias} is an inductance equivalent value, the bias current direction in which this is the maximum value is θ _{bias_min} . That is, the bias current direction in which the calculated inductance equivalent value exhibits an extreme value may be set to θ bias — _min . This θ _{bias_min} is the pole magnetic position of the motor 107.

  7 and 8 are flowcharts showing the magnetic pole position estimation processing in the first embodiment. Among these, steps A11 to A19 in FIG. 7 show the magnetic pole position estimation processing by the first magnetic pole position estimation unit 111.

That is, in the control device having the configuration shown in FIG. 1, first, current control for the motor 107 is started through the current control unit 102 (step A12) with initial values of current and voltage set (step A11). In addition, the control device starts superimposing the high frequency voltage through the high frequency voltage superimposing unit 113 (step A13).
When a bias current command value is input to the control device (step A14), a bias current having a predetermined phase is supplied to the motor 107, a high frequency voltage is applied to the bias current, and a high frequency current is observed at the same time. become able to.

  The control device detects a high-frequency current flowing corresponding to this high-frequency voltage (step A15), and obtains an inductance equivalent value in the direction of the same phase as the bias current through the first magnetic pole position estimation unit 111 (step A16). Specifically, based on the high-frequency voltage applied so that a current change occurs near the current operating point of the bias current and the high-frequency current flowing corresponding to this high-frequency voltage, the inductance equivalent value in the same phase direction as the bias current Ask for.

  While changing the bias current from 0 to 360 degrees, the control device sequentially obtains inductance equivalent values in the same phase direction as the bias current at that time (steps A17 to A18). As a result, the bias current phase where the inductance equivalent value finally becomes an extreme value is set as the magnetic pole position of the motor 107 (step A19). The magnetic pole position θ1 obtained at this time is set as the initial rotation angle.

Next, the operation of the above configuration will be described.
First, the occurrence of the magnetic saturation phenomenon will be described.

  FIG. 9 is a cross-sectional view schematically showing the configuration of the SPMSM. In this example, a 4-pole surface magnet type PMSM (SPMSM) is shown, in which 801 is a stator and 802 is a rotor. Further, the dq axis in the figure indicates a coordinate system that coincides with the magnetic pole of the rotor.

  The magnetic flux of the bias current schematically represents the magnetic flux generated by the bias current when the phase of the bias current is changed in the state of FIG. In FIG. 9, when the magnetic flux direction and the bias current magnetic flux are in the same direction (1), magnetic saturation occurs and the inductance in the bias current direction decreases. In the states (2) and (3), there is no magnet magnetic flux or a reverse direction to the magnetic flux of the bias current, so magnetic saturation does not occur and inductance does not decrease.

  FIG. 10 shows typical characteristics of the amplitude of the bias current and the magnetic flux in the bias current direction when the bias current phase angle is set to three patterns of the d-axis positive direction, the q-axis direction, and the d-axis negative direction.

  In FIG. 10, the gradient of the magnetic flux with respect to the amplitude of the bias current is the inductance. If the bias current is increased in the positive direction, magnetic saturation occurs and the inclination becomes smaller. This characteristic varies depending on the direction of the bias current phase with respect to the magnetic pole, that is, the d-axis. This shows that when the bias current phase with respect to the d axis is 0 degrees, that is, when the bias current phase is in the positive direction of the d axis, magnetic saturation starts with the smallest bias current.

  On the other hand, even when a bias current is passed in the negative direction of the d-axis, magnetic saturation occurs if the amplitude of the bias current is increased. For this reason, it can be seen that the difference in inductance due to the difference in the phase of the bias current becomes smaller at this bias current amplitude.

  That is, when the bias current amplitude value is small (for example, operating point a in FIG. 9), magnetic saturation does not occur at any phase angle and no inductance difference appears. When the bias current amplitude value is large (operating point c), magnetic saturation occurs at any phase, and the difference in inductance does not appear.

  When estimating the magnetic pole position using magnetic saturation, it is important to accurately pass a bias current in which a difference in inductance clearly appears. In FIG. 10, the current value in the range of the shaded portion including the operating point b Is appropriate.

  Further, regarding the magnetic saturation phenomenon, the d-axis reference bias current phase angle and the inductance in the bias current direction are shown in correspondence with each other as shown in FIG.

  The bias current operating points a, b, and c in FIG. 11 coincide with the operating points in FIG. That is, in the characteristics at the bias current operating point b, when the bias current phase angle and the phase difference between the d-axis are near zero, the magnetic flux and the magnetic flux due to the bias current are in the same direction, so that the inductance is reduced due to magnetic saturation. At the operating points a and c, the difference in inductance hardly appears due to the reasons described above, and the phase angle at which the inductance takes an extreme value due to detection noise or the like cannot be accurately estimated.

  As described above, the magnetic pole position is estimated using the high-frequency current amplitude in the same direction by utilizing the fact that the inductance in the bias current direction changes due to the magnetic saturation phenomenon. That is, the phase angle at which the bias current phase angle coincides with the d-axis is calculated by calculating the inductance equivalent value from the amplitude and frequency of the high-frequency voltage flowing at least in the same direction as the bias current due to the superimposed high-frequency voltage. Can be measured.

(B) Configuration of Second Magnetic Pole Position Estimation Unit Next, the configuration of the second magnetic pole position estimation unit will be described.

  FIG. 12 is a block diagram showing a configuration relating to a second magnetic pole position estimation unit of the control device for the synchronous motor. The same parts as those in FIG. 1 are denoted by the same reference numerals, and detailed description thereof will be omitted. Although not shown in the drawings, it is assumed that the components shown in FIG. 1 are actually included in the configuration of FIG.

  A coordinate conversion unit (st / UVW) 1102a and a coordinate conversion unit (UVW / st) 1102b are provided instead of the coordinate conversion unit (αβ / UVW) 103a and the coordinate conversion unit (UVW / αβ) 103b in FIG. In addition to the first magnetic pole position estimation unit 111, a second magnetic pole position estimation unit 1101 is provided.

Although not shown in FIG. 12, the first magnetic pole position estimation unit 111 is biased based on the high-frequency currents i _{α_hf} and i _{β_hf} detected by the high-frequency current detection unit 110 as described in FIG. An inductance equivalent value _Lbias in the current direction is calculated. Further, the first magnetic pole position estimation unit 111, an inductance equivalent value _{L bias} between a bias current direction theta _bias is changed from 0 to 360 degrees and outputs a bias current direction theta _{Bias_min} as a minimum value.

In the configuration of FIG. 12, a coordinate conversion unit (st / UVW) 1102a and a coordinate conversion unit (UVW / st) 1102b perform coordinate conversion between a three-phase fixed coordinate system and an orthogonal two-axis (st axis) rotational coordinate system. As shown in FIG. 13, the st-axis rotation coordinate system 1201 is a coordinate system that rotates at a bias current phase angle θ _bias with the α axis as a reference.

The processes in the coordinate conversion units 1102a and 1102b are shown in Expression 8 and Expression 9.

In addition, as a current command that is an input to the current control unit 102, a bias current amplitude command _Ibias is directly input as an s-axis current command.

Second magnetic pole position estimation unit 1101, based on the high-frequency voltage command _{v S_hf} and _{v T_hf} and the high-frequency current _{i S_hf} and _{i T_hf} detected by the high-frequency detecting unit 110, associated with the phase difference Δθ of the bias current and the magnetic pole position, wherein The quantity R is calculated.

Ideally, the characteristic amount R is ideally obtained as Δθ itself, but it is sufficient that an amount substantially proportional to Δθ is obtained in the vicinity of zero of Δθ. FIG. 14 shows such characteristic amount characteristics. In FIG. 14A, the characteristic amount is zero at the point of Δθ = 0, but the characteristic amount may have an offset. In FIG. 14B, the offset value R _ofs at the point of Δθ = 0 is provided. In this case, a target value for convergence calculation described later may be set to the offset value R _ofs .

  For example, when the high frequency voltage is applied only in the same direction as the bias current, the feature amount R can be obtained as the feature amount R by measuring the amplitude of the high frequency current that appears in the direction orthogonal to the bias current.

That is, the high frequency current shown in Equation 10 is given.

At this time, the high-frequency current that appears in the direction orthogonal to the bias current is a t-axis high-frequency current and has the characteristics of Equation 11.

Here, L _sat is a coefficient that fluctuates due to magnetic saturation that occurs when the bias current is in the vicinity of the magnetic pole position. When the bias current is not near the magnetic pole position and magnetic saturation does not occur, L _sat = 0, and when magnetic saturation occurs, it has a value other than zero.

From the high-frequency current of the formula 11, when extracting the amplitude component of the cos (ω _{hf t),} the amplitude is a function of sin (2.DELTA..theta), since a characteristic which is substantially proportional to [Delta] [theta] in [Delta] [theta] = 0 near, used as the feature quantity be able to.

  As described above, the configuration in which the high-frequency voltage is applied only in the same direction as the bias current has an advantage that electromagnetic noise due to the generation of the high-frequency current can be suppressed to be smaller than that in the other configurations.

Subsequently, the second magnetic pole position estimation unit 1101 performs a convergence operation based on the feature amount R and outputs a bias current phase θ _{bias_set} .

  FIG. 15 is a block diagram showing the configuration of the convergence calculation in the second magnetic pole position estimation unit 1101.

Gain multipliers 1401 and 1402 multiply predetermined gains K _{p_pll} and K _{i_pll} , respectively. Integrators 1403 and 1404 integrate the respective inputs over time. Regarding the initial integration value, the integrator 1403 has an initial value of zero, and the integrator 1404 has an initial value of θ _{bias_min} .

θ _{bias_min} is the magnetic pole position estimated by the first magnetic pole position estimation unit 111 in FIG. By setting this as an initial value, the bias current phase θ _{bias_set} starts to converge from θ _{bias_min,} so the second magnetic pole position estimation unit 1101 is calculated from the phase near the magnetic pole position estimated by the first magnetic pole position estimation unit 111. The estimation process can be started.

The convergence determination unit 1405 terminates the convergence calculation based on whether the difference between the feature values R and R _ofs is a certain value or less, whether the time during which the convergence calculation is performed has elapsed for a certain period of time, or other criteria. Judgment is made and θ _{bias_set} at that time is output as the final estimated magnetic pole position θ _{bias_end} . Examples of the other determination criteria include a time during which a difference between feature amounts is continuously below a certain value elapses for a predetermined time.

When configured as in FIG. 15, it is possible to modify the theta _{Bias_set} to match the characteristic amount R to the offset amount _{R ofs} This convergence calculation. Note that R _ofs may be set to zero in accordance with the characteristics of the feature amount R.

  By using the second magnetic pole position estimation unit 1101, the estimated magnetic pole position can be estimated by being converged to the zero cross point of the feature amount R and Δθ. Therefore, it is possible to estimate the magnetic pole position with higher accuracy than the first magnetic pole position estimation unit 111 and other methods.

  In the flowchart of FIG. 8, steps B <b> 11 to B <b> 19 indicate magnetic pole position estimation processing by the second magnetic pole position estimation unit 1101.

That is, in the control device having the configuration shown in FIG. 12, first, with the magnetic pole position θ1 obtained by the first magnetic pole position estimating unit 111 set as an initial value (step B11), the motor 107 is controlled through the current control unit 102. Current control is started (step B12). Further, the control device starts superimposing the high frequency voltage through the high frequency voltage superimposing unit 113 (step B13).
Here, when the bias current command value is input to the control device (step B14), a bias current having a predetermined phase is supplied to the motor 107, a high frequency voltage is applied to the bias current, and the high frequency current is observed at the same time. become able to.

  The control device detects the high-frequency current flowing corresponding to this high-frequency voltage (step B15), and obtains the bias current phase angle by the convergence calculation through the second magnetic pole position estimation unit 1101 (steps B16 to B18). Specifically, a high-frequency voltage applied so that a bias current having a predetermined magnitude flows with the magnetic pole position θ1 as an initial phase and a current change occurs in the vicinity of the operating point of the bias current, and a high-frequency current flowing corresponding to the high-frequency voltage. Based on the above, a feature amount related to the phase difference between the bias current phase and the magnetic pole position is calculated, and a convergence calculation is performed to correct the bias current phase so that the feature amount converges to a predetermined value.

  The control device sets the bias current phase finally obtained from the result of the convergence calculation as the magnetic pole position θ2 of the motor 107 (step B19).

(Comparison between the first magnetic pole position and the second magnetic pole position)
FIG. 16 is a block diagram showing a configuration of a magnetic pole position comparison portion of the synchronous motor control device. The same parts as those in FIGS. 1 and 12 are denoted by the same reference numerals, and detailed description thereof will be omitted. Although omitted from the drawings, it is assumed that the components shown in FIGS. 1 and 12 are actually included in the configuration of FIG.

  The control apparatus for a synchronous motor in this embodiment includes a first magnetic pole position estimation unit 111 shown in FIG. 1 and a second magnetic pole position estimation unit 1101 shown in FIG. Further, the control device includes a speed control current command generation unit 1511, a magnetic pole position comparison unit 1512, an external storage unit 1513, and a drive control unit 1514 as configurations related to the two estimation units.

Speed control current command generation unit 1511 outputs to the current control unit 102 calculates a current command i _Arufaref and i _Betaref such as speed of the motor 107 coincides with the speed command S _c.

  The magnetic pole position comparison unit 1512 generates a magnetic pole position difference Δθ based on the temporary pole position θ1 output from the first magnetic pole position estimation unit 111 and the magnetic pole position θ2 output from the second magnetic pole position estimation unit 1101. Is calculated as shown in Equation 12.

Δθ = | θ1-θ2 | (Formula 12)
The magnetic pole position comparison unit 1512 compares the magnetic pole position difference Δθ with the magnetic pole position difference threshold θth stored in the external storage unit 1513. The drive control unit 1514 controls driving of the motor 107 based on the comparison result of the magnetic pole position comparison unit 1512.

Here, both θ1 and θ2 are magnetic pole positions detected based on a decrease in inductance caused by magnetic saturation. In principle, Δθ is zero because the same angle is indicated. However, for example, when the current detection units 106a and 106b are affected by external devices and disturbances occur in the current response values i _u and i _w , the first magnetic pole position estimation unit 111 has a true extreme value. Other points may be detected as the magnetic pole position θ1. This is because the first magnetic pole position estimation unit 111 inductance equivalent value _{L bias} or high-frequency current amplitude _{Ihf_bias} is performing operations such as an extreme value.

  In this case, the second magnetic pole position estimation unit 1101 that performs the convergence calculation using θ1 as the initial phase requires a longer time than usual in order to converge the phase to the true magnetic pole position. For example, when the end condition of the convergence calculation is a certain time, the detected magnetic pole position θ2 is far away from θ1, and it is considered that the true magnetic pole position is not detected.

When θ2 is controlling the motor 107 in a state not true magnetic pole position (erroneous magnetic pole position), occurs a phenomenon such as not satisfied abnormal noise or occurs, or the speed command S _c. In such a case, the magnetic pole position difference Δθ calculated in Equation 12 is a value other than zero.

Therefore, as shown in the flowchart of FIG. 8, when the magnetic pole position difference Δθ <threshold θth is not satisfied, that is, when the difference Δθ between θ1 and θ2 is greater than or equal to the preset threshold θth (No in Step B20), The position comparison unit 1512 outputs a control inhibition command C _inh to the drive control unit 1514. When the control prohibition command C _inh is input, the drive control unit 1514 prohibits the control of the motor 107 by cutting off the rotation command _Crot to the speed control current command generation unit 1511 (step B21).

On the other hand, when the magnetic pole position difference Δθ <threshold θth is satisfied, that is, when the difference Δθ between θ1 and θ2 is smaller than the preset threshold θth (Yes in Step B20), the magnetic pole position comparison unit 1512 sends the drive control unit 1514. On the other hand, the control prohibition command C _inh is not output. As a result, the rotation command C _rot is output from the drive control unit 1514 to the speed control current command generation unit 1511, and the control of the motor 107 is started (step B22).

  As described above, according to the first embodiment, by comparing the magnetic pole position θ1 and the magnetic pole position θ2, the estimation accuracy of the magnetic pole position can be further increased, and the motor 107 can be normally driven and controlled. If the correct magnetic pole position cannot be detected due to the influence, the motor 107 can be safely stopped.

(Second Embodiment)
Next, a second embodiment will be described.

In the first embodiment, the difference Δθ between the magnetic pole position θ1 obtained by the first magnetic pole position estimator 111 and the magnetic pole position θ2 obtained by the second magnetic pole position estimator 1101 is equal to the threshold _θth . If it exceeds, the control of the motor 107 is immediately prohibited. In contrast, in the second embodiment, when the difference Δθ between the magnetic pole position θ1 and the magnetic pole position θ2 is greater than or equal to the threshold _θth , the magnetic pole position estimation process is repeated a predetermined number of times.

  FIG. 17 is a block diagram illustrating a configuration of a repetitive mechanism of magnetic pole position estimation processing of the synchronous motor control device according to the second embodiment. The same parts as those in FIG. 16 are denoted by the same reference numerals, and detailed description thereof will be omitted. Although not shown in the drawings, it is assumed that the components shown in FIGS. 1 and 12 are actually included in the configuration of FIG.

  The control apparatus for a synchronous motor in the present embodiment includes an external storage unit 1515 and a repetition determination unit 1516.

The external storage unit 1515 stores the number of times Nrp that is a criterion for repeatedly determining the magnetic pole position estimation process. In the example of FIG. 17, the external storage unit 1513 and the external storage unit 1515 are provided independently, but the threshold value θth and the number of times Nrp may be stored in one storage unit. When the control prohibition command C _inh1 is output from the magnetic pole position comparison unit 1512, the repetition determination unit 1516 determines whether or not to re-execute the magnetic pole position estimation process based on the number of times Nrp.

  18 and 19 are flowcharts showing the magnetic pole position estimation processing in the second embodiment. Among these, steps A11 to A19 in FIG. 18 show the magnetic pole position estimation processing by the first magnetic pole position estimation unit 111. Steps B11 to B19 in FIG. 19 show the magnetic pole position estimation processing by the second magnetic pole position estimation unit 1101.

Here, in the second embodiment, when the magnetic pole position difference Δθ <threshold θth is not satisfied, that is, when the difference Δθ between θ1 and θ2 is equal to or larger than the preset threshold θth (No in step B20), the magnetic pole position The comparison unit 1512 outputs a control prohibition command C _inh1 to the repetition determination unit 1516. Repetition determination unit 1516, when this control prohibition command C _inh1 is input, determines whether the number of repetitions of the magnetic pole position estimation process has reached the Nrpth (step B23).

  If it has not reached the Nrpth time (No in Step B23), as described in the first embodiment, the magnetic pole position estimation processing by the first magnetic pole position estimation unit 111 and the magnetic pole by the second magnetic pole position estimation unit 1101 The position estimation process is executed again. As a result, when the magnetic pole position difference Δθ <threshold θth is not satisfied, the first and second magnetic pole position estimation processes are repeated up to Nrp times.

  When the magnetic pole position difference Δθ <threshold θth is satisfied in this repetition. That is, if the difference Δθ between θ1 and θ2 becomes smaller than the preset threshold θth (Yes in Step B20), the repeat determination unit 1516 does not output the control prohibition command Cinh2 to the drive control unit 1514, and the drive control unit 1514 The rotation command Crot is output to the current command generator 1511 for speed control, and the control of the motor 107 is started (step B22).

On the other hand, when the magnetic pole position difference Δθ <threshold θth is not satisfied in the repetition, and the number of times the magnetic pole position estimation is repeated reaches the Nrpth, the control prohibition command C _inh2 is output from the repetition determination unit 1516 to the drive control unit 1514. The When the control prohibition command C _inh is input, the drive control unit 1514 prohibits the control of the motor 107 by cutting off the rotation command _Crot to the speed control current command generation unit 1511 (step B21).

  As described above, according to the second embodiment, the magnetic pole position can be correctly detected by repeating the first and second magnetic pole position estimation processes according to the comparison result between the magnetic pole position θ1 and the magnetic pole position θ2. For example, it is possible to avoid a situation in which the operation of the motor 107 is stopped because the magnetic pole position cannot be detected due to the influence of disturbance or the like.

(Third embodiment)
Next, a third embodiment will be described.

  In the third embodiment, when the magnetic pole position cannot be detected by the first and second magnetic pole position estimation processes, the magnetic pole position θ1 is corrected and the second magnetic pole position estimation process is executed again. It is.

  The apparatus configuration is the same as that of FIG. 17 in the second embodiment. Here, with reference to FIG. 20 and FIG. 21, only the process as the third embodiment will be described.

  20 and 21 are flowcharts showing the magnetic pole position estimation processing in the third embodiment. Among these, steps A11 to A19 in FIG. 20 show the magnetic pole position estimation processing by the first magnetic pole position estimation unit 111. Steps B11 to B19 in FIG. 21 show the magnetic pole position estimation processing by the second magnetic pole position estimation unit 1101.

If the magnetic pole position difference Δθ <threshold θth is not satisfied, that is, if the difference Δθ between θ1 and θ2 is greater than or equal to the preset threshold θth (No in step B20), the magnetic pole position comparison unit 1512 proceeds to the repetition determination unit 1516. A control inhibition command C _inh1 is output. Repetition determination unit 1516, when this control prohibition command C _inh1 is input, determines whether the number of repetitions of the magnetic pole position estimation process has reached the Nrpth (step B23).

  Here, in the third embodiment, when the number of repetitions of the magnetic pole position estimation process has not reached the Nrpth time (No in Step B23), the repetition determination unit 1516 corrects the magnetic pole position θ1 by the following Expression 13. (Step B24), the corrected magnetic pole position θ1 ′ is set as an initial value in the second magnetic pole position estimation unit 1101 (Step B25).

θ1 ′ = θ2 + sign (θ1−θ2) * θo (Formula 13)
θ1 is the magnetic pole position obtained by the first magnetic pole position estimation unit 111, θ2 is the magnetic pole position obtained by the second magnetic pole position estimation unit 1101, θo is a predetermined value such as 30 degrees, and sign is a value in parentheses. This function returns 1 if is positive, and returns (-1) if it is negative. The second magnetic pole position estimation unit 1101 executes the second magnetic pole position estimation process again using the corrected magnetic pole position θ1 ′ obtained by Equation 13 as an initial value.

  As described above, when the magnetic pole position difference Δθ <threshold θth is not satisfied, the second magnetic pole position estimation process is performed again using the corrected magnetic pole position θ1 ′, so that the motor 107 and the inverter 105 to be controlled are performed. Thus, it is possible to avoid erroneous estimation of the magnetic pole position due to the disturbance of the feature amount generated in the characteristic.

  That is, when Δθ does not show the ideal feature as shown in FIG. 14 and deviates from the ideal value at a cycle of 60 ° as shown in FIG. 22, for example, the true magnetic pole position is set to 0 °, and θth = It is assumed that 15 ° and θo = 30 ° are set.

Here, if θ1 = 12 ° and θ2 = 28 ° are obtained by the first first magnetic pole position estimation process and the second magnetic pole position estimation process,
Δθ = | θ1-θ2 | = 16 °
It becomes.

  That is, since the phase difference Δθ is larger than the threshold θth = 15 °, the second magnetic pole position estimation process is performed again. At this time, the corrected magnetic pole position θ1 ′ has the following value according to the above equation 13.

θ1 ′ = 28 ° + (− 1) * 30 °
= -2 °
With this θ1 ′ as an initial value, the second magnetic pole position estimation process is performed. In this case, the first magnetic pole position estimation process starts from a position of 12 °, but the second time starts from a position of −2 °. As a result, the waveform of the phase difference Δθ shown in FIG. 22 is also shifted, so that it is possible to avoid obtaining a convergence result (feature amount zero) at a singular point of 30 °.

  As a result, if the magnetic pole position difference Δθ <threshold θth is not satisfied, the second magnetic pole position estimation is repeated again. In this repetition, if the magnetic pole position difference Δθ <threshold θth is satisfied, that is, if the difference Δθ between θ1 and θ2 becomes smaller than the preset threshold θth, the repeat determination unit 1516 prohibits the drive control unit 1514 from controlling. The drive control unit 1514 outputs the rotation command Crot to the speed control current command generation unit 1511 without outputting the command Cinh2, and the control of the motor 107 is started.

On the other hand, when the magnetic pole position difference Δθ <threshold θth is not satisfied in the repetition, and the number of times the magnetic pole position estimation is repeated reaches the Nrpth, the control prohibition command C _inh2 is output from the repetition determination unit 1516 to the drive control unit 1514. The When this control prohibit command C _inh is input, the drive control unit 1514 prohibits the control of the motor 107 by blocking the rotation command C _rot with respect to the speed control current command generation unit 1511.

  As described above, according to the third embodiment, the motor 107 is corrected by correcting the magnetic pole position θ1 according to the comparison result between the magnetic pole position θ1 and the magnetic pole position θ2 and executing the second magnetic pole position estimation process again. In addition, it is possible to avoid erroneous estimation of the magnetic pole position due to the disturbance of the characteristic amount generated by the characteristics of the inverter 105 and to detect the correct magnetic pole position.

  It is also possible to combine the second embodiment and the third embodiment. For example, if the second magnetic pole position estimation process is repeated twice in the third embodiment, the second magnetic pole position estimation process is repeated twice in the second embodiment, and the second magnetic pole position estimation process is repeated twice. . When this repetition is performed three times, the drive of the motor 107 is stopped. As a result, it is possible to further strengthen the erroneous estimation of the magnetic pole position and the avoidance of stopping the motor control.

  In particular, in a synchronous motor used for an elevator hoisting machine, it is necessary to avoid abnormal noise due to incorrect estimation of the magnetic pole position, and operation service is interrupted due to stoppage of motor control. Therefore, the method of the present embodiment that can detect the magnetic pole position and start the motor control is effective.

  According to at least one embodiment described above, in a synchronous motor that does not have magnetic saliency or has low magnetic saliency, when the estimation accuracy of the magnetic pole position deteriorates due to the influence of disturbance or the like, It is possible to provide a control apparatus for a synchronous motor in which control is stopped or estimation is performed again so as to obtain sufficient accuracy for controlling the motor.

  In addition, although some embodiment of this invention was described, these embodiment is shown as an example and is not intending limiting the range of invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.

  DESCRIPTION OF SYMBOLS 101 ... Bias current command generation part, 102 ... Current control part, 103a ... Coordinate conversion part ((alpha) (beta) / UVW), 103b ... Coordinate conversion part (UVW / (alpha) (beta)), 104 ... Triangular wave PWM modulation part, 105 ... Inverter, 106a and 106b DESCRIPTION OF SYMBOLS ... Current detection part, 107 ... Motor, 108 ... Brake, 109 ... Magnetic pole position detection part, 110 ... High frequency current detection part, 111 ... First magnetic pole position estimation part, 112 ... Bias current phase angle command generation part, 113 ... High frequency Voltage superimposing section, 201a, 201b, 201c ... stator coil, 202 ... αβ axis fixed coordinate system, 203 ... dq axis rotary coordinate system, 204 ... rotor, 801 ... stator, 802 ... rotor, 1101 ... second magnetic pole position Estimating unit, 1102a ... coordinate conversion unit (st / UVW), 1102b ... coordinate conversion unit (UVW / st), 1201 ... st axis rotation seat System, 1401 and 1402 ... Gain multiplier, 1403 and 1404 ... Integrator, 1405 ... Convergence determination unit, 1511 ... Current command generation unit for speed control, 1512 ... Magnetic pole position comparison unit, 1513 ... External storage unit, 1514 ... Drive control Part, 1515... External storage part, 1516.

Claims (6)

A bias current having a predetermined magnitude and a phase change from 0 to 360 degrees is supplied to the synchronous motor, and a high-frequency voltage applied so that a current change occurs near the current operating point of the bias current and the high-frequency voltage The first magnetic pole position estimation is performed to obtain an inductance equivalent value in the direction of the same phase as the bias current based on the high-frequency current flowing and to estimate the bias current phase at which the inductance equivalent value is an extreme value as the first magnetic pole position. Means,
The first magnetic pole position estimated by the first magnetic pole position estimating means is used as an initial phase, a bias current of a predetermined magnitude is passed, and a high frequency voltage applied so that a current change occurs near the operating point of the bias current. And a feature value related to the phase difference between the bias current phase and the magnetic pole position based on the high-frequency current flowing corresponding to the high-frequency voltage, and the bias current phase is corrected so that the feature value converges to a predetermined value. A second magnetic pole position estimating means for performing a convergence calculation, estimating a bias current phase obtained as a result of the convergence as a second magnetic pole position, and determining the second magnetic pole position as a magnetic pole position of the synchronous motor;
Drive for controlling the drive of the synchronous motor based on the difference between the first magnetic pole position estimated by the first magnetic pole position estimating means and the second magnetic pole position estimated by the second magnetic pole position estimating means And a control means for the synchronous motor. The drive control means includes
When the difference between the first magnetic pole position and the second magnetic pole position is smaller than a preset threshold value, the control of the synchronous motor is started, and the first magnetic pole position and the second magnetic pole position are 2. The synchronous motor control device according to claim 1, wherein control of the synchronous motor is prohibited when the difference between the two is equal to or greater than the threshold value. The drive control means includes
When the difference between the first magnetic pole position and the second magnetic pole position is greater than or equal to a preset threshold value, the magnetic pole position estimation processing by the first and second magnetic pole position estimation means is repeatedly executed. The control apparatus for a synchronous motor according to claim 1, wherein: The drive control means includes
The magnetic pole position estimation process by the first and second magnetic pole position estimating means is repeatedly executed up to a preset number of times, and the first magnetic pole position and the second magnetic pole position are reached even when the number of times is reached. 4. The synchronous motor control device according to claim 3, wherein control of the synchronous motor is prohibited when a difference between the synchronous motor and the motor is greater than or equal to the threshold value. The drive control means includes
When the difference between the first magnetic pole position and the second magnetic pole position is greater than or equal to a preset threshold value, the second magnetic pole position is corrected by adding a predetermined value, and the corrected magnetic pole position is initialized. 2. The synchronous motor control device according to claim 1, wherein the magnetic pole position estimation processing by the second magnetic pole position estimation means is repeatedly executed as the position. The drive control means includes
The magnetic pole position estimation process by the second magnetic pole position estimating means is repeatedly executed up to a preset number of times, and the difference between the first magnetic pole position and the second magnetic pole position even when the number of times is reached. 6. The control apparatus for a synchronous motor according to claim 5, wherein control of the synchronous motor is prohibited when the value is equal to or greater than the threshold value.

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