JP5434943B2 - Rotation sensor - Google Patents

Description

  The present invention relates to a rotation sensor that calculates a relative rotation angle of a measurement object that rotates relative to a signal output from a magnetoelectric conversion element.

  Conventionally, as a technique related to the rotation sensor that calculates the relative rotation angle as described above, for example, a digital tracking R / D converter disclosed in Patent Document 1 below is known. In this converter, a sine wave signal and a cosine wave signal from the rotation detector according to the rotation of the rotor that is the object of rotation detection, specifically, rotation detection signals sinθ · f (t), cosθ · f (t). Is inputted, the first output signal sin (θ−φ) · f (t) is obtained based on sinφ and cosφ corresponding to the digital angle output φ inputted as feedback. The first output signal sin (θ−φ) · f (t) is quantized (digital signal) by positive / negative determination by a comparator and then synchronously detected, so that the control deviation ε = sin (θ -Φ) to obtain the digital angle output φ. Thereby, the R / D conversion processing can be digitized, and the conversion performance can be improved (stability, high speed, noise resistance).

JP 2000-353957 A

  By the way, in the rotation sensor as disclosed in the above-mentioned Patent Document 1, due to a structural error such as a shape error or a layout error in a measurement object or a magnetoelectric conversion element, a phase shift between a sine wave signal and a cosine wave signal, etc. May occur. Such a structural error is different for each sensor, and there is a problem that the measurement accuracy of the relative rotation angle is lowered in a sensor having a large structural error.

  The present invention has been made to solve the above-described problems, and the object of the present invention is to provide a rotation that can more accurately calculate the relative rotation angle while suppressing the occurrence of measurement errors due to structural errors and the like. It is to provide a sensor.

In order to achieve the above object, in the rotation sensor according to claim 1, the signal level is N periods (where N is a natural number) according to the strength of the magnetic field during one rotation of the relative rotating measurement object. And a plurality of magnetoelectric transducers arranged so as to output a phase difference between the sine wave signal and the cosine wave signal. In a rotation sensor configured to obtain a relative rotation angle with respect to the measurement object using the sine wave signal and the cosine wave signal output from an element, the sine wave signal output from the plurality of magnetoelectric conversion elements and Using the cosine wave signal, feedback control is performed so that the deviation between the relative rotation angle θ with respect to the measurement object and the calculation angle φ obtained by calculation converges to a predetermined value, and the relative rotation is performed. An angle calculation unit that calculates a rotation angle θ, and an output unit that outputs a signal corresponding to the calculation angle φ calculated by the angle calculation unit, and relatively rotates the relative rotation angle θ to be a reference angle. Further, for the measurement object, a deviation from the reference angle included in the sine wave signal is measured in advance as an angle α based on a signal output as the sine wave signal from each magnetoelectric transducer, and the cosine wave Based on a signal output as a signal, a deviation from the reference angle included in the cosine wave signal is measured in advance as an angle β, and the angle calculation unit is configured to output the sine wave output from the plurality of magnetoelectric transducers. Subtract the signal sin (Nθ + α) multiplied by cos (Nφ + β) from the cosine wave signal cos (Nθ + β) multiplied by sin (Nφ + α). For edges, is the variable portion (Nθ-Nφ) is performed a feedback control such that the predetermined value, characterized by calculating the relative rotation angle theta.
sin (Nθ + α) × cos (Nφ + β) −cos (Nθ + β) × sin (Nφ + α)
= (Sinαsinβ + cosαcosβ) × sin (Nθ−Nφ)

  According to a second aspect of the present invention, in the rotation sensor according to the first aspect, a first addition unit that outputs an angle obtained by adding the angle β to an input angle, and an angle that is input from the first addition unit Cosine value output means for outputting a cosine value in accordance with the cos (Nφ + β) is obtained by inputting a calculation angle Nφ to the first addition means to obtain a cosine value output from the cosine value output means. It is calculated based on this.

  According to a third aspect of the present invention, in the rotation sensor according to the first or second aspect, the second addition means for outputting an angle obtained by adding the angle α to the input angle and the second addition means are input. Sine value output means for outputting a sine value corresponding to the angle to be output, and sin (Nφ + α) is a sine output from the sine value output means by inputting a calculation angle Nφ to the second addition means. It is calculated based on a value.

According to a fourth aspect of the present invention, in the rotation sensor according to any one of the first to third aspects, the sine wave signal with respect to a reference amplitude based on a signal output as the sine wave signal from each of the magnetoelectric transducers. The amplitude ratio of the cosine wave signal is previously measured as the first amplitude ratio A, and the amplitude ratio of the cosine wave signal to the reference amplitude is previously measured as the second amplitude ratio B based on the signal output as the cosine wave signal. Then, the angle calculating unit multiplies the cosine wave signal Bcos (Nθ + β) by Asin (Nφ + α) by multiplying the sine wave signal Asin (Nθ + α) output from the plurality of magnetoelectric transducers by Bcos (Nφ + β). Feedback control is performed so that the variable part (Nθ−Nφ) is the predetermined value for the right side of the following formula obtained by solving the left side of the following formula to subtract the product. Characterized by calculating the relative rotation angle theta.
Asin (Nθ + α) × Bcos (Nφ + β) −Bcos (Nθ + β) × Asin (Nφ + α)
= AB (sinαsinβ + cosαcosβ) × sin (Nθ−Nφ)

  According to the first aspect of the present invention, the sine wave signal is included in the measurement object that is relatively rotated so that the relative rotation angle θ becomes the reference angle, based on a signal output as a sine wave signal from each magnetoelectric transducer. The deviation from the reference angle is measured in advance as the angle α, and the deviation from the reference angle included in the cosine wave signal is measured in advance as the angle β based on the signal output as the cosine wave signal. Then, the angle calculation unit multiplies the cosine wave signal cos (Nθ + β) by sin (Nφ + α) from the one obtained by multiplying the sine wave signal sin (Nθ + α) output from the plurality of magnetoelectric transducers by the cos (Nφ + β). The relative rotation angle θ is calculated by performing feedback control so that the variable part (Nθ−Nφ) becomes the predetermined value for the right side of the above formula obtained by solving the left side of the above formula to be subtracted. .

Since the sine wave signal and the cosine wave signal output from each magnetoelectric conversion element have a deviation (phase deviation) from a reference angle due to a structural error or the like, the deviation is measured in advance as an angle α and an angle β, By reflecting these angles α and β on the calculated calculation angle Nφ, it is possible to calculate the relative rotation angle θ in consideration of the structural error and the like. In particular, when the left side is solved, as shown on the right side, it can be expressed as a multiplication of a constant represented by (sin αsin β + cos αcos β) and a variable represented by sin (Nθ−Nφ). Since the relative rotation angle θ is calculated by performing feedback control so that −Nφ converges to the predetermined value, the calculation process in the feedback control is not complicated.
Therefore, it is possible to more accurately calculate the relative rotation angle θ while suppressing the occurrence of measurement errors due to structural errors and the like.

  In the invention of claim 2, the cos (Nφ + β) is calculated based on the cosine value output from the cosine value output means by inputting the calculation angle Nφ to the first adding means.

  When the cosine value output means is configured to output cos (Nφ + β) by inputting the calculation angle Nφ, the angle β differs for each rotation sensor, so a dedicated cosine value output means for the angle β is prepared. There is a need to. Therefore, a general-purpose cosine value output means can be adopted by configuring (Nφ + β) output from the first addition means to the cosine value output means, and the first addition means is added. By simply doing this, it is possible to standardize parts and reduce the cost of the cosine value output means.

  In the invention of claim 3, the sin (Nφ + α) is calculated based on the sine value output from the sine value output means by inputting the calculation angle Nφ to the second addition means.

  When the sine value output means is configured to output sin (Nφ + α) by inputting the calculation angle Nφ, the angle α is different for each rotation sensor, so a dedicated sine value output means for the angle α is prepared. There is a need to. Therefore, a general-purpose sine value output means can be adopted by configuring (Nφ + α) output by the second addition means to the sine value output means, and the second addition means is added. As a result, standardization of parts and cost reduction can be achieved for the sine value output means.

  In the fourth aspect of the invention, the ratio of the amplitude of the sine wave signal to the reference amplitude is measured in advance as the first amplitude ratio A based on the signal output as a sine wave signal from each magnetoelectric transducer, and output as a cosine wave signal. The ratio of the amplitude of the cosine wave signal with respect to the reference amplitude is measured in advance as the second amplitude ratio B based on the received signal. Then, the angle calculation unit multiplies the sine wave signal Asin (Nθ + α) output from the plurality of magnetoelectric transducers by Bcos (Nφ + β) to the cosine wave signal Bcos (Nθ + β) multiplied by Asin (Nφ + α). The relative rotation angle θ is calculated by performing feedback control so that the variable part (Nθ−Nφ) becomes the predetermined value for the right side of the above formula obtained by solving the left side of the above formula to be subtracted. .

  Since the sine wave signal and the cosine wave signal output from each magnetoelectric transducer have a deviation (hereinafter referred to as distortion) with respect to the reference amplitude due to a structural error or the like, this distortion is expressed by the first amplitude ratio A to the reference amplitude. In addition to the angle α and the angle β described above, the first amplitude ratio A and the second amplitude ratio B are reflected on the feedback calculation angle Nφ in addition to the angle α and the angle β described above. Thus, the relative rotation angle θ taking into account distortion is calculated, and the relative rotation angle θ can be calculated more accurately by further suppressing the occurrence of measurement errors due to structural errors and the like.

It is explanatory drawing which shows the main structures of the rotation sensor of 1st Embodiment with a block. It is explanatory drawing which shows the flow of the signal between each block shown in FIG. It is explanatory drawing which shows the flow of the signal between each block in 2nd Embodiment. It is explanatory drawing which shows the flow of the signal between each block in 3rd Embodiment. It is explanatory drawing which shows the main structures of the rotation sensor of 4th Embodiment with a block. It is explanatory drawing which shows an example of the use condition of the sensor chip shown in FIG. 5, (a) is a longitudinal cross-sectional view of a sensor chip and a permanent magnet, (b) is a top view of the permanent magnet shown to (a). It is a longitudinal cross-sectional view which shows the state which the permanent magnet shown to Fig.6 (a) rotated 180 degrees. It is explanatory drawing which shows the structure of a sensor chip typically, (a) is a top view, (b) is AA arrow sectional drawing of (a). (A) is a top view of the magnetoresistive element area | region E1 and Hall element area | region E2, (b) is explanatory drawing which shows the arrangement | positioning angle of Hall element H1, H2. It is a top view which shows typically the structure of AMR sensor M1. It is a top view which shows typically the structure of AMR sensor M2. It is an equivalent circuit of the AMR sensor M1. It is an equivalent circuit of the AMR sensor M2. It is explanatory drawing which shows each output signal of AMR sensor M1, M2 and Hall element H1, H2. It is explanatory drawing of Hall element H2, (a) is a top view which shows a part of Hall element H2 and its periphery, (b) is AA arrow sectional drawing of (a), (c) is (a). It is BB arrow sectional drawing of. It is explanatory drawing which shows the flow of the signal between each block shown in FIG. It is explanatory drawing which shows the structure of the initial value table 53d shown in FIG. It is explanatory drawing which shows output waveforms, such as a Hall element, (a) is an output waveform of Hall element H1, (b) is an output waveform of comparison circuit 53a, (c) is an output waveform of Hall element H2, (d) is The output waveform of the comparison circuit 53b, (e) is the output waveform of the output unit 70.

[First Embodiment]
Hereinafter, a rotation sensor according to a first embodiment of the present invention will be described with reference to the drawings. FIG. 1 is an explanatory diagram showing the main configuration of the rotation sensor of the first embodiment in blocks. FIG. 2 is an explanatory diagram showing a signal flow between the blocks shown in FIG.

  A main configuration of the rotation sensor of the present embodiment will be described. As shown in FIG. 1, the rotation sensor 1 of this embodiment includes a rotation detector 2 made of a resolver and the like, and a detection circuit 50 electrically connected to the rotation detector 2. The rotation detector 2 includes an excitation winding 3a and a two-phase output winding (hereinafter referred to as a sine wave phase coil 3b and a cosine wave phase coil 3c), and is supplied from an excitation signal generator (not shown) of the detection circuit 50. Is applied to the excitation winding 3a, the sine wave phase coil 3b and the cosine according to the relative rotation angle θ of the rotor 4 to be measured that rotates relative to the windings 3a to 3c. A two-phase rotation detection signal of a sine wave signal and a cosine wave signal having a phase difference is output from the wave phase coil 3 c to the detection circuit 50. That is, the rotation detector 2 is a one-phase excitation two-phase output type detector that outputs a signal output for one rotation when the rotor 4 is rotated once. Note that the sine wave phase coil 3b and the cosine wave phase coil 3c correspond to an example of a “magnetoelectric conversion element” described in the claims, and the rotor 4 is a “measuring object” described in the claims. It can correspond to an example.

  In the sine wave phase coil 3b and the cosine wave phase coil 3c configured as described above, the output signal sin (θ + α) is output from the sine wave phase coil 3b in the case of the relative rotation angle θ due to the structural error or the like. It can be assumed that the output signal cos (θ + β) is output from the cosine wave phase coil 3c.

  Here, the angle α is output as a sine wave signal from the sine wave phase coil 3b with respect to the sine wave phase coil 3b and the rotor 4 that are relatively rotated so that the relative rotation angle θ becomes a reference angle, for example, 0 °. It is set based on the signal. That is, the angle α is a deviation (phase deviation) from the reference angle due to a structural error or the like related to the sine wave phase coil 3b and the rotor 4, and is measured in advance and reflected in a sinROM (61g) described later.

  Further, the angle β is based on a signal output as a cosine wave signal from the cosine wave phase coil 3c with respect to the cosine wave phase coil 3c and the rotor 4 that are relatively rotated such that the relative rotation angle θ becomes the reference angle. Are set. That is, the angle β is a deviation (phase deviation) from the reference angle due to a structural error or the like related to the cosine wave phase coil 3c and the rotor 4, and is measured in advance and reflected in a cosROM (61f) described later.

  As shown in FIG. 1, the detection circuit 50 includes an amplification unit 51, an angle calculation unit 60, and an output unit 70. The amplifying unit 51 amplifies output signals output from the sine wave phase coil 3b and the cosine wave phase coil 3c. The angle calculation unit 60 calculates a calculation angle φ corresponding to the relative rotation angle θ of the rotor 4 using the amplified signal output from the amplification unit 51. The output unit 70 receives the calculation angle φ calculated by the angle calculation unit 60, and outputs a linear characteristic signal having a voltage Vo corresponding to the calculation angle φ in one cycle while the rotor 4 makes one rotation. To do.

Next, the main electrical configuration of the rotation sensor 1 will be described.
The amplification unit 51 includes differential amplification circuits 51a and 51b. The differential amplifier circuit 51a differentially amplifies the output signal sin (θ + α) of the sine wave phase coil 3b, and the differential amplifier circuit 51b differentially amplifies the output signal cos (θ + β) of the cosine wave phase coil 3c. The angle calculation unit 60 is a tracking loop type digital angle conversion circuit, and includes a signal creation unit 61, a deviation calculation unit 62, a positive / negative determination unit 63, and an up / down counter (U / D counter) 64.

  The angle calculation unit 60 uses signals output from the sine wave phase coil 3b and the cosine wave phase coil 3c, and the deviation between the relative rotation angle θ with respect to the rotor 4 and the calculation angle φ obtained by the calculation converges to a predetermined value. Thus, the feedback control is performed to calculate the relative rotation angle θ.

  The signal creation unit 61 uses the signal sin (θ + α) output from the differential amplifier circuit 51a and the signal cos (θ + β) output from the differential amplifier circuit 51b, and uses the signal (sinαsinβ + cosαcosβ) sin (θ−φ). Create This signal (sinαsinβ + cosαcosβ) sin (θ−φ) will be described later.

  The deviation calculation unit 62 calculates the deviation (θ−φ) using the signal (sin α sin β + cos α cos β) sin (θ−φ) output from the signal creation unit 61. The positive / negative determining unit 63 determines whether the deviation (θ−φ) calculated by the deviation calculating unit 62 is a positive value or a negative value. The up / down counter 64 adds (counts up) or subtracts (counts down) the count value according to the determination result of the positive / negative determination unit 63.

  Here, processing contents executed by the signal creating unit 61 will be described with reference to FIG. Each block indicated by reference numerals 61a to 61g in FIG. 2 indicates the content of processing executed by the signal creation unit 61, or a signal or data generated by the processing.

  The signal creation unit 61 creates a signal of Equation (5) shown in FIG. 2 by multiplying the signal sin (θ + α) by a signal cos (φ + β) created as will be described later and developing it (61c). This multiplication can be performed using a known multiplication circuit.

  Further, the signal creation unit 61 creates a signal of Formula (6) shown in FIG. 2 by multiplying the signal cos (θ + β) by a signal sin (φ + α) created as will be described later and developing the signal (61d). ). This multiplication can be performed using a known multiplication circuit.

  Subsequently, the signal creation unit 61 subtracts the mathematical formula (6) from the mathematical formula (5) and divides it into a variable part and a constant part, so that the signal of the mathematical formula (7) shown in FIG. A sin signal having φ as a variable is created (61e). This subtraction can be performed using a known subtraction circuit.

  Next, the deviation calculation unit 62 performs an inverse sine operation (arcsine operation) on the signal (sin α sin β + cos α cos β) × sin (θ−φ) generated by the signal generation unit 61 to obtain a deviation (θ−φ) (62). Next, the positive / negative determining unit 63 determines whether the deviation (θ−φ) obtained by the deviation calculating unit 62 is a positive value or a negative value.

  When θ−φ≈0, sin (θ−φ) = θ−φ can be approximated. Therefore, the deviation (θ−φ) is not calculated by the deviation calculating unit 62 and the signal ( It is also possible to use a method of determining that the positive value is obtained when sinαsinβ + cosαcosβ) × sin (θ−φ) is greater than 0 and negative when the value is smaller than 0. If this method is used, it is not necessary to perform an inverse sine operation on the signal (sin α sin β + cos α cos β) × sin (θ−φ).

  Next, when the determination result of the positive / negative determination unit 63 is positive, the up / down counter 64 adds n to the least significant bit (LSB) of the counter and adds the count value. If the result is negative, subtract n from the least significant bit of the counter. The count value of the up / down counter 64 is a digital angle, that is, a calculation angle φ (65). In the present embodiment, the above-described n is set to 1. However, in order to improve the convergence, the value of n is changed according to the magnitude of the signal (sin α sin β + cos α cos β) × sin (θ−φ). May be. For example, if the signal (sinαsinβ + cosαcosβ) × sin (θ−φ) is less than or equal to a predetermined threshold, n = 1, and if the signal (sinαsinβ + cosαcosβ) × sin (θ−φ) exceeds the predetermined threshold, n = 5 It can be.

  Further, the signal creation unit 61 creates signals cos (φ + β) and sin (φ + α) using the calculation angle φ (count value) output from the up / down counter 64 (61f, 61g). The signal cos (φ + β) is generated by, for example, preparing in advance a cosROM having a table in which the calculation angle φ (count value) and the data cos (φ + β) are associated with each other, and generating data cos ( (φ + β) can be read and the read data can be converted into an analog signal. The signal sin (φ + α) is created by preparing, for example, a sinROM having a table in which the calculation angle φ (count value) and the data sin (φ + α) are associated in advance, and the data associated with the calculation angle φ. It is possible to read sin (φ + α) and convert the read data into an analog signal.

  Then, the signal creation unit 61 again multiplies the signal sin (θ + α) by the signal cos (φ + β) to create the signal represented by the mathematical formula (5) in FIG. Further, the signal cos (θ + β) is multiplied by the signal sin (φ + α) again to create a signal represented by Expression (6) in FIG. That is, the deviation (θ−φ) is fed back to the signals cos (φ + β) and sin (φ + α), and the signal (sinαsinβ + cosαcosβ) × sin (θ−φ) changes. This feedback is repeated until the deviation (θ−φ) converges to a predetermined value, 0 in this embodiment.

  Next, the output unit 70 outputs a signal obtained by converting the calculation angle φ output from the up / down counter 64 into an analog value. Specifically, the output unit 70 latches the calculation angle φ output from the up / down counter 64, converts the latched calculation angle φ to the analog voltage Vo when the deviation (θ−φ) becomes zero, and calculates An angle signal having the characteristic that the voltage (Vo) increases linearly corresponding to 0 to 360 ° of the angle φ is generated and output.

  As described above, in the rotation sensor 1 according to the present embodiment, a signal output as a sine wave signal from the sine wave phase coil 3b for the rotor 4 that is relatively rotated so that the relative rotation angle θ becomes the reference angle. The deviation from the reference angle included in the sine wave signal is measured in advance as the angle α, and the reference included in the cosine wave signal is based on the signal output as the cosine wave signal from the cosine wave phase coil 3c. The deviation from the angle is measured in advance as the angle β. Then, the angle calculation unit 60 multiplies the sine wave signal sin (θ + α) output from the sine wave phase coil 3b by cos (φ + β) (formula (5) in FIG. 2), and the cosine wave phase coil 3c. Multiplying the output cosine wave signal cos (θ + β) by sin (φ + α) Subtracting equation (6) in FIG. 2 Subtracts sin (θ + α) × cos (φ + β) −cos (θ + β) × sin (φ + α) Thus, (sin α sin β + cos α cos β) × sin (θ−φ) is subjected to feedback control so that the variable portion (θ−φ) becomes the predetermined value, and the relative rotation angle θ is calculated.

As described above, the deviation (phase deviation) from the reference angle caused by the structural error or the like is measured in advance as the angle α and the angle β, and the angle α and the angle β are reflected in the calculation angle φ to be fed back. The relative rotation angle θ can be calculated in consideration of errors and the like. In particular, when sin (θ + α) × cos (φ + β) −cos (θ + β) × sin (φ + α) obtained by subtracting equation (6) from equation (5) in FIG. 2 is solved, a constant represented by (sinαsinβ + cosαcosβ), sin The relative rotation angle can be expressed as multiplication with a variable represented by (θ−φ) (see Equation (7)), and feedback control is performed so that θ−φ converges to the predetermined value as in the prior art. Since θ is calculated, the calculation processing in the feedback control is not complicated.
Therefore, it is possible to more accurately calculate the relative rotation angle θ while suppressing the occurrence of measurement errors due to structural errors and the like.

[Second Embodiment]
Next, a rotation sensor 1 according to a second embodiment of the present invention will be described with reference to FIG. FIG. 3 is an explanatory diagram showing a signal flow between blocks in the second embodiment.
In the signal creation unit 61 of the rotation sensor 1 according to the second embodiment, the processing shown in the blocks 61h and 61j is performed instead of the processing shown in the block 61f, and the processing shown in the block 61g is executed in the blocks 61i and 61k. The point which performs the process shown mainly differs from the rotation sensor which concerns on the said 1st Embodiment. Therefore, substantially the same components as those of the rotation sensor of the first embodiment described above are denoted by the same reference numerals, and description thereof is omitted.

  As shown in FIG. 3, the processing shown in the block 61h is performed using a known addition circuit so that an angle φ + β obtained by adding the angle β to the input calculation angle φ is output to the block 61j. . The processing shown in the block 61j is performed using a cosROM in which necessary nonlinear characteristics are written in advance so as to output a cosine value cos (φ + β) corresponding to the angle φ + β input from the block 61h. That is, the block 61h and the block 61j are equivalent to the block 61f described above, and cos (φ + β) is calculated based on the cosine value output from the block 61j by inputting the calculation angle φ to the block 61h. The The block 61h may correspond to an example of “first addition unit” described in the claims, and the block 61j may correspond to an example of “cosine value output unit” described in the claims.

  The processing shown in the block 61i is performed using a known addition circuit so as to output an angle φ + α obtained by adding the angle α to the input calculation angle φ to the block 61k. Further, the processing shown in the block 61k is performed using a sinROM in which necessary nonlinear characteristics are written in advance so as to output a sine value sin (φ + α) corresponding to the angle φ + α input from the block 61i. That is, the block 61i and the block 61k are equivalent to the block 61g described above, and sin (φ + α) is calculated based on the sine value output from the block 61k by inputting the calculation angle φ to the block 61i. The The block 61i may correspond to an example of “second addition means” recited in the claims, and the block 61k may correspond to an example of “sine value output means” recited in the claims. Further, the angle α and the angle β are measured in advance and stored in the memory of the detection circuit 50 or the like.

  When configured to output the cosine value cos (φ + β) or the sine value sin (φ + α) by inputting the calculation angle φ as in the blocks 61f and 61g described in the first embodiment, Since the angle α and the angle β are different for each rotation sensor, it is necessary to prepare dedicated sinROM and cosROM for the angle.

  Therefore, in the present embodiment, (φ + β) output from the block 61h is input to the cosROM (61j) and (φ + α) output from the block 61i is input to the sinROM (61k). A standard sinROM and cosROM can be adopted, and by simply adding a block 61h and a block 61i made of a known adder circuit, standardization of parts and cost reduction of the sinROM and cosROM can be achieved.

  Note that only the block 61f may be replaced with the blocks 61h and 61j that are characteristic parts of the present embodiment, or only the block 61g may be replaced with the blocks 61i and 61k that are characteristic parts of the present embodiment, depending on the measurement target and the like. May be.

[Third Embodiment]
Next, a rotation sensor 1 according to a third embodiment of the present invention will be described with reference to FIG. FIG. 4 is an explanatory diagram showing a signal flow between blocks in the third embodiment.
In the signal generation unit 61 of the rotation sensor 1 according to the third embodiment, the first amplitude ratio is controlled in order to suppress a deviation (hereinafter referred to as distortion) with respect to the reference amplitude in the sine wave signal and the cosine wave signal due to a structural error or the like. The point that A and the second amplitude ratio B are reflected in the feedback control is mainly different from the rotation sensor according to the first embodiment. Therefore, substantially the same components as those of the rotation sensor of the first embodiment described above are denoted by the same reference numerals, and description thereof is omitted.

  In the present embodiment, in the sine wave phase coil 3b and the cosine wave phase coil 3c, due to the structure error or the like, distortion in which the amplitude deviates from the reference amplitude (design value) when the structure error or the like does not affect occurs. Therefore, in the case of the relative rotation angle θ, the output signal Asin (θ + α) reflecting the first amplitude ratio A is output from the sine wave phase coil 3b, and the second amplitude ratio B is reflected from the cosine wave phase coil 3c. The output signal Bcos (θ + β) is output.

  Here, the first amplitude ratio A is a ratio of the amplitude of the sine wave signal to the reference amplitude, and the sine wave phase coil 3b and the sine wave phase coil 3b are rotated relative to a predetermined relative rotation angle. Is set based on a signal output as a sine wave signal. That is, the first amplitude ratio A is a deviation with respect to the reference amplitude due to distortion caused by a structural error or the like related to the sine wave phase coil 3b and the rotor 4, and is measured in advance and reflected in a sinROM (61g ′) described later. .

  Further, the second amplitude ratio B is the ratio of the amplitude of the cosine wave signal to the reference amplitude, and the cosine wave phase coil 3c and the rotor 4 that are relatively rotated to a predetermined relative rotation angle from the cosine wave phase coil 3c. It is set based on a signal output as a cosine wave signal. That is, the second amplitude ratio B is a deviation from the reference amplitude due to distortion caused by a structural error or the like related to the cosine wave phase coil 3c and the rotor 4, and is measured in advance and reflected in a cosROM (61f ′) described later. .

Hereafter, the processing content which the signal preparation part 61 of the angle calculating part 60 of this embodiment performs is demonstrated using FIG.
The signal creation unit 61 creates a signal of Formula (5) ′ shown in FIG. 4 by multiplying the signal Asin (θ + α) by a signal Bcos (φ + β) created as will be described later (61c). By multiplying the signal Bcos (θ + β) by a signal Asin (φ + α) created as described later and developing the signal, a signal of Formula (6) ′ shown in FIG. 4 is created (61d). Subsequently, the signal generator 61 subtracts the equation (6) ′ from the equation (5) ′ to divide it into a variable portion and a constant portion, thereby obtaining the signal of the equation (7) ′ shown in FIG. A sin signal having (θ−φ) as a variable is created (61e).

  Next, the deviation calculation unit 62 performs an inverse sine operation (arc sine calculation) on the signal AB (sin α sin β + cos α cos β) × sin (θ−φ) generated by the signal generation unit 61, obtains the deviation (θ−φ), and determines whether it is positive or negative The unit 63 determines whether the deviation (θ−φ) obtained by the deviation calculating unit 62 is a positive value or a negative value.

  Next, when the determination result of the positive / negative determination unit 63 is positive, the up / down counter 64 adds n to the least significant bit (LSB) of the counter and adds the count value. If the result is negative, subtract n from the least significant bit of the counter.

  Further, the signal creation unit 61 creates signals Bcos (φ + β) and Asin (φ + α) using the calculation angle φ (count value) output from the up / down counter 64 (61f ′, 61g ′). The signal Bcos (φ + β) is generated by, for example, preparing in advance a cosROM having a table in which the calculation angle φ (count value) and the data Bcos (φ + β) are associated with each other, and data Bcos ( (φ + β) can be read and the read data can be converted into an analog signal. The signal Asin (φ + α) is created by preparing a sinROM having a table in which the calculation angle φ (count value) and the data Asin (φ + α) are associated in advance, and data associated with the calculation angle φ. Asin (φ + α) can be read out and the read data can be converted into an analog signal.

  Then, the signal creation unit 61 again multiplies the signal Asin (θ + α) by the signal Bcos (φ + β) to create a signal represented by Equation (5) ′ in FIG. Further, the signal Bcos (θ + β) is multiplied by the signal Asin (φ + α) again to create a signal represented by the equation (6) ′ in FIG. That is, the deviation (θ−φ) is fed back to the signals Bcos (φ + β) and Asin (φ + α), and the signal AB (sinαsinβ + cosαcosβ) × sin (θ−φ) changes. This feedback is repeated until the deviation (θ−φ) converges to a predetermined value, 0 in this embodiment.

  As described above, in the rotation sensor 1 according to the present embodiment, the ratio of the amplitude of the sine wave signal to the reference amplitude is the first amplitude ratio A based on the signal output as the sine wave signal from the sine wave phase coil 3b. As a second amplitude ratio B, the ratio of the amplitude of the cosine wave signal to the reference amplitude is measured in advance based on a signal output as a cosine wave signal from the cosine wave phase coil 3c. Then, the cosine wave phase coil 3c is obtained by multiplying the sine wave signal Asin (θ + α) output from the sine wave phase coil 3b by the angle calculation unit 60 by Bcos (φ + β) (equation (5) ′ in FIG. 4). Subtracting a signal obtained by multiplying the cosine wave signal Bcos (θ + β) output from Asin (φ + α) (the equation (6) ′ in FIG. 4) by subtracting Asin (θ + α) × Bcos (φ + β) −Bcos (θ + β) × Asin ( For AB (sinαsinβ + cosαcosβ) × sin (θ−φ) obtained by solving (φ + α), feedback control is performed so that the variable portion (θ−φ) becomes the predetermined value, and the relative rotation angle θ is Calculated.

  As described above, the distortion with respect to the reference amplitude caused by the structural error or the like is measured in advance as the first amplitude ratio A and the second amplitude ratio B with respect to the reference amplitude, and the first amplitude ratio is added to the angle α and the angle β described above. By reflecting A and the second amplitude ratio B in the feedback calculation angle φ, the relative rotation angle θ considering the distortion in addition to the phase shift is calculated, and the generation of the measurement error due to the structural error or the like is further increased. Thus, the relative rotation angle θ can be calculated more accurately.

[Fourth Embodiment]
Next, a rotation sensor 1a according to a fourth embodiment of the present invention will be described with reference to the drawings. FIG. 5 is an explanatory diagram showing the main configuration of the rotation sensor 1a of the fourth embodiment in blocks. FIG. 6 is an explanatory view showing an example of a usage state of the sensor chip 15 shown in FIG. 5, (a) is a longitudinal sectional view of the sensor chip 15 and the permanent magnet 12, and (b) is a permanent magnet shown in (a). 12 is a plan view of FIG. FIG. 7 is a longitudinal sectional view showing a state in which the permanent magnet 12 shown in FIG.

  As shown in FIG. 5, the rotation sensor 1 a of this embodiment includes a sensor chip 15 and a detection circuit 50 a electrically connected to the sensor chip 15. The sensor chip 15 includes two anisotropic magnetoresistive sensors (hereinafter referred to as AMR sensors) M1 and M2 made of magnetoresistive elements, and two Hall elements H1 and H2.

  As shown in FIG. 6A, the sensor chip 15 is arranged at a position facing the rotating surface 12c along the diameter of the permanent magnet (magnet generator) 12 to be detected. The sensor chip 15 is supported by a support member (not shown), and is fixed so that the arrangement position does not change. Further, as shown in FIG. 6B, the permanent magnet 12 has a disk shape, one of which is divided into the same size in the radial direction is an N-pole permanent magnet 12a and the other is S. It is a pole permanent magnet 12b. As shown in FIG. 6A, the permanent magnet 12 is attached to the tip of the rotating shaft 13, and rotates in the direction indicated by the arrow F1.

  The permanent magnet 12 generates a magnetic field from the N-pole permanent magnet 12a toward the S-pole permanent magnet 12b, and generates a magnetic field B1 parallel to the surface 15a of the sensor chip 15. In the illustrated example, the magnetic field B1 passes through the Hall element H2 from the Hall element H1. When the permanent magnet 12 rotates 180 ° from the state shown in FIG. 6A as shown in FIG. 7, the direction of the magnetic field B1 changes by 180 °. In the illustrated example, the magnetic field B1 passes through the Hall elements H2 to H1. The permanent magnet 12 may correspond to an example of “measuring object” described in the claims.

As shown in FIG. 5, the detection circuit 50 a includes amplification units 51 and 52, an initial value determination unit 53, an angle calculation unit 60, and an output unit 70 as in the second embodiment. The amplifying unit 51 amplifies output signals output from the AMR sensors M1 and M2. The angle calculation unit 60 calculates the relative rotation angle θ of the permanent magnet 12 using the amplified signal output from the amplification unit 51.
The amplifying unit 52 amplifies output signals output from the Hall elements H1 and H2. The initial value determining unit 53 compares each amplified signal output from the amplifying unit 52 with a threshold value, and the initial value θ0 of the relative rotation angle θ of the permanent magnet 12 is determined from how many times to how many times based on the comparison result. The initial value φ0 of the calculation angle φ corresponding to the determined angle range is determined.

  The output unit 70 receives the calculation angle φ calculated by the angle calculation unit 60 and outputs a linear characteristic signal having a voltage Vo corresponding to the calculation angle φ in one cycle while the permanent magnet 12 makes one rotation. To do.

  Next, the structure of the sensor chip 15 will be described. 8A and 8B are explanatory views schematically showing the structure of the sensor chip 15, wherein FIG. 8A is a plan view and FIG. 8B is a cross-sectional view taken along line AA in FIG. FIG. 9A is a plan view of the magnetoresistive element region E1 and the Hall element region E2, and FIG. 9B is an explanatory diagram showing the arrangement angles of the Hall elements H1 and H2. In each drawing, the dimensions of the hall elements H1 and H2 are drawn larger than the actual dimensions for easy understanding. Also, the shape of the magnetoresistive element is drawn larger than the actual size in order to make the formation direction of the element easy to understand.

  As shown in FIG. 8, the sensor chip 15 includes a silicon substrate 10, an insulating film 90 formed on the surface of the silicon substrate 10, and AMR sensors M1 and M2 (magnetoelectric conversion) formed on the surface of the insulating film 90. Element) and Hall elements H1 and H2 (detection elements) formed in the silicon substrate 10. The AMR sensor M1 includes magnetoresistive elements R1 to R4, and the AMR sensor M2 includes magnetoresistive elements R5 to R8. The Hall elements H1 and H2 are arranged so as to overlap below the magnetoresistive elements R1 to R8 with the insulating film 90 interposed therebetween.

  As shown in FIG. 9B, the Hall elements H1 and H2 are arranged such that the angle formed by the magnetic detection surfaces HP1 and HP2 of each magnetic detection unit HP is 90 °. That is, the Hall elements H1 and H2 are arranged so that the phase difference between the output signals is 90 °. A line extending horizontally from the relative rotation center P1 of the sensor chip 15 toward the magnetoresistive element R2 is a reference line L3, a line parallel to the magnetic detection surface HP1 of the Hall element H1 is L4, and the position of the reference line L3 is a reference. When the angle is 0 °, the Hall element H1 is arranged such that an angle γ formed by its magnetic detection surface HP1 and the reference line L3 is 90 °.

  In addition, the Hall element H2 is arranged so that its magnetic detection surface HP2 and the reference line L3 are parallel to each other. Further, the magnetic detection surface HP1 of the Hall element H1 and the easy axis of the magnetoresistive element R2 arranged at the reference angle of 0 ° form an angle of 90 °. That is, the Hall element H1 outputs a sin θ signal having the same phase with respect to the relative rotation angle θ, and the Hall element H2 outputs a cos θ signal whose phase is 90 ° different from that of the Hall element H1. The signal levels of the sin θ signal and the cos signal change in two cycles according to the strength of the magnetic field while the permanent magnet 12 rotates once.

  Here, a region where the magnetoresistive elements R1 to R8 are disposed is referred to as a magnetoresistive element region E1, and a region where the Hall elements H1 and H2 are disposed is referred to as a Hall element region E2. FIG. 9A is created based on FIG. The magnetoresistive element region E1 has a quadrangular shape, and its area is substantially equal to the minimum area required for arranging the magnetoresistive elements R1 to R8. The hall element region E2 has a T-shape, and its area is substantially equal to the minimum area necessary for arranging the hall elements H1 and H2.

  As illustrated, the entire Hall element region E2 is overlapped below the magnetoresistive element region E1, and a part of the Hall element region E2 does not protrude from the end of the magnetoresistive element region E1. Further, the intersection of the diagonal lines L1 and L2 of the magnetoresistive element region E1 coincides with the relative rotation center P1 of the sensor chip 15.

  That is, the relative rotation center P1 of the sensor chip 15 is located on an extension line of the relative rotation axis C1 (FIG. 6A) of the permanent magnet 12, and the relative rotation center P1 of the sensor chip 15 and the permanent magnet 12 are relative to each other. The center of rotation exists on the same axis. Therefore, even when the sensor chip 15 is relatively rotated around the relative rotation center P1 in a state where the permanent magnet 12 is not rotating, the relative rotation angle of the sensor chip 15 with respect to the permanent magnet 12 can be detected.

Since the sensor chip 15 has the above-described structure, the sensor chip 15 has a size (horizontal width) in the substrate surface direction as compared with the conventional rotation sensor in which the AMR sensor and the Hall element are arranged in the substrate surface direction of the semiconductor substrate. Can be small.
The AMR sensors M1 and M2 and the Hall element regions H1 and H2 are overlapped in a direction corresponding to the relative rotation axis C1 of the permanent magnet 12 (FIG. 6A).
Therefore, since the sensor chip 15 can be reduced in the direction of the relative rotation center of the permanent magnet 12, the space facing the rotation surface 2c of the permanent magnet 12 can be effectively used.

  Next, the structure of the AMR sensors M1 and M2 will be described. FIG. 10 is a plan view schematically showing the structure of the AMR sensor M1. FIG. 11 is a plan view schematically showing the structure of the AMR sensor M2. FIG. 12 is an equivalent circuit of the AMR sensor M1, and FIG. 13 is an equivalent circuit of the AMR sensor M2. FIG. 14 is an explanatory diagram showing output signals of the AMR sensors M1, M2 and the Hall elements H1, H2.

The magnetoresistive elements R <b> 1 to R <b> 8 are formed in a shape obtained by folding a plurality of band-shaped regions, that is, in a meander shape (meandering shape). The resistance values of the magnetoresistive elements R1 to R8 change mainly depending on the intensity and direction of the magnetic field parallel to the surface of the silicon substrate 10, and output a signal having a level corresponding to the resistance value. That is, the magnetoresistive elements R1 to R8 are elements that generate an anisotropic magnetoresistive effect.
In this embodiment, the magnetoresistive elements R1 to R8 are formed of a ferromagnetic metal thin film. As the ferromagnetic material, NiFe (permalloy), NiCo, or the like can be used. The ferromagnetic metal thin film can be formed by sputtering or vapor deposition.

  As shown in FIG. 10, the AMR sensor M1 includes four magnetoresistive elements R1 to R4. The magnetoresistive elements R <b> 1 to R <b> 4 are arranged so that the angle formed by the extending direction of the strip-shaped elements in the adjacent magnetoresistive elements is 90 °. In other words, the magnetoresistive elements R <b> 1 to R <b> 4 are arranged so that the current direction (magnetization easy axis) of the adjacent magnetoresistive elements forms an angle of 90 °. That is, each set of the magnetoresistive elements R1, R4 and R2, R3 is arranged such that the phase between the output signals is 90 ° different in each set.

  As shown in FIG. 12, the magnetoresistive elements R1 and R4 are electrically connected in series to form a half bridge circuit. An output terminal 31 for taking out the midpoint output Vout1 is electrically connected to the midpoint of the half bridge circuit. The magnetoresistive elements R2 and R3 are also electrically connected in series to form a half bridge circuit. An output terminal 32 for taking out the midpoint output Vout2 is electrically connected to the midpoint of the half bridge circuit.

  Both half-bridge circuits are connected in parallel to form a full-bridge circuit that outputs a cos 2θ signal. The full bridge circuit is electrically connected to a power supply terminal 30 for supplying power Vcc and a terminal 33 for electrically connecting to the ground G1. In the full bridge circuit, the magnetoresistive elements R1 and R2 facing each other output the (R0−ΔRcos2θ) signal, and the magnetoresistive elements R3 and R4 output the (R0 + ΔRcos2θ) signal. Here, R0 is a resistance value of the magnetoresistive element in the absence of a magnetic field, and ΔR is a resistance value change amount.

Since each of the midpoint outputs Vout1 and Vout2 vibrates around Vcc / 2, it is possible to suppress an offset of the output waveform caused by a change in environmental temperature or the like.
The output terminals 31 and 32 are connected to a differential amplifier circuit (indicated by reference numeral 51a in FIG. 5), and the midpoint outputs Vout1 and Vout2 are differentially amplified. Therefore, the output amplitude of the AMR sensor M1 can be doubled compared to the case where the AMR sensor M1 is configured by one half bridge circuit, so that the magnetic detection sensitivity can be increased.

  As shown in FIG. 11, the AMR sensor M2 includes four magnetoresistive elements R5 to R8. The magnetoresistive elements R5 to R8 are arranged so that the angle formed by the extending direction of the strip-shaped elements in the adjacent magnetoresistive elements is 90 °. In other words, the magnetoresistive elements R5 to R8 are arranged so that the current direction (magnetization easy axis) of the adjacent magnetoresistive elements forms an angle of 90 °. That is, each set of magnetoresistive elements R5, R7 and R8, R6 is arranged such that the phase between the output signals is 90 ° different in each set.

  As shown in FIG. 13, the magnetoresistive elements R5 and R7 are electrically connected in series to form a half bridge circuit. An output terminal 37 for taking out the midpoint output Vout3 is electrically connected to the midpoint of the half bridge circuit. The magnetoresistive elements R8 and R6 are also electrically connected in series to form a half bridge circuit. An output terminal 38 for taking out the midpoint output Vout4 is electrically connected to the midpoint of the half bridge circuit.

  Both half bridge circuits are connected in parallel to form a full bridge circuit that outputs a sin 2θ signal. The full bridge circuit is electrically connected to a power supply terminal 36 for supplying power Vcc and a terminal 39 for electrically connecting to the ground G2. The magnetoresistive elements R5 and R6 facing each other in this full bridge circuit output (R0 + ΔRsin2θ) signals, and the magnetoresistive elements R7 and R8 output (R0−ΔRsin2θ) signals.

Since each of the midpoint outputs Vout3 and Vout4 vibrates around Vcc / 2, it is possible to suppress an offset of the output waveform caused by a change in environmental temperature or the like.
The output terminals 37 and 38 are connected to a differential amplifier circuit (indicated by reference numeral 51b in FIG. 5), and the midpoint outputs Vout3 and Vout4 are differentially amplified. For this reason, the output amplitude of the AMR sensor M2 can be doubled as compared with the case where the AMR sensor M2 is configured by one half bridge circuit, so that the magnetic detection sensitivity can be increased.

  As shown in FIG. 8A, the magnetoresistive elements of the AMR sensors M1 and M2 are alternately arranged concentrically, and the magnetoresistive elements R1 to R4 of the adjacent AMR sensor M1 and the magnetism of the AMR sensor M2. The resistance elements R5 to R8 are arranged so that the direction of current (magnetization easy axis) forms an angle of 45 °. The amount of change ΔR in the electrical resistance of the anisotropic magnetoresistive element is maximized when the angle between the direction of the current flowing in its own metal thin film (magnetization easy axis) and the direction of the magnetic field is 90 ° and 270 °. , 0 ° and 180 °.

  Therefore, as shown in FIG. 14, the AMR sensor M1 outputs a sin signal whose one wavelength is an electrical angle of 180 °, and the AMR sensor M2 has a phase difference of 45 ° with respect to the AMR sensor M1 and one wavelength has an electrical angle. A 180 ° cos signal is output.

  As shown in FIG. 8B, the magnetoresistive elements R <b> 1 to R <b> 8 constituting the AMR sensors M <b> 1 and M <b> 2 are disposed on the surface layer portion of the silicon substrate 10 through the insulating film 90. The AMR sensors M1 and M2 mainly detect a change in magnetic flux density of the magnetic field B1 parallel to the magnetoresistive element region E1, that is, the silicon substrate 10. The Hall elements H1 and H2 are built in the silicon substrate 10, and are disposed below the magnetoresistive elements R1 to R8 through the insulating film 90. In this embodiment, each of the Hall elements H1 and H2 is a vertical Hall element having a CMOS (Complementary Metal Oxide Semiconductor) structure. The insulating film 90 is a silicon oxide film.

  The Hall elements H1 and H2 are arranged so that the phase difference between the output signals is 90 °. For this reason, as shown in FIG. 14, when the permanent magnet 12 rotates 360 °, the Hall element H1 outputs a sin signal having an electrical angle of 360 °, and the Hall element H2 has an electrical angle of 360 °. The cos signal is output. When the permanent magnet 12 makes two or more revolutions, 360 ° and 0 ° are treated as being continuous.

  Next, the structure of the Hall elements H1 and H2 will be described. Since the Hall elements H1 and H2 have the same structure, the Hall element H2 will be described as an example here. 15A and 15B are explanatory views of the Hall element H2, in which FIG. 15A is a plan view showing the Hall element H2 and a part of the periphery thereof, FIG. 15B is a cross-sectional view taken along line AA in FIG. [FIG. 2] is a sectional view taken along line BB in FIG.

  The Hall element H2 has a high breakdown voltage CMOS transistor (HVCMOS) structure. The Hall element H2 includes a P-type (first conductivity type) silicon substrate (P-sub) 10 and an N-type (second conductivity type) semiconductor region formed in the depth direction from the surface layer portion of the silicon substrate 10. (Nwell) 91, a P-type (first conductivity type) diffusion layer (Pwell) 92 that surrounds the entire circumference of the semiconductor region 91, and a depth direction from the surface layer portion of the silicon substrate 10. P-type (first conductivity type) diffusion layers (Pwell) 93, 99 that divide a region from the surface layer portion to a predetermined depth into three semiconductor regions 91a, 91b, 91c, and respective surface layers of the semiconductor regions 91a, 91b, 91c Contact regions (N + diffusion layers (impurity diffusion regions)) 94 to 98 formed in the portion.

  Terminals S, V1, V2, G3, and G4 are electrically connected to the contact regions 94 to 98 through wiring. Terminals S, G3, and G4 are terminals for supplying a driving current, and terminals V1 and V2 are terminals for taking out a Hall voltage signal. That is, the contact regions 97 and 98 are current supply pairs, and the contact regions 95 and 96 are voltage output pairs. Therefore, the Hall elements H1 and H2 shown in FIG. 8A are arranged so that the lines connecting the current supply pairs are orthogonal to each other. Further, the lines connecting the voltage output pairs are arranged so as to be orthogonal to each other.

As shown in FIG. 15C, a region sandwiched between the contact regions 95 and 96 is a magnetic detection unit (hole plate) HP. Further, in the magnetic detection part HP, both surfaces parallel to the line connecting the contact regions 95 and 96 become the magnetic detection surface HP2, respectively. That is, the Hall element H2 outputs the Hall voltage signal corresponding to the magnetic field applied from the magnetic detection surface HP2 to the magnetic detection unit HP from the terminals V1 and V2.
As shown in FIG. 15B, when a constant drive current i is supplied from the terminal S to the terminal G3 and from the terminal S to the terminal G4, the drive current i is supplied from the contact region 94 to the magnetic detection unit HP, Then, it flows to the contact regions 97 and 98 through the semiconductor region 91 below the diffusion layers 93 and 98, respectively.

  That is, a drive current including a component perpendicular to the substrate surface (sensor chip surface) flows through the magnetic detection unit HP. For this reason, a magnetic field (for example, a magnetic field indicated by an arrow B1 in FIG. 15C) including a component parallel to the substrate surface (sensor chip surface) is applied to the magnetic detection unit HP in a state where the drive current is supplied. The Hall voltage VH corresponding to the magnetic field is generated between the terminals V1 and V2 by the Hall effect. The Hall voltage VH changes according to the angle formed by the magnetic detection surface HP2 and the direction of the magnetic field, that is, the incident angle of the magnetic field with respect to the magnetic detection surface HP2.

  As shown in FIGS. 6A and 7, the Hall elements H <b> 1 and H <b> 2 are arranged so that the magnetic detection surfaces HP <b> 1 and HP <b> 2 are perpendicular to the surface of the silicon substrate 10. 12 and a magnetic field B1 parallel to the surface of the silicon substrate 10 penetrates the magnetic detection surfaces HP1 and HP2 vertically. In the illustrated state, the magnetic field B1 penetrates perpendicularly to the magnetic detection surface HP2 of the Hall element H2, but when the permanent magnet 12 rotates 90 ° from the illustrated position, the magnetic field B1 changes to the magnetic detection surface of the Hall element H1. It penetrates perpendicularly to HP1. That is, the Hall elements H1 and H2 mainly detect changes in the magnetic flux density of the magnetic field B1 parallel to the surface of the silicon substrate 10.

  The N-type semiconductor region 91 is formed deeper than the N-type semiconductor region in the low breakdown voltage CMOS transistor structure, and accordingly, the P-type diffusion layers 92, 93, 98 are also P-type in the low breakdown voltage CMOS transistor structure. It is formed deeper than the diffusion layer. In this embodiment, the P-type diffusion layers 92, 93 and 98 are each formed to a depth approximately half that of the N-type semiconductor region 91.

Thus, since the Hall element H1 has the N-type semiconductor region 91 formed deeply, the carrier mobility is increased and the Hall effect can be increased, so that the Hall voltage VH can be increased. The detection sensitivity with respect to the magnetic field can be increased.
In addition, since the Hall element H1 is manufactured by a CMOS process, it is more cost-effective than a vertical Hall element manufactured by a bipolar process.

  In the sensor chip 15 configured as described above, the angle α ′ and the first amplitude ratio from the AMR sensor M1 in the case of the relative rotation angle θ due to the structural error and the like, as in the third embodiment. An output signal A′sin (2θ + α ′) reflecting A ′ is output, and an output signal B′cos (2θ + β ′) reflecting the angle β ′ and the second amplitude ratio B ′ is output from the AMR sensor M2. Can be assumed.

  Here, the angle α ′ is based on a signal output as a sine wave signal from the AMR sensor M1 with respect to the AMR sensor M1 and the permanent magnet 12 that are relatively rotated so that the relative rotation angle θ becomes a reference angle, for example, 0 °. Is set. That is, the angle α ′ is a deviation (phase deviation) from the reference angle due to a structural error or the like related to the AMR sensor M1 and the permanent magnet 12, and is measured in advance and stored in a memory or the like of the detection circuit 50a.

  Further, the angle β ′ is set based on a signal output as a cosine wave signal from the AMR sensor M2 with respect to the AMR sensor M2 and the permanent magnet 12 that are relatively rotated so that the relative rotation angle θ becomes the reference angle. Is. That is, the angle β ′ is a deviation (phase deviation) from the reference angle due to a structural error or the like related to the AMR sensor M2 and the permanent magnet 12, and is measured in advance and stored in a memory or the like of the detection circuit 50a.

  The first amplitude ratio A ′ is the ratio of the amplitude of the sine wave signal to the reference amplitude, and the sine wave signal from the AMR sensor M1 for the AMR sensor M1 and the permanent magnet 12 rotated relative to a predetermined relative rotation angle. Is set based on the signal output as. That is, the first amplitude ratio A ′ is a deviation from the reference amplitude due to distortion caused by a structural error or the like regarding the AMR sensor M1 and the permanent magnet 12, and is measured in advance and reflected in the above-described sinROM (61i).

  Further, the second amplitude ratio B ′ is a ratio of the amplitude of the cosine wave signal to the reference amplitude, and the cosine wave signal from the AMR sensor M2 with respect to the AMR sensor M2 and the permanent magnet 12 rotated relative to a predetermined relative rotation angle. Is set based on the signal output as. That is, the second amplitude ratio B ′ is a deviation from the reference amplitude due to distortion caused by a structural error or the like related to the AMR sensor M2 and the permanent magnet 12, and is measured in advance and reflected in the above-described cosROM (61j).

  Next, the main electrical configuration of the rotation sensor 1a according to the present embodiment will be described. FIG. 16 is an explanatory diagram showing a signal flow between the blocks shown in FIG. FIG. 17 is an explanatory diagram showing the structure of the initial value table 53d shown in FIG. 18A and 18B are explanatory diagrams showing the output waveform of the Hall element, where FIG. 18A shows the output waveform of the Hall element H1, FIG. 18B shows the output waveform of the comparison circuit 53a, FIG. 18C shows the output waveform of the Hall element H2, (D) is an output waveform of the comparison circuit 53b, and (e) is an output waveform of the output unit.

  As shown in FIG. 5, the amplification unit 52 includes amplification circuits 52a and 52b. The amplifier circuit 52a amplifies the detection signal sin θ output from the Hall element H1, and the amplifier circuit 52b amplifies the detection signal cos θ output from the Hall element H2. The initial value determining unit 53 includes comparison circuits 53a and 53b, an initial value reading unit 53c, and an initial value table 53d.

  The comparison circuit 53a compares the signal level VH1 of the detection signal (FIG. 18 (a)) output from the amplification circuit 52a with a threshold value (0V), and a pulse signal (FIG. 18 (b)) corresponding to the comparison result. Is output. The comparison circuit 53b compares the signal level VH2 of the detection signal (FIG. 18C) output from the amplification circuit 52b with a threshold value (0V), and a pulse signal corresponding to the comparison result (FIG. 18D). Is output.

  The initial value determination unit 53 uses the comparison results between the signal levels VH1 and VH2 of the detection signals output from the Hall elements H1 and H2 and the threshold value (0 V), that is, the outputs of the comparison circuits 53a and 53b. An angle range including the initial value θ0 of the relative rotation angle θ is determined. The absolute value of the difference between the initial value θ0 of the relative rotation angle θ and the initial value φ0 of the calculation angle φ that can occur within the determined angle range is less than 90 ° (| θ0−φ0 | <90 °). The initial value φ0 of the calculation angle φ is determined using the initial value table 53d.

  As shown in FIG. 18B, the comparison circuit 53a maintains a high level (H) when the input angle θ is 0 to 180 °, and a low level (when the input angle θ is 180 to 360 °). A pulse signal maintaining L) is output. Further, as shown in FIG. 18D, the comparison circuit 53b maintains a high level (H) when the input angle θ is 90 to 270 ° and low when the input angle θ is 270 to 90 °. A pulse signal that maintains the level (L) is output.

  Accordingly, when the signal level of the pulse signal output from the comparison circuit 53a is H and the signal level of the pulse signal output from the comparison circuit 53b is L in the initial state where the permanent magnet 12 is not rotating. The initial value θ0 of the relative rotation angle θ of the permanent magnet 12 can be determined to exist in the first quadrant (0 ° ≦ θ <90 °). When the signal level of the pulse signal output from the comparison circuit 53a is H and the signal level of the pulse signal output from the comparison circuit 53b is also H, the initial value of the relative rotation angle θ of the permanent magnet 12 is set. It can be determined that θ0 exists in the second quadrant (90 ° ≦ θ <180 °).

When the signal level of the pulse signal output from the comparison circuit 53a is L and the signal level of the pulse signal output from the comparison circuit 53b is H, the initial value of the relative rotation angle θ of the permanent magnet 12 is set. It can be determined that θ0 exists in the third quadrant (180 ° ≦ θ <270 °). Further, when the signal level of the pulse signal output from the comparison circuit 53a is L and the signal level of the pulse signal output from the comparison circuit 53b is also L, the initial value of the relative rotation angle θ of the permanent magnet 12 is set. It can be determined that θ0 exists in the fourth quadrant (270 ° ≦ θ <360 °).
That is, the quadrant (angle range) where the initial value θ0 of the relative rotation angle θ of the permanent magnet 12 exists can be determined by using a combination of signal levels of the pulse signals output from the comparison circuits 53a and 53b. .

  In this embodiment, the angle range including the initial value θ0 is divided by a value 4 obtained by dividing 0 to 360 ° of the relative rotation angle θ by the phase difference 90 ° between the output signals of the Hall elements H1 and H2. Four angle ranges are set. That is, as shown in FIG. 17, the first quadrant (0 ° ≦ θ0 <90 °), the second quadrant (90 ° ≦ θ0 <180 °), the third quadrant (180 ° ≦ θ0 <270 °), and Four quadrants composed of the fourth quadrant (270 ° ≦ θ0 <360 °) are set as the angle range.

  As shown in FIG. 17, the initial value table 53d includes a signal level VH1 (H or L) of the pulse signal output from the comparison circuit 53a and a signal level VH2 (H of the pulse signal output from the comparison circuit 53b. Or L) and the initial value φ0 of the calculation angle φ are associated with each other. In this embodiment, the combinations H and L of the signal levels VH1 and VH2 and the initial value 45 ° are the combinations H and H of the signal levels VH1 and VH2 and the initial value 135 ° are the combinations L and H of the signal levels VH1 and VH2. The initial value 225 ° is associated with the combinations L and L of the signal levels VH1 and VH2 and the initial value 315 °, respectively, and the combinations of the signal levels VH1 and VH2 are all different in each quadrant.

  Each initial value is a digital angle, which is a count value by an up / down counter 64 constituting an angle calculation unit 60 described later, and the count value is stored in the initial value table 53d as an initial value. The initial value table 53d can be stored in a storage medium such as a ROM or a flash ROM.

  The initial value reading unit 53c (FIG. 5) reads the initial value φ0 associated with the combination of the signal levels VH1 and VH2 of each pulse signal output from the comparison circuits 53a and 53b with reference to the initial value table 53d. put out. For example, when the combination of the signal levels VH1 and VH2 of the pulse signals output from the comparison circuits 53a and 53b is H and L, the initial value reading unit 53c sets 45 ° as the initial value φ0 from the initial value table 53d. Read out.

  In the present embodiment, AMR sensors M1 and M2 corresponding to magnetoelectric transducers are arranged in the magnetic field of a permanent magnet (magnet generator) 12 that rotates relatively. Then, the AMR sensors M1 and M2 output a sinNθ signal and a cosNθ signal in which the relative rotation angle with respect to the permanent magnet 12 is θ and N is a natural number according to the strength of the magnetic field during one rotation of the permanent magnet 12. It has become. In the example of the present embodiment, the AMR sensors M1 and M2 output the sin 2θ signal and the cos 2θ signal whose signal level changes in two periods according to the strength of the magnetic field while the permanent magnet 12 rotates once. , N = 2.

  The differential amplifier circuit 51a of the amplifier 51 differentially amplifies the output signal A′sin (2θ + α ′) of the AMR sensor M1, and the differential amplifier circuit 51b outputs the output signal B′cos (2θ + β ′) of the AMR sensor M2. Is differentially amplified. The angle calculation unit 60 is a tracking loop type digital angle conversion circuit, and includes a signal creation unit 61, a deviation calculation unit 62, a positive / negative determination unit 63, and an up / down counter (U / D counter) 64.

  The angle calculator 60 uses the signals output from the AMR sensors M1 and M2 to perform feedback control so that the deviation between the relative rotation angle θ with respect to the permanent magnet 12 and the calculated angle φ obtained by the calculation converges to a predetermined value. To calculate the relative rotation angle θ. The angle calculation unit 60 uses the initial value φ0 determined by the initial value determination unit 53 as the initial value φ0 of the calculation angle φ when starting the calculation of the relative rotation angle θ (54).

  The signal creation unit 61 uses the signal A′sin (2θ + α ′) output from the differential amplifier circuit 51a and the signal B′cos (2θ + β ′) output from the differential amplifier circuit 51b, and uses the signal A′B. '(Sinα'sinβ' + cosα'cosβ ') sin (2θ-2φ) is created. This signal A′B ′ (sinα′sinβ ′ + cosα′cosβ ′) sin (2θ−2φ) will be described later.

  The deviation calculation unit 62 calculates the deviation (2θ-2φ) using the signal A′B ′ (sin α′sin β ′ + cos α′cos β ′) sin (2θ-2φ) output from the signal creation unit 61. The positive / negative determining unit 63 determines whether the deviation (2θ−2φ) calculated by the deviation calculating unit is a positive value or a negative value. The up / down counter 64 adds (counts up) or subtracts (counts down) the count value according to the determination result of the positive / negative determination unit 63.

Here, processing contents executed by the signal creating unit 61 will be described with reference to FIG.
The signal creation unit 61 multiplies the signal A′sin (2θ + α ′) by a signal B′cos (2φ + β ′) created as described later, and develops the result, so that the formula (5) ″ shown in FIG. A signal is created (61c). Further, the signal creation unit 61 multiplies the signal B′cos (2θ + β ′) by a signal A′sin (2φ + α ′) created as will be described later, and develops it, thereby expressing the formula (6) ′ shown in FIG. 'Signal is created (61d). Subsequently, the signal generating unit 61 subtracts the formula (6) '' from the formula (5) '' to divide it into a variable part and a constant part, whereby the signal of the formula (7) '' shown in FIG. That is, a sin signal having a deviation (2θ−2φ) as a variable is created (61e).

  Next, the deviation calculation unit 62 performs an inverse sine operation (arcsine operation) on the signal A′B ′ (sin α′sin β ′ + cos α′cos β ′) × sin (2θ−2φ) generated by the signal generation unit 61 to obtain the deviation. (2θ-2φ) is obtained (62). Next, the positive / negative determining unit 63 determines whether the deviation (2θ−2φ) obtained by the deviation calculating unit 62 is a positive value or a negative value.

  Next, when the determination result of the positive / negative determination unit 63 is positive, the up / down counter 64 adds n to the least significant bit (LSB) of the counter and adds the count value. If the result is negative, subtract n from the least significant bit of the counter. The count value of the up / down counter 64 is a digital angle, that is, a calculation angle φ (65).

  Further, the signal generation unit 61 uses the calculation angle φ (count value) output from the up / down counter 64, and outputs the calculation angle φ from the block 61j by inputting the calculation angle φ to the block 61h, as in the second embodiment. B′cos (2φ + β ′) is created based on the cosine value to be generated, and A′sin (2φ + α ′) is created based on the sine value output from the block 61k by inputting the calculation angle φ to the block 61i. To do.

  Each φ of 2φ + β ′ (61h) and 2φ + α ′ (61i) is a variable that varies depending on the count value of the up / down counter 64. Before the permanent magnet 12 starts rotating, that is, before the rotation sensor 1a detects the relative rotation angle θ, the initial value reading unit 53c uses the initial value φ0 read from the initial value table 53d as the calculation angle φ.

  Then, the signal creation unit 61 again multiplies the signal A′sin (2θ + α ′) by the signal B′cos (2φ + β ′) to create a signal represented by Expression (5) ″ in FIG. Further, the signal B′cos (2θ + β ′) is multiplied by the signal A′sin (2φ + α ′) again to create a signal represented by the equation (6) ″ in FIG. That is, the deviation (2θ−2φ) is fed back to the signals B′cos (2φ + β ′) and A′sin (2φ + α ′), and the signal A′B ′ (sinα′sinβ ′ + cosα′cosβ ′) × sin (2θ− 2φ) changes. This feedback is repeated until the deviation (2θ-2φ) converges to a predetermined value, 0 in this embodiment.

  Next, the output unit 70 outputs a signal obtained by converting the calculation angle φ output from the up / down counter 64 into an analog value. Specifically, the output unit 70 latches the calculation angle φ output from the up / down counter 64, converts the latched calculation angle φ to the analog voltage Vo when the deviation (θ−φ) becomes zero, and calculates An angle signal having the characteristic that the voltage (Vo) increases linearly corresponding to 0 to 360 ° of the angle φ is generated and output.

  As described above, in the rotation sensor 1a according to this embodiment, the AMR sensors M1 and M2 have a sine wave signal whose signal level changes in two cycles according to the strength of the magnetic field while the permanent magnet 2 makes one rotation. It is configured to output a cosine wave signal. Then, the deviation (phase deviation) from the reference angle caused by the structural error or the like is measured in advance as the angle α ′ and the angle β ′, and the distortion with respect to the reference amplitude caused by the structural error or the like is the first amplitude ratio A to the reference amplitude. By measuring in advance as 'and the second amplitude ratio B', and reflecting these angles α 'and β', the first amplitude ratio A 'and the second amplitude ratio B' in the feedback calculation angle φ, the structure error The relative rotation angle θ can be calculated in consideration of the above.

In particular, A′B′sin (2θ + α ′) × cos (2φ + β ′) − A′B′cos (2θ + β ′) × sin (2φ + α) obtained by subtracting equation (6) ″ from equation (5) ″ in FIG. When ') is solved, it can be expressed as multiplication of a constant represented by A'B' (sin α'sin β '+ cos α'cos β') and a variable represented by sin (2θ-2φ) (formula (7) ' Since the relative rotation angle θ is calculated by performing feedback control so that 2θ−2φ converges to the predetermined value, the calculation process in the feedback control is not complicated as in the conventional technique.
Therefore, it is possible to more accurately calculate the relative rotation angle θ while suppressing the occurrence of measurement errors due to structural errors and the like.

  Magnetoelectric transducers such as AMR sensors M1 and M2 output a sin Nθ signal and a cos Nθ signal in which the relative rotation angle with respect to the permanent magnet 12 is θ and N is a natural number according to the strength of the magnetic field during one rotation of the permanent magnet 12. It may be configured to. Even in this case, since the relative rotation angle θ is calculated by performing feedback control so that Nθ−Nφ converges to the predetermined value, the calculation processing in the feedback control is not complicated.

  In the fourth embodiment, similarly to the third embodiment, a block 61f is adopted instead of the blocks 61h and 61j, and the operation angle φ is input to the block 61f, whereby the signal B′cos (2φ + β ') May be configured to output. In this case, the block 61f is configured by a dedicated cosROM that reflects the angle β ′ and the second amplitude ratio B ′. Further, instead of the blocks 61i and 61k, a block 61g may be adopted, and a signal A′sin (2φ + α ′) may be output by inputting a calculation angle φ to the block 61g. In this case, the block 61f is configured by a dedicated sinROM that reflects the angle α ′ and the first amplitude ratio A ′.

In addition, this invention is not limited to said each embodiment and its modification, You may actualize as follows.
(1) Applications of the rotation sensors 1 and 1a according to the present invention are not limited as long as the detection target is a relative rotation. For example, a crank angle sensor that detects a crank angle of a crankshaft provided in an internal combustion engine, a cam angle sensor that detects a cam angle of a camshaft, a steering angle sensor that detects a steering angle of a steering device provided in a vehicle, etc. Can be applied. The present invention can also be applied to a sensor that detects the angle of a joint provided in a robot.

(2) In each of the above embodiments, a sine wave signal and a cosine wave signal whose signal level changes in N periods (where N is a natural number) according to the strength of the magnetic field while the measurement object that rotates relative to each other makes one rotation are output. Instead of the sine wave phase coil 3b, the cosine wave phase coil 3c, and the AMR sensors M1 and M2, other sensors that can function as magnetoelectric conversion elements can be used as the magnetoelectric conversion element.

(3) In the configuration according to each of the above embodiments, the angle calculation unit 60, the initial value determination unit 53, and the output unit 70 can be realized by hardware such as a discrete circuit, or by software using a microcomputer. It can also be realized.

(4) In the configuration according to each of the above embodiments, the content performed by the initial value determination unit 53 may be performed by the output unit 70.

(5) In the configuration according to each of the embodiments described above, the output unit 70 can also be configured to output a digital value without converting the calculation angle into an analog signal.

(6) In the configuration according to the fourth embodiment, the detection circuit 50a can be formed on the silicon substrate 10, and the sensor chip 15 and the detection circuit 50a can be integrated.
Moreover, it can replace with the silicon substrate 10 and the board | substrate formed with compound semiconductors, such as GaAs, InAs, and InSb, can also be used.
Further, instead of the permanent magnet 12, a member coated with magnetic ink can be used. A member magnetized on the surface of the conductive member can also be used.

DESCRIPTION OF SYMBOLS 1,1a ... Rotation sensor 2 ... Rotation detector 3b ... Sine wave phase coil (magnetoelectric conversion element)
3c ... cosine phase coil (magnetoelectric transducer)
4 ... Rotor (measurement target)
12 ... Permanent magnet (measurement object)
DESCRIPTION OF SYMBOLS 15 ... Sensor chip 50, 50a ... Detection circuit 60 ... Angle calculating part 61 ... Signal preparation part 70 ... Output part A ... 1st amplitude ratio B ... 2nd amplitude ratio H1, H2 ... Hall element M1, M2 ... AMR sensor (magnetoelectricity) Conversion element)
α, β ... Angle θ ... Relative rotation angle φ ... Calculation angle

Claims (4)

A sine wave signal and a cosine wave signal whose signal levels change in N periods (where N is a natural number) according to the strength of the magnetic field while the measurement object that rotates relative to each other rotates, respectively, and the sine wave signal And a plurality of magnetoelectric transducers arranged so as to produce a phase difference between the cosine wave signals,
In a rotation sensor configured to obtain a relative rotation angle with respect to the measurement object using the sine wave signal and the cosine wave signal output from each magnetoelectric conversion element,
Using the sine wave signal and the cosine wave signal output from the plurality of magnetoelectric transducers, the deviation between the relative rotation angle θ with respect to the measurement object and the calculation angle φ obtained by calculation converges to a predetermined value. An angle calculation unit for calculating the relative rotation angle θ by performing feedback control;
An output unit that outputs a signal corresponding to the calculation angle φ calculated by the angle calculation unit,
The reference angle included in the sine wave signal based on a signal output as the sine wave signal from each magnetoelectric conversion element with respect to the measurement object relatively rotated so that the relative rotation angle θ becomes a reference angle. And measuring in advance as the angle α, based on the signal output as the cosine wave signal, and measuring in advance as the angle β the deviation from the reference angle included in the cosine wave signal,
The angle calculator is
Subtracting a value obtained by multiplying the cosine wave signal cos (Nθ + β) by sin (Nφ + α) from a value obtained by multiplying the sine wave signal sin (Nθ + α) output from the plurality of magnetoelectric transducers by cos (Nφ + β). The variable portion (Nθ−Nφ) of the right side of the following equation obtained by solving the left side of the equation is subjected to feedback control so that the predetermined value is obtained, and the relative rotation angle θ is calculated. Rotation sensor.
sin (Nθ + α) × cos (Nφ + β) −cos (Nθ + β) × sin (Nφ + α)
= (Sinαsinβ + cosαcosβ) × sin (Nθ−Nφ) First addition means for outputting an angle obtained by adding the angle β to an input angle;
Cosine value output means for outputting a cosine value corresponding to the angle input from the first addition means,
2. The rotation according to claim 1, wherein cos (Nφ + β) is calculated based on a cosine value output from the cosine value output unit when a calculation angle Nφ is input to the first addition unit. Sensor. Second addition means for outputting an angle obtained by adding the angle α to the input angle;
Sine value output means for outputting a sine value according to the angle input from the second addition means,
3. The sin (Nφ + α) is calculated based on a sine value output from the sine value output means by inputting a calculation angle Nφ to the second addition means. Rotation sensor. Based on the signal output as the sine wave signal from each magnetoelectric conversion element, the ratio of the amplitude of the sine wave signal to the reference amplitude is measured in advance as a first amplitude ratio A and output as the cosine wave signal. Based on the signal, a ratio of the amplitude of the cosine wave signal to the reference amplitude is measured in advance as a second amplitude ratio B,
The angle calculator is
Subtracting the cosine wave signal Bcos (Nθ + β) multiplied by Asin (Nφ + α) from the one obtained by multiplying the sine wave signal Asin (Nθ + α) output from the plurality of magnetoelectric transducers by Bcos (Nφ + β) The variable portion (Nθ−Nφ) of the right side of the following equation obtained by solving the left side of the equation is subjected to feedback control so that the predetermined value is obtained, and the relative rotation angle θ is calculated. The rotation sensor according to any one of claims 1 to 3.
Asin (Nθ + α) × Bcos (Nφ + β) −Bcos (Nθ + β) × Asin (Nφ + α)
= AB (sinαsinβ + cosαcosβ) × sin (Nθ−Nφ)

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