JP5411659B2 - Multipath distortion equalization apparatus and reception apparatus in OFDM signal reception - Google Patents

Description

  The present invention relates to a multipath distortion equalization apparatus and a reception apparatus that compensate for multipath distortion when receiving radio waves of an orthogonal frequency division multiplexing (OFDM system).

  In general, in a radio transmission path, a radio wave emitted from one transmission antenna may receive a reflected wave and a direct wave reflected from a building or a mountain. A plurality of radio waves of the same signal having different arrival times due to different propagation paths are called multipath waves.

  Conventionally, in domestic terrestrial digital television broadcasting, an ISDB-T (Integrated Services of Digital Broadcasting Terrestrial) system based on a modulation system called an OFDM system has been adopted. When receiving an OFDM radio wave, it may be a multipath wave. In OFDM terrestrial digital television broadcasting, signals having the same content are transmitted from a plurality of transmitting stations or relay stations using a single frequency network (SFN) using the same frequency. That is, radio waves transmitted from a plurality of transmission antennas have different arrival times because of their different arrival paths, and become multipath waves.

  The received multipath wave (received wave) with the highest level is called the main wave (desired wave), and the incoming wave other than the main wave of the multipath wave (other radio waves other than the desired wave) is called the delayed wave. . The difference between the arrival time of the main wave and the arrival time of the delay wave is referred to as the delay time with reference to the arrival time of the main wave. The delay time has a polarity (delay or advance). When the sign of the delay time is positive, it indicates “delay”, and when the sign of the delay time is negative, it indicates “advance”.

  When multipath distortion occurs in the multipath wave demodulated by the receiver or the equalizer and the level of the delayed wave exceeds a predetermined value, the bit error rate (BER) limit of the demodulated signal is exceeded, It may become impossible to receive.

  In order to improve such a situation and enable reception even for multipath waves, a method in which a guard interval is added by a signal method is adopted. In this method, at the time of reception, a multipath distortion is equalized using a scattered pilot (SP) inserted in a part of the signal, and normal reception can be performed without increasing the BER. . Therefore, in this method, there is no problem when the delay time is equal to or shorter than the guard interval.

  Here, the equalization of multipath distortion is expressed by a mathematical expression as follows. For simplicity, it is assumed that there is one delay wave (in the case of one delay wave). First, the transfer function H (f) of the propagation path of the delayed wave is expressed by Expression (1).

In Equation (1), R is the delay wave level / main wave level (delay wave amplitude / main wave amplitude), t _d is the delay time normalized by the sampling interval, and α is the high-frequency phase difference between the main wave and the delay wave. , F is a frequency (variable) normalized by the fundamental frequency, and N is the total number of data to be analyzed. The time including the total number of data to be analyzed is called a period in analysis.

Of these, the total number N of data corresponds to the period normalized by the sampling interval in the time domain. Specifically, if the effective symbol length T _u is the period in the analysis and the sampling interval is Δt, there is a relationship of T _u = NΔt.
The total number N of data corresponds to a sampling frequency that is standardized with a fundamental frequency in the frequency domain. Specifically, when the basic frequency f ₀ and the sampling frequency f _s are used, there is a relationship of f _s = Nf ₀ .
Note that Equation (1) is H (f) = 1 when there is no delay (t _d = 0). That is, the first term on the right side represents a term representing the main wave, and the second term represents the term representing the delay.

The received multipath wave is expressed by equation (2) using the transfer function H (f) shown in equation (l).

Here, C _m is the main wave (Main Wave), C _r is the received wave: shows the (Received Wave sum of the delay wave and the dominant wave). Multipath distortion equalization is to equalize a received wave to obtain a main wave. Therefore, equation (3) obtained by modifying equation (2) represents equalization of multipath distortion. The received wave here indicates a carrier converted into a signal in the frequency domain.

That is, in order to equalize multipath distortion, which is delay distortion, it is necessary to obtain a transfer function by some means. The transfer function is expressed by Expression (4) obtained by modifying Expression (3).

In the equation (4), the main wave C _m (f) indicating the denominator on the right side cannot generally be clearly obtained with respect to the change in the frequency f, but the SPs inserted at a predetermined interval are used. Yes, if you use it. In a large number of carriers in the frequency direction (identification numbers 0, 1, 2,...), For example, if SPs are arranged on the 0th carrier, the 3rd carrier, the 6th carrier,. 3) are arranged on one carrier. This is referred to as an L-carrier interval (interval of three carriers). In the system (mode 3) adopted in Japan, L = 3. However, the present invention is not limited to this, and for example, L = 6 or the like can be adopted. In short, the SP is inserted at a predetermined interval with respect to a change in the frequency f (variable) normalized by the fundamental frequency. In addition, the SP value indicating the SP arrangement in the segment configuration is defined by the standard. Therefore, for example, at the frequency of SP arranged at a predetermined interval among the frequencies f (variables), the transfer function can be obtained using the above-described equation (4).

Since the input multipath wave is an infinite data string during reception, in order to obtain SP from this received signal, first, the number of data to be subjected to analysis by Fourier transform using discrete value processing using finite data Get (cut out the data). At this time, it is cut out in synchronization with the effective symbol. Note that the time indicated by the product of the number of cut data (for example, the total number of data N) and the sampling interval Δt is a period (for example, T _u ). Then, for example, the signal is converted into a frequency domain signal by conversion such as Fast Fourier Transform (FFT), and the carrier is demodulated for each symbol (symbol number is s). However, the signal in the case of extracting SP in this way is not represented by a single variable of only the frequency f but by a binary variable of the symbol number s and the frequency f. In this case, the above equation (4) can be rewritten as equation (5).

When SP data is used in this way, the data interval (frequency interval) in the frequency domain increases from one interval to L intervals. Therefore, when the total number of data is N, the total number of data N _SP in the frequency domain becomes N / L. Similarly, when the effective symbol length is T _u , the period T _sp of the time domain signal is T _u / L.

The sampling theorem indicates that the frequency that can be reproduced without folding in the frequency domain is within ½ of the sampling frequency f _s . That is, the frequency is in the frequency of f _s / 2 + δ beyond the f _s / 2, resulting aliasing is reproduced frequency -f _s / 2 + _δ. Expressing this sampling theorem in the time domain, it can be said that the reproducible time is within half of the period.

  In the time domain, the “expressed time” changes with respect to an increase in “event time” such as delay time when the “event time” exceeds 1/2 of the period T (T / 2). , “Represented time” returns to −T / 2 and increases in a relationship equal to the increase in “event time”. Thus, the “expressed time” when the “event time” exceeds T / 2 is referred to as turnaround time. It should be noted that the turnaround time can be similarly defined when the “event time” exceeds −T / 2.

The transfer function H (s, f) at the SP frequency shown in the above equation (5) includes R, t _d , and α, respectively, as in the above equation (1). R, t _d , and α included in the transfer function H (s, f) in the equation (5) can be obtained by performing an inverse Fourier transform on the transfer function using, for example, IFFT (Inverse FFT). The calculation formula of the inverse Fourier transform at this time is represented by Formula (6).

Here, h (t) represents the level of the intensity of the incoming radio wave with respect to the arrival time t in the multipath wave, and is referred to as a delay profile. In the delay profile h (t) of Expression (6), the integral value at t = 0 indicates the level of the main wave, and becomes an impulse with h (0) = N. Further, the integrated value at t = t _d indicates the level of the delayed wave, and becomes an impulse with h (t _d ) = Re ^jα N. When these mathematical expressions of the main wave level and the delayed wave level are respectively divided on both sides, the following expression (7) is established. The left side of Expression (7) represents the relative value of the level of the delayed wave with respect to the level of the main wave as a complex number, and is called a complex delayed wave level.

Therefore, by measuring the delay profile h (t), R, t _d , and α included in the transfer function H (s, f) of the above-described equation (5) can be obtained.

  As means for obtaining a delay profile, there are known a method of performing an IFFT on a transfer function and a method of performing an IFFT on a power spectrum of a multipath wave (power spectrum method: see Patent Document 1 and Non-Patent Document 1). In the delay profile measurement by the IFFT method of the transfer function, “period / 2” without folding is the measurable delay time. On the other hand, the advantage of the power spectrum method is that a reference signal such as SP is unnecessary, and there is no principle restriction on measurable delay time. Specifically, the measurable delay time can be lengthened by lengthening the signal period and making the turn-back time longer than the measurement target time. The disadvantage of this power spectrum method is that the IFFT output of the power spectrum also shows a delayed wave and its conjugate complex number. That is, the impulse of the delayed wave appears in “± delay time”.

It is necessary to equalize delay distortion (multipath distortion) for carriers (information carriers) arranged at frequencies other than the SP frequency. On the other hand, the transfer function H (s, f) is a transfer function at the frequency of SP, and is used to obtain R, t _d , and α. Therefore, before performing equalization using equation (3), a transfer function is generated for all frequencies (however, the frequency of the information carrier) between SP and SP inserted in the received signal at predetermined intervals. This must be used for equalization.

The transfer function used for this equalization is generated as follows. That is, after substituting R, t _d , and α obtained from the measurement of the delay profile h (t) and the above equation (7) into the above equation (1), all the frequencies indicating the frequency of the information carrier are shown. The right side is calculated for each frequency f. As a result, a total of N data is generated as a transfer function. By substituting these transfer functions into the above equation (3), the main wave C _m (f) free from multipath distortion can be obtained.

JP 2005-268831 A

Kazuhiko Kuruyama, 3 others, “Development of a long-range delay profile measurement device under SFN environment”, Journal of the Institute of Image Information and Television Engineers, 2007, vol.61, No.7, p.990-996

However, when using SP data of L carrier intervals, the period of the signal in the time domain of the received signal is assumed to be T _sp (= T _u / L) having a period shorter than the effective symbol length T _u. When the delay time t _d, which is the “event time”, exceeds T _sp / 2, a problem of turnaround time occurs due to the sampling theorem.

In the above equation (6), the impulse time representing the level of the delayed wave corresponds to the “expressed time” with respect to the increase in the “event time”. When T _sp / 2 is exceeded, the impulse time representing the level of the delayed wave in equation (6) is different from t _d . This time is t _err . When the transfer function at the SP frequency is generated when the delay time t _d exceeds T _sp / 2, a transfer function H _err (s, f) shown in Expression (8) is generated.

Further, when the delay time t _d exceeds T _sp / 2, an equation representing equalization similar to the equation (3) is expressed by the equation (9).

Here, C _err (s, f) indicates a main wave level corresponding to H _err (s, f), and C (s, f) indicates a main wave level corresponding to H (s, f). Indicates. In Expression (9), the delay time t _{err of} H _err (s, f) indicated by the denominator on the right side is different from the delay time t _{d of} H (s, f) indicated by the numerator on the right side. Therefore, the main wave level C (s, f) cannot be obtained from the left side C _err (s, f) of equation (9).

Thus, theoretically, the problem of the turnaround time occurs when the delay time t _d exceeds T _sp / 2. In practice, if the delay time t _d exceeds the range of ± T _sp (= ± T _u / L) in the receiver, the arrival time of the detected delay wave is mistaken for the turn-back time, so multipath distortion is reduced. There was a problem that it was impossible to equalize. For example, the digital terrestrial television broadcasting employs the SFN system, and the delay time exceeds a range of ± T _sp (= ± T _u / L), for example, as the number of relay stations installed increases. The number of cases where good reception is interrupted or reception becomes impossible due to arrival of the information has increased. In short, the equalization of the multipath distortion utilizing SP, for placement conditions SP, the delay time can be detected is limited to 1 (± T _sp) following L amount of the effective symbol length T _u. That is, in the OFDM signal received, can only be equalized received signals up to the effective symbol length T _u of L portion of 1 (± T _sp) delay time had the following values.

The present invention has been made in view of the above problems, in the OFDM signal receiving, even when the delay time exceeds the L component of the effective symbol length T _u 1 (± T _sp) The problem is to equalize multipath distortion.

  To achieve the above object, a multipath distortion equalization apparatus according to claim 1 of the present invention equalizes from a multipath wave indicating a received wave including a main wave and a delayed wave that arrive as an OFDM radio wave. Transfer function generation means for generating a transfer function at a target frequency, and time domain data cut out from an input multipath wave at a predetermined cycle to be converted into a frequency domain signal by Fourier transform for each cycle. Multipath distortion equalization apparatus for OFDM signal reception comprising: equalization means for generating a time domain signal of the main wave by performing inverse Fourier transform on a result obtained by dividing each frequency domain signal by the transfer function In this case, the transfer function generation unit includes a first transfer function generation unit and a second transfer function generation unit.

  According to such a configuration, the multipath distortion equalization apparatus is configured such that, in the transfer function generation unit, the scattered pilot data extracted from the input multipath wave by the first transfer function generation unit and the scatter stored in advance. A transfer function at the frequency of the scattered pilot for the propagation path of the delayed wave is generated from the standard value of the tard pilot. Here, since the value of the scattered pilot is determined by the standard, the transfer function at the frequency of the scattered pilot is calculated from the ratio between the scattered pilot value of the multipath wave and the standard value of the scattered pilot. Can be obtained.

  The second transfer function generating means has delay profile measuring means. The delay profile measuring means measures a delay profile by the power spectrum method for each period from time domain data cut out from the input multipath wave at a predetermined period, and determines the delay wave level and the main wave level. Find the ratio and delay time. The power spectrum method is a method that can freely increase the signal period in accordance with the delay time of the measurement target. Therefore, in the power spectrum method, it is possible to detect a long delay time having a longer time width than a measurable time width when the delay time is obtained by a transfer function at the frequency of the scattered pilot. Therefore, no error due to aliasing occurs in the delay time detected by the power spectrum method. Here, since two delay times are detected in the delayed wave in the power spectrum method, it is necessary to specify the true delay time.

For this purpose, the second transfer function generation means obtains a high-frequency phase difference between the main wave and the delay wave from the delay profile, and measures the delay profile into a frequency and time domain function including the transfer function at the frequency of the scattered pilot. When the two delay times having different polarities obtained from the above are substituted, the delay time of the frequency average of the frequency and time domain functions that is theoretically zero and the one that does not become zero is not zero. Is specified as the true delay time. Then, the second transfer function generating means uses the specified true delay time, the ratio between the level of the delayed wave and the level of the main wave, and the high-frequency phase difference to be equalized for the propagation path of the delayed wave. Generate a transfer function at the frequency This transfer function is expressed by, for example, the above formula (1). Therefore, multipath distortion equalizer as transfer function in the frequency of an equalization target was also corresponds to the case where the true delay time exceeds the L component of the effective symbol length T _u 1 (± T _sp) transmission A function can be generated. Then, the multipath distortion equalization apparatus equalizes the multipath wave using this transfer function by the equalizing means. As a result, multipath distortion equalizer can equalize the multipath distortion even when the delay time exceeds the L component of the effective symbol length _{T u 1 (± T sp)} .

  The multipath distortion equalization apparatus according to claim 2 is the multipath distortion equalization apparatus according to claim 1, wherein the transfer function at the frequency of the scattered pilot and the transfer function at the frequency to be equalized are The product of the term representing the main wave, the component of the ratio of the delay wave level to the main wave level, the high-frequency phase difference component of the main wave and the delay wave, and the time component indicating the delay time The second transfer function generation means includes a complex delay wave level generation means, a delay time determination means, and a transfer function calculation means. Here, these transfer functions are expressed by, for example, the aforementioned equation (1).

  According to such a configuration, the multipath distortion equalization apparatus uses the complex delay wave level generation unit of the second transfer function generation unit to convert a term representing a delay wave included in the transfer function at the frequency of the scattered pilot into a delay profile. For each delay time detected when measuring the time, a complex delay wave level indicating a level drop caused by the delay is generated as a function of the frequency and time domain by dividing by the time component indicating the delay time. . Then, the multipath distortion equalization apparatus calculates a statistic including at least an average of the generated complex delay wave levels with respect to the frequency to be equalized by the delay time determination unit, and calculates the calculated complex delay wave level. The delay time that generated the complex delay wave level whose frequency average does not theoretically become zero is specified as the true delay time.

  Here, from the definition of the complex delay wave level, theoretically, the average value of the complex delay wave level does not become 0 in the true delay time, and the complex delay occurs in the delay time that causes an error due to aliasing. The average wave level is zero. Further, as a statistic to be calculated, if only one true delay time can be specified from the delay times obtained theoretically from the measurement of the delay profile, in addition to the average of the complex delay wave level, It may be a statistic such as variance using the average. Then, the multipath distortion equalization apparatus uses the transfer function calculation means to equalize the propagation path of the delayed wave using the identified true delay time and the complex delay wave level that generated the true delay time. Calculate the transfer function at the target frequency. Here, this transfer function is expressed by, for example, the aforementioned equation (1).

  The multipath distortion equalization apparatus according to claim 3 is the multipath distortion equalization apparatus according to claim 2, wherein the delay time determination means includes an average calculation means, a deviation calculation means, a comparison means, It was decided to prepare.

  According to such a configuration, the multipath distortion equalization apparatus calculates, for each complex delay wave level, the average of the complex delay wave level with respect to a change in the frequency to be equalized by the average calculation unit of the delay time determining unit. Then, the multipath distortion equalization apparatus calculates the deviation according to the delay time for generating the complex delayed wave level by using the average of the complex delayed wave level calculated by the average calculating unit by the deviation calculating unit, respectively. To do. Then, the multipath distortion equalization apparatus compares the magnitudes of the two deviations respectively obtained by the deviation calculating means by the comparing means, and determines the delay time that generated the complex delay wave level with the smaller deviation as the true delay time. Is identified as being.

  The multipath distortion equalization apparatus according to claim 4 is the multipath distortion equalization apparatus according to claim 1, wherein the transfer function at the frequency of the scattered pilot and the transfer function at the frequency to be equalized are A term representing the main wave, a component of the ratio of the level of the delay wave to the level of the main wave for each of the plurality of delay waves having a delay or advance with respect to the main wave, and the high-frequency level of the main wave and the delay wave A term representing each delayed wave by the product of the phase difference component and the time component indicating the delay time, and the second transfer function generating means includes an initial phase difference calculating means, a delay time polarity determining means, and a transfer function calculating Means.

  According to such a configuration, the multipath distortion equalization apparatus subtracts the term representing the main wave from the transfer function at the frequency of the scattered pilot by the initial phase difference calculating means of the second transfer function generating means, and then delays the delayed wave. Generate an objective function as a function of frequency and time domain for each scattered pilot frequency by dividing by the time component indicating the absolute value of the detected delay time for each scattered pilot frequency. The added sum is obtained as a frequency average, and an initial phase difference indicating the phase of the delayed wave with respect to the phase of the carrier center frequency of the main wave is calculated from the obtained sum as the high-frequency phase difference of the delayed wave. Then, the multipath distortion equalization apparatus determines whether the value of the sum of the objective functions is 0 by the delay time polarity determination means, and when the value does not become 0, the delay time has a true polarity. If it is determined that the value is 0, it is determined that the polarity is not true. Then, the multipath distortion equalization apparatus uses the transfer function calculation means to detect the component of the ratio between the level of the delayed wave and the level of the main wave detected when measuring the delay profile and the absolute value of the delay time of the delayed wave. Then, using the initial phase difference calculated for the delayed wave and the determination result of the delay time polarity determining means, a transfer function at the frequency to be equalized for the propagation path of the delayed wave is calculated.

  In order to achieve the above object, a receiving apparatus according to claim 5 of the present invention includes a plurality of multipath distortion equalization apparatuses according to any one of claims 1 to 4. The period for extracting time domain data from the multipath wave input to each multipath distortion equalizer is an integer multiple of 3 or more of the effective symbol length, and the timing for extracting each multipath distortion equalizer. However, the receiving apparatus includes a receivable signal extracting unit and a receivable signal combining unit.

  According to such a configuration, the receiving apparatus acquires, by the receivable signal extracting means, the time domain signal of the main wave generated by the equalizing means of each multipath distortion equalizing apparatus for each period, and the effective symbol length. A predetermined number of effective signals satisfying the level of a receivable signal located in the middle except for the first part and the end part of a period that does not satisfy a predetermined receivable signal level among main wave signals having a period that is an integer multiple of The symbol length portion is extracted for each multipath distortion equalizer. Since the receiving apparatus includes receivable signal combining means, a predetermined number of effective symbol length portions of the main signal extracted for each multipath distortion equalization apparatus by the receivable signal extracting means are continuously provided. The main wave signal of the period can be restored. In general, in a multipath wave, when the delay time is relatively long and the delay wave level is close to the main wave level, when the frequency domain signal is returned to the time domain signal, inter-symbol interference causes the first and last ones. CNR (Carrier to Noise Ratio) may deteriorate for symbols. However, in this receiving apparatus, the multipath wave input to the multipath distortion equalization apparatus is cut out at a cycle that is an integral multiple of the effective symbol length, and the signal of the signal equalized by each multipath distortion equalization apparatus. Among them, since the effective symbol length portion satisfying the receivable signal level is synthesized, the low CNR signal is removed from the synthesized signal.

According to the multipath distortion equalization apparatus of claim 1, a delay time longer than the time width measurable by the transfer function at the frequency of the scattered pilot is detected by the power spectrum method, and the long delay time and the scatter time are detected. The multipath wave is equalized using a transfer function corresponding to the delay time whose polarity is determined using the transfer function at the frequency of the tard pilot. Therefore, in the OFDM signal receiving, it is possible to equalize the multipath distortion even when the delay time exceeds the L component of the effective symbol length _{T u 1 (± T sp)} .

  According to the multipath distortion equalization apparatus of the second aspect, the complex delay wave level is obtained by using the transfer function at the frequency of the scattered pilot and the delay time detected when measuring the delay profile. By statistically processing the delay wave level, it is possible to estimate the transfer function after determining the true delay time and perform equalization.

  According to the multipath distortion equalization apparatus according to claim 3, when determining the true delay time from the two delay times detected when measuring the delay profile, in addition to the average of the complex delay wave level, Since the deviation obtained from the average is used, the true delay time can be correctly obtained even if noise is included in the multipath wave.

  According to the multipath distortion equalization apparatus of claim 4, the ratio of the main wave level and the delay wave level and the delay time are obtained by the power spectrum method, and the delay time is measured by the delay profile measurement method using the scattered pilot. Equalization can be performed by obtaining the polarity and the high-frequency phase difference and using these to estimate the transfer function.

According to the receiving apparatus according to claim 5, while equalizing the multipath distortion even when the delay time of the received OFDM signal exceeds 1 (± T _sp) of L portion of the effective symbol length T _u, the CNR Can be increased.

It is a block diagram which shows the structure of the multipath distortion equalization apparatus which concerns on 1st Embodiment of this invention. It is a block diagram which shows the structure of the receiver which concerns on embodiment of this invention. It is a figure which shows an effective symbol length typically. It is a figure which shows typically arrangement | positioning of SP per segment. It is a block diagram which shows the structure of the multipath distortion equalization apparatus which concerns on 2nd Embodiment of this invention. It is a figure which shows typically arrangement | positioning of SP per 13 segments. It is a block diagram which shows the structure of the modification of the receiver which concerns on embodiment of this invention. 5 is a graph showing an initial phase difference detected by the multipath distortion equalizer according to the present invention, where (a) shows a case where the delay time is 20 μs, and (b) shows a case where the delay time is 300 μs. It is a graph which shows the constellation measured on the conditions of 1 delay wave of 20 microsecond delay time using the multipath distortion equalization apparatus which concerns on this invention, Comprising: (a) is before equalization and (b) is after equalization. Respectively. It is a graph which shows the constellation measured on condition of 1 delay wave of 600 microsecond delay time using the multipath distortion equalization apparatus which concerns on this invention, Comprising: (a) is before equalization, (b) is after equalization. Respectively. It is a graph which shows the constellation measured on the conditions of 2 delay waves using the multipath distortion equalization apparatus which concerns on this invention, Comprising: (a) has shown before equalization, (b) has each shown after equalization. . It is a graph which shows CNR after equalization measured on the conditions of delay time delay using the multipath distortion equalizer concerning the present invention. It is a graph which shows CNR after equalization measured on condition of advance delay time using the multipath distortion equalization device concerning the present invention. It is explanatory drawing which shows notionally the method of obtaining the signal output which can be received using two multipath distortion equalization apparatuses of this invention.

  An embodiment for implementing a multipath distortion equalization apparatus and reception apparatus of the present invention will be described in detail with reference to the drawings. Hereinafter, the multipath distortion equalizer according to the first embodiment, the receiver, and the multipath distortion equalizer according to the second embodiment will be sequentially described. It should be noted that the mathematical formulas (formulas (1) to (9)) described in the background art and the summary of the invention will be appropriately incorporated for explanation.

(First embodiment)
[Configuration of Multipath Distortion Equalizer]
The multipath distortion equalization apparatus 1 compensates for distortion due to multipath in OFDM signal reception, and roughly includes a transfer function generation means 2 and an equalization means 3 as shown in FIG. . The multipath distortion equalization apparatus 1 shown in FIG. 1 shows only the components of the reception apparatus 100 shown in FIG. 2, and may include a reception unit such as a reception antenna omitted in FIG. The receiver 100 will be described later.

  The transfer function generation unit 2 generates a transfer function used for equalizing the multipath wave by the equalization unit 3. The transfer function generation unit 2 is roughly divided into a first transfer function generation unit 10 and a second transfer function generation unit 20. And.

<First transfer function generating means>
The first transfer function generation unit 10 generates a transfer function at the frequency of the scattered pilot (SP) in order to generate a transfer function used by the equalization unit 3. The first transfer function generation means 10 generates a transfer function at the SP frequency for the propagation path of the delayed wave from the SP data extracted from the input multipath wave V _r (t). In the present embodiment, as shown in FIG. 1, the first transfer function generation means 10 includes a cutout unit 11, an FFT 12, an SP extraction unit 13, an SP standard value storage unit 14, a division unit 15, a transfer function, And a storage unit 16.

The cut unit 11 cuts the input multipath wave V _r (t) in synchronization with the effective symbol and outputs the cut signal to the FFT 12. The multipath wave V _r (t) input here is a data sequence after A / D conversion of the in-phase component and the quadrature component of the synchronous detection waveform in the time domain. For example, the synchronous detection of the receiving apparatus 100 shown in FIG. This is generated by the unit 120. This multipath wave V _r (t) is similarly input to the equalization means 3 and the delay profile measurement means 30 of the second transfer function generation means 20.

The cutout unit 11 cuts out only the effective symbols from the signal in which effective symbols and guard intervals appear alternately as shown in FIG. 3 and outputs the result to the FFT 12. The cutting time is, for example, about 1 [ms]. In FIG. 3, the effective symbol length T _u of the effective symbol (symbol number 0, 1), the ratio of the length T _g of the guard interval (hatched portion in FIG. 3) was set to 8: 1 (1 symbol The length is 9/8 times the effective symbol length). Similarly, the equalization unit 3 and the delay profile measurement unit 30 also include a cutout unit (not shown) that cuts out the input multipath wave V _r (t), but the cutout period is different and longer. . Details will be described later.

The FFT 12 performs fast Fourier transform on the time domain data of the multipath wave cut out by the cutout unit 11 and outputs the result to the SP extraction unit 13. Then, a demodulated carrier C _r (s, f) is obtained (see equation (2)). The demodulated carrier C _r (s, f) includes an information carrier and SP. The SPs are arranged at a predetermined interval (one for every L carriers) with respect to the change in the frequency f.
The SP extraction unit 13 extracts only SP (SP symbols) from the entire carriers (SP symbols and data symbols) obtained by the FFT 12, and outputs them to the division unit 15. The extracted SP is used as multipath wave data.

  The SP standard value storage unit 14 stores SP standard values, and is composed of, for example, a general memory. As a specific example, the arrangement of SPs per segment in mode 3 is shown in FIG. In FIG. 4, a blank cell symbol indicates a data symbol, and a symbol described as “SP” in a cell indicates a pilot symbol (SP symbol). As shown in FIG. 4, the vertically long band-shaped carriers indicated by the rows are arranged in the horizontal direction. One of the three carriers arranged side by side contains SP, and the remaining three of the two carriers do not contain SP. In addition, in a carrier including SP, one SP is included in four symbols. For this reason, the SP is transmitted at a rate of once every four symbol periods by one of the three carriers. From another point of view, SP is transmitted by one carrier in 12 in each symbol period indicated by a row. In the ISDB-T system, the number of carrier segments is 13, and one frame is composed of 204 symbols. In mode 3, the total number of carriers is 5617, the effective symbol length is 1008 [μs], and the number of carriers per segment is 432. Of these 432, 384 is an information carrier excluding SP, TMCC (Transmission and Multiplexing Configuration and Control) and AC (Auxiliary Channel) which are control signals.

The division unit 15 performs the division shown in the above equation (5) and stores the division result in the transfer function storage unit 16. Specifically, in the calculation of Expression (5), the carrier C _r (s, f) demodulated for the SP extracted by the SP extraction unit 13 is used as the right-hand side numerator. In the calculation of equation (5), the SP standard value stored in the SP standard value storage unit 14 is used as the right-side denominator. The division result by the division unit 15 is a transfer function H (s, f) which is the left side of the equation (5). The frequency interval (data interval) of the transfer function H (s, f) is L times the carrier interval as in SP (corresponding to the horizontal direction in FIG. 4). Therefore, the period of the transfer function H (s, f) here, (corresponding to the vertical direction in FIG. 4) of the L component of the effective symbol length _{T u 1 (T sp = T} u / L) become.
The transfer function storage unit 16 stores a transfer function H (s, f), which is a division result of the division unit 15, and is composed of, for example, a general memory.

<Second transfer function generating means>
The second transfer function generation unit 20 generates a transfer function used by the equalization unit 3 using the transfer function at the SP frequency generated by the first transfer function generation unit 10. The second transfer function generation means 20 obtains a high-frequency phase difference (α) between the main wave and the delay wave from the delay profile, and has a frequency and time including the transfer function (H (s, f)) at the SP frequency. When the two delay times with different polarities obtained from the delay profile measurement are assigned to the function, the delay time in which the frequency average of these frequency and time domain functions is not zero is specified as the true delay time. The transfer function is generated using the specified true delay time, the ratio (R) between the level of the delayed wave and the level of the main wave, and the high-frequency phase difference (α).

  In the first embodiment, as shown in FIG. 1, the second transfer function generation means 20 includes a complex delay wave level generation means 21, a delay time determination means 22, a delay profile measurement means 30, and a transfer function calculation means 40. And. Here, for the convenience of explanation, the outlines of the delay profile measuring means 30, the complex delayed wave level generating means 21 and the delay time determining means 22 are described, and then the complex delayed wave level generating means 21, the delay time determining means 22 and the transmission are transmitted. Details of the function calculating means 40 will be described using mathematical expressions.

<Delay profile measurement means>
The delay profile measuring means 30 measures the delay profile from the input multipath wave V _r (t) by the power spectrum method and obtains the ratio (R) between the delay wave level and the main wave level and the delay time. It is. This delay profile measuring means 30 can be constituted by, for example, processing means of a terrestrial digital SFN wave measuring apparatus described in Patent Document 1. According to this power spectrum method, conjugate complex numbers also appear as impulses representing delayed waves, and therefore, positive and negative delay times with different polarities (delay or advance) are detected as delay times. A positive delay time indicates “delay”, and a negative delay time indicates “advance”.

The delay profile measuring unit 30 is provided with a cutout portion similar to the cutout portion 11 although not shown. However, this not-illustrated cut-out unit is a long-cycle cut-out unit in which the cut-out length (predetermined cycle) of the input multipath wave V _r (t) is an integral multiple of one effective symbol length. That is, as shown in FIG. 3, a signal in which valid symbols and guard intervals appear alternately is cut out. The delay profile measuring means 30 measures the delay profile by collectively performing the Fourier transform processing such as FFT on the corresponding data in a lump with the cycles cut out by the long-cycle cutout unit (not shown).

When the delay time exceeds T _sp / 2, the delay time detected from the delay profile obtained by IFFT of the transfer function H (s, f) at the SP frequency is a folded time. Therefore, when a transfer function corresponding to all frequencies is generated by the conventional method, a delay time that causes an error due to aliasing is used as shown in the above equation (8). Multipath distortion cannot be equalized.

<Outline of Complex Delay Wave Level Generation Unit and Delay Time Determination Unit>
The complex delay wave level generation means 21 sets a term representing a delay wave included in the transfer function H (s, f) at the SP frequency for each delay time detected when measuring the delay profile. By dividing by the time component shown, a complex delayed wave level indicating a level drop caused by delay is generated as a function of frequency and time domain, respectively.

  The delay time determination unit 22 calculates a statistic including an average of the two complex delay wave levels generated by the complex delay wave level generation unit 21 with respect to the frequency to be equalized. When the frequency average is obtained using the complex delay wave level, one is theoretically 0 and the other is theoretically 0. Then, the delay time determination means 22 selects the other (the other) complex delay wave level that does not theoretically become zero, and sets the selected delay time (the other) complex delay wave level as the true delay time. It is specified as

  In the present embodiment, the complex delayed wave level generating means 21 obtains the complex delayed wave level from the following equation (11). Further, in this embodiment, the delay time determining means 22 obtains deviations calculated from the average of each complex delay wave level with respect to the frequency, and determines the delay time generated with the smaller complex delay wave level as the true delay time. It was decided to decide as.

<Details of complex delay wave level generation means>
Here, first, the principle of generating the multiple-loop delayed wave level will be described using mathematical expressions.
The complex delay wave level is Re ^jα . In equation (1), when the second term representing the delay is moved to the left side and divided by the time component (delay time rotator), a complex delayed wave level Re ^jα is obtained as in equation (10). .

Therefore, as shown on the right side of Expression (10), the complex delay wave level can be generated by dividing “transfer function −1” by the rotor having the delay time t _d . However, at the stage of treatment with the complex delayed wave level generating means 21, an unknown true delay time t _d, and two delay times having the detected polarity with the delay profile measuring unit 30 has a candidate ing. Therefore, by using the equation (10), two complex delay wave levels corresponding to two delay times are generated by the following method.

First, H (s, f) is used for the transfer function shown in the numerator on the right side of Equation (10).
In addition, the rotor time is expressed as t as an arbitrary value while corresponding to the delay time. Further, two delay times having the polarity detected by the delay profile measuring means 30 are distinguished as t ₁ and t ₂ . Further, the complex number obtained as the complex delayed wave level by the division of Expression (10) is represented by a binary function form of the time t and the frequency f of the rotor as shown on the left side of Expression (11). .

となる。したがって、２つの複素遅延波レベルＲ（ｔ₁，ｆ）、Ｒ（ｔ₂，ｆ）は、それぞれ式（１２）、式（１３）のように表せる。 R (t ₁ , f) and R (t ₂ , f) are obtained by associating the complex number shown in equation (11) with two delay times t ₁ and t ₂ . Here, assuming that the true delay time is t _d and t ₁ = t _d , t ₂ = −t _d . The numerator on the right side of equation (11) is

It becomes. Therefore, the two complex delay wave levels R (t ₁ , f) and R (t ₂ , f) can be expressed as shown in equations (12) and (13), respectively.

Expression (12) shows a case where a true delay time is given to the rotor. In this case, R (t ₁ , f) is a constant regardless of the change in the frequency f. On the other hand, Expression (13) shows a case where the rotor is given a time that causes an error due to folding. In this case, R (t ₂ , f) is a vector that rotates as the frequency f changes.

  In order to generate two complex delay wave levels by the above-described method, the complex delay wave level generation means 21 includes a subtraction value storage unit 23, a rotor storage unit 24, a division unit 25, as shown in FIG. It has.

  The subtraction value storage unit 23 stores a subtraction value obtained by subtracting “1” from the transfer function H (s, f) at the SP frequency generated by the first transfer function generation unit 10. Consists of memory and the like. “H (s, f) −1” represents a numerator on the right side of the above-described formula (11).

The rotor storage unit 24 stores the rotor e ^{−j2πft / N} that changes the frequency f of the signal, and includes, for example, a general memory. Here, the rotor storage unit 24 stores two delay times t ₁ and t ₂ having the polarity detected by the delay profile measuring means 30.

The division unit 25 similarly divides the subtraction value stored in the subtraction value storage unit 23 by the two rotors stored in the rotor storage unit 24. That is, the division unit 25 performs the calculations of the above equations (12) and (13). The obtained quotient and the two complex delay wave levels R (t ₁ , f) and R (t ₂ , f) are sent to the delay time determining means 22.

<Details of delay time determination means>
In order to determine the delay time by comparing the magnitudes of the deviations calculated from the average frequency of the complex delay wave level, the delay time determining means 22, as shown in FIG. 1, an average calculation unit 26, a deviation calculation unit 27, And a comparison unit 28.

Here, since the complex delay wave levels R (t ₁ , f) and R (t ₂ , f) vary depending on the delay time, a generalized delay time representing the delay time (t ₁ or t ₂ ) is expressed as t. It shall be denoted by _i (i = 1, 2). The delay time determining means 22 specifies which of t ₁ and t ₂ is the true delay time by using the above-described equations (12) and (13).

The average calculator 26 calculates the average of the complex delay wave level with respect to the change in frequency. In the present embodiment, the average calculation unit 26 calculates the average A _i of the complex delay wave level with respect to the change in the frequency f for each delay time t _{i according} to the equation (14).

Here, n is a data number, L is an SP carrier interval, and f ₀ is a carrier width. That is, there is a relationship of f = nLf ₀ with the frequency f. Incidentally, when the total number of data is N, the total number of data N _SP in the frequency domain becomes N / L.

The average of R (t ₁ , f) with respect to the change of the frequency f = nLf ₀ is Re ^jα as shown in the above equation (12), and the average of R (t ₂ , f) is the above equation (13 ), As shown in FIG. Therefore, it is possible to specify the true delay time from this difference. However, since there is actually noise, it may be difficult to make a clear determination. Therefore, in this embodiment, the deviation is determined from this average.

The deviation calculation unit 27 uses the respective averages A _i calculated by the average calculation unit 26 to obtain the deviations D _i corresponding to the delay times t _i using the equation (15). An average deviation can be used instead of the standard deviation.

The comparator 28 compares the two deviations (D ₁ and D ₂ ) obtained from the equation (15) with respect to the delay time t _i , and generates the complex delay wave level with the smaller deviation. Is determined to be the true delay time t _d . On the other hand, the delay time in which the complex delay wave level with the larger deviation is generated is a time at which an error due to aliasing occurs. The deviation on the time side that causes this error is much larger than the noise. Therefore, according to the present embodiment, even in the presence of noise, the difference between the two deviations is clear, and a large and small comparison determination is easy, and the determination is not erroneous.

Further, the delay time determining means 22 determines the complex delay wave level with the smaller deviation as the complex delay wave level. The true delay time t _d determined by the delay time determining means 22 and the complex delay wave level are output to the transfer function calculating means 40.

<Transfer function calculation means>
The transfer function calculation means 40 uses the true delay time specified by the delay time determination means 22 and the complex delay wave level that generated the true delay time, and uses the transfer function at each frequency for the propagation path of the delay wave. H (f) is calculated. In the present embodiment, the transfer function calculation means 40 uses the complex delay wave level Re ^jα with the smaller deviation and the delay time (true delay time t _d ) that generated the complex delay wave level to calculate the transfer function. The above equation (1) is generated by calculation on the right side. The transfer function H (f) generated for each frequency f is output to the equalizing means 3.

<Equalization means>
The equalizing means 3 converts the input multipath wave V _r (t) into a frequency domain signal, and the converted frequency domain signal C _r (f) is transferred to the transfer function generated by the transfer function calculating means 40. The result divided by H (f) is subjected to inverse Fourier transform to generate a time domain signal C _m (f) of the main wave of the multipath wave. In the present embodiment, the equalizing means 3 performs an FFT to _cut the input multipath wave V _r (t) at a predetermined period T _cut and convert the cut time domain signal into a frequency domain signal. The inverse Fourier transform is performed by IFFT.

As shown in FIG. 1, the equalization unit 3 includes an FFT 51, a division unit 52, and an IFFT 53. Although illustration is omitted, a cutout portion similar to the cutout portion 11 is provided in the preceding stage of the FFT 51. However, this not-illustrated cut-out unit is a long-cycle cut-out unit in which the cut-out length (predetermined cycle) of the input multipath wave V _r (t) is an integral multiple of one effective symbol length. That is, as shown in FIG. 3, a signal in which valid symbols and guard intervals appear alternately is cut out. The equalizing means 3 performs processing such as FFT on the corresponding data in a lump with the cycles cut out by the long-cycle cutout portion (not shown) in this way, and equalizes multipath distortion.

If the cutout period in this case is p times one effective symbol length, the frequency interval of the spectrum is f ₀ / p when the fundamental frequency is f ₀ . Here, the value of the integer p is an arbitrary integer of at least 3 in order to remove the front and rear and leave the middle. For example, in the case of a mode 3 signal, the effective symbol length is 1008 [μs]. If p = 32 and the cutout length is about 32 [ms], a high CNR signal can be expected.

The reason why it is preferable to connect a large number of symbols in this way is as follows. When the delay time is relatively long such as about 1 [ms] and the level of the delayed wave is close to the level of the main wave, when the frequency domain signal is returned to the time domain signal, it is caused by intersymbol interference. The CNR may deteriorate for the first and last symbols. However, if the effective symbol length is 32 times, a good signal can be obtained with the remaining symbols, and therefore a method of obtaining a continuous high CNR signal can be achieved by using a plurality of this method. In order to ignore the error of the ratio of the delay wave level / main wave level, it is preferably 16 times or more. Below, the cut length in the equalization means 3 is described with _Tcut (= p * _Tu ). Note that the delay profile measuring unit 30 also includes a not-shown cutting unit that cuts the input multipath wave V _r (t) at the same cycle as the equalizing unit 3.

The FFT 51 performs FFT on the multipath wave _cut out with the period T _cut and converts it into a spectrum, and outputs the received wave C _r (f) indicating the numerator in the above-described equation (3).

The division unit 52 performs the calculation of the above-described equation (3). Specifically, the division unit 52 uses C _r (f) output from the FFT 51 for the right side numerator of Expression (3), and the transfer function calculation means 40 generates the right side denominator of Expression (3). The transferred transfer function H (f) is used. As a result, a spectrum C _m (f) equivalent to the main wave equalizing the multipath distortion is obtained as a quotient of division.

The IFFT 53 performs IFFT on the spectrum C _m (f), which is a quotient of division, and returns it to a time domain signal having the same period (period T _cut ) as the input, and outputs a multipath distortion equalization signal V _eq (t).

  The first transfer function generation means 10, the second transfer function generation means 20, and the equalization means 3 described above can be configured by, for example, a semiconductor memory or an electric circuit, and are realized by processing a program. You can also.

[Operation of Multipath Distortion Equalizer]
Next, the operation flow of the multipath distortion equalization apparatus will be described with reference to FIG.
The multipath distortion equalization apparatus 1 includes a multipath wave that is input as a data sequence after A / D conversion of the in-phase component and the quadrature component of the synchronous detection waveform in the time domain by the clipping unit 11 in the first transfer function generation unit 10 V _r (t) is cut out in synchronization with the effective symbol (short cycle). The FFT 12 performs fast Fourier transform on the cut data and outputs a demodulated carrier C _r (s, f)), and the SP extraction unit 13 extracts only SP from the entire carrier. The division unit 15 divides the SP C _r (s, f) extracted by the SP extraction unit 13 by the SP standard value based on the above-described equation (5), thereby obtaining the transfer function H (s, f). obtain.

On the other hand, in the multipath distortion equalization apparatus 1, the delay profile measurement unit 30 of the second transfer function generation unit 20 uses a cycle (long cycle) that is an integral multiple of the effective symbol length from the input multipath wave V _r (t). A delay profile is measured by the power spectrum method using the cut data, and two delay times t ₁ and t ₂ having polarity are detected. Two delay times t ₁ and t ₂ are detected for each delay wave.

Then, the multipath distortion equalization apparatus 1 uses the complex delay wave level generation means 21 to complex delay wave levels R (t ₁ , f) and R (t ₂ , f corresponding to the _two delay times t ₁ and t _2. ) Respectively. Here, the complex delayed wave level generation means 21 converts the subtraction value stored in the subtraction value storage unit 23 by the division unit 25 based on the equations (12) and (13) described above, to the rotor storage unit 24. Similarly, each of the two rotors stored in is divided.

Subsequently, the multipath distortion equalization apparatus 1 uses the delay time determination unit 22 to calculate the two delay times t ₁ and t ₂ from the complex delay wave levels R (t ₁ , f) and R (t ₂ , f). one of out specified as true delay time t _d. Here, the delay time determination unit 22 calculates the average A _i of the complex delay wave level with respect to the change in frequency based on the above-described equation (14) by the average calculation unit 26. Then, the deviation calculating unit 27 obtains the deviation D _i according to the delay time t _i using each average A _i based on the above-described equation (15). Further, the comparator 28 identifies the delay time that generated the complex delay wave level with the smaller deviation as the true delay time t _d .

Subsequently, the multipath distortion equalization apparatus 1 uses the complex delay wave level Re ^jα with a smaller deviation and the true delay time t _d by the transfer function calculation means 40 to calculate the right side of the above equation (1). By calculating for each frequency f, a transfer function H (f) is generated and output to the equalizing means 3.

Then, the multipath distortion equalization apparatus 1 uses the equalization means 3 to receive data obtained by the FFT 51 using data cut from the multipath wave V _r (t) at a period (long period) that is an integral multiple of the effective symbol length. Using the wave C _r (f) and the transfer function H (f) generated by the transfer function calculating means 40, the division unit 52 performs the calculation of the above-described equation (3), and the IFFT 53 performs the inverse Fourier transform. To do. As a result of the above operation, the multipath distortion equalization apparatus 1 equalizes multipath distortion from the multipath wave V _r (t).

[Receiver]
2 receives, for example, a terrestrial digital television broadcast, and includes a multipath distortion equalization apparatus 1, a frequency conversion unit 110 and a synchronous detection unit 120 provided as a reception unit in the preceding stage. And a receiver 130 provided in the subsequent stage.

  When the antenna reception signal from the reception antenna 140 is supplied, the frequency conversion unit 110 down-converts the OFDM signal (multipath wave) of the selected channel into an intermediate frequency signal (IF (Intermediate Frequency) signal). Is.

The synchronous detection unit 120 performs quadrature synchronous detection on the IF signal converted by the frequency conversion unit 110, and multi-paths a data sequence obtained by performing A / D conversion on the in-phase component and the quadrature component of the obtained synchronous detection waveform in the time domain. The input data V _r (t) is input to the multipath distortion equalization apparatus 1. The in-phase component of the A / D converted data string indicates the real number axis component (I data) of the symbol (carrier symbol), and the quadrature component indicates the imaginary number axis component (Q data) of the symbol.

The receiver 130 decodes the multipath distortion equalization signal V _eq (t) demodulated and equalized by the multipath distortion equalization apparatus 1 and displays the decoded video signal and audio signal.

According to the multipath distortion equalization apparatus 1 and the receiving apparatus 100 of this embodiment, the delay times t ₁ and t ₂ longer than the time width measurable by the transfer function H (s, f) at the SP frequency are used as the power spectrum. Using the transfer function H (f) corresponding to the delay time t _d detected by the method and the polarity determined using the long delay times t ₁ and t ₂ and the transfer function H (s, f) at the SP frequency. To equalize multipath waves. Therefore, in the OFDM signal receiving, it is possible to equalize the multipath distortion even when the delay time t _d has exceeded the effective symbol length T _u of L portion of 1 (± T _sp).

In the delay profile measurement of the power spectrum method, as described in Non-Patent Document 1, even if the delay time exceeds the range of ± 1 [ms], the absolute value of the delay time and the delay wave level can be measured. . Therefore, the multipath distortion equalization apparatus 1 according to the present embodiment can perform multipath distortion equalization regardless of the level of the delayed wave even when a multipath wave of about 1 [ms] is received, for example. Can be received. This is because, in ISDB-T system, a third of the effective symbol length T _u 1 (336 in total [μs]: ± 168 [μs ]) are well beyond the scope of.

(Second Embodiment)
In the embodiment, the Fourier transform for converting the time domain signal into the frequency domain signal is performed by the fast Fourier transform (FFT). However, when the speed is not important, a discrete Fourier transform (DFT) is used. ). Further, in the above embodiment, for the sake of simplicity, one delay wave is formulated with respect to the main wave, but a plurality of delay waves can be similarly formulated. Furthermore, it is possible to analyze the received wave including noise. Hereinafter, such a case will be described as a second embodiment.

[Outline of Multipath Distortion Equalizer Configuration]
A configuration of a multipath distortion equalization apparatus 1A of the second embodiment is shown in FIG. The same configurations as those of the multipath distortion equalization apparatus 1 shown in FIG. 1 are denoted by the same reference numerals, description thereof will be omitted as appropriate, and differences will be described. A reception wave (v _r (t)) received by the multipath distortion equalization apparatus 1A as a multipath wave is a main wave (hereinafter referred to as a desired wave), a plurality of delay waves, and additive white Gaussian noise (AWGN: Additive White Gaussian Noise (hereinafter referred to as Gaussian noise).

As shown in FIG. 5, the multipath distortion equalization apparatus 1 </ b> A roughly includes a transfer function generation unit 2 </ b> A, an equalization unit 3 </ b> A, and a long period cutout unit 70. The long-period cutting unit 70 cuts the input multipath wave V _r (t) at a length (long cycle) that is an integral multiple of one effective symbol length, and outputs it to the transfer function generation means 2A and equalization means 3A. . Specifically, in the case of an OFDM signal in mode 3 that is actually operated, the signal is cut out by 32 times (approximately 32 [ms]) of the effective symbol length (1.008 [ms]). Note that, as in the first embodiment, the long-period cutout unit 70 may be individually provided inside the transfer function generating unit 2A and the equalizing unit 3A. Further, the multipath distortion equalization apparatus 1A shown in FIG. 5 shows only the components of the receiving apparatus 100 shown in FIG. 2, and may include a receiving unit such as a receiving antenna omitted in FIG.

  The transfer function generating means 2A generates a transfer function used for equalizing the multipath wave by the equalizing means 3A, and is roughly divided into a first transfer function generating means 10A and a second transfer function generating means 20A. And.

  In the second embodiment, the first transfer function generation means 10A is different in that it includes a DFT 12A that performs discrete Fourier transform, as shown in FIG. On the other hand, as shown in FIG. 5, the second transfer function generating means 20A includes a delay profile measuring means 30, a transfer function calculating means 40A, and a transfer function analyzing means 60.

<Transfer function analysis means>
The transfer function analyzing means 60 detects the initial phase difference using the SP data extracted by the first transfer function generating means 10A and the SP standard value, and determines the polarity of the delay time. Phase difference detection means 61 and delay time polarity determination means 62 are provided.

The initial phase difference detecting means 61 detects a phase difference (hereinafter referred to as an initial phase difference) at the carrier center frequency between the desired wave and the delayed wave from the SP data, and outputs it to the transfer function calculating means 40A.
The delay time polarity determining means 62 determines whether the polarity of the delay time is delayed or advanced from the SP data, and outputs the determination result to the transfer function calculating means 40A as the delay time polarity.

  The delay profile measuring means 30 outputs the ratio of the delayed wave level to the desired wave level (delayed wave level / desired wave level) measured by the power spectrum method and the absolute value of the delay time to the transfer function calculating means 40A. .

  The transfer function calculating unit 40A generates a transfer function (H (f)) by combining the initial phase difference, the delay time polarity, the delay wave level / desired wave level, and the absolute value of the delay time.

<Equalization means>
In the second embodiment, the equalization means 3A includes a DFT 51A that performs discrete Fourier transform, a division unit 52A, and an IDFT 53A that performs inverse discrete Fourier transform, as shown in FIG. In the case of an OFDM signal in mode 3, the equalizing means 3A performs DFT (Discrete Fourier Transform) on the received wave v _r (t) clipped by 32 times the effective symbol length (1.008 [ms]), and then receives the signal C _The equalized time domain is obtained by dividing _r (f) by the transfer function H (f) obtained by the transfer function calculating means 40A and further performing IDFT (Inverse Discrete Fourier Transform) on the signal C _eq (f). Desired wave v _cq (t) is obtained.

[Equalization principle of multipath distortion equalizer]
The principle of equalization of the multipath distortion equalization apparatus 1A of the second embodiment using a specific example of the OFDM signal in the actually operated mode 3 and the mathematical expressions (formulas (21) to (24)). Will be described.

As shown in FIG. 3, the effective symbol length (T _u ) of the OFDM signal in mode 3 is 1.008 [ms], and the number of effective symbol data (N _u ) is 8192. Hereinafter, the number of valid symbol data is represented as _Nu . The sampling interval is about 0.123 [μs] (= T _u / N _u ), and the guard interval length (T _g ) is 126 [μs]. In the ISDB-T method, SPs are effectively arranged at a ratio of 1 to 3 (L = 3) in the frequency direction, and ± 168 [μs] (336 [μs] in total, minus is advanced) If it is a delay time until (means)), no problem occurs.

The equalizing means 3A performs equalization in the frequency domain, and the cut-out time for obtaining the received wave in the frequency domain is an integral multiple of the effective symbol length (for example, p = 32 times). Therefore, the total number of data in this cut-off time is p × N _u .

In FIG. 5, if the time domain signal of desired wave + delayed wave (s) is v _i (t), the frequency domain received signal (C _r (f)) after DFT is expressed by equation (21). .

Here, t indicates the time normalized by the sampling interval. Specifically, t is expressed by time / sampling interval (T _u / N _u ). Further, f is a carrier number for discriminating a carrier, and is defined by a frequency normalized by a discrete frequency interval, that is, a frequency / discrete frequency interval. Specifically, when the effective symbol length is 1.008 [ms] and p = 32, the reciprocal of the effective symbol length is about 0.992 [kHz]. In this case, the discrete frequency interval is about 0.992. [KHz] / 32. If p = 1, the discrete frequency interval is represented by about 0.992 [kHz].

In Equation (21), N _w (f) is a frequency domain signal of Gaussian noise and noise voltage due to intersymbol interference caused by the delay time exceeding the guard interval. Intersymbol interference is described in Non-Patent Document 1.

On the other hand, the transfer function (H (f)) when M delayed waves (identification numbers m = 1, 2,..., M) arrive can be obtained by extending the above-described equation (1) to obtain the equation (22). ). Here, R _m indicates the m-th delayed wave level / desired wave level. Further, t _dm indicates a delay time standardized by the sampling interval for the m-th delayed wave. Specifically, t _dm is expressed by delay time / sampling interval (T _u / N _u ). Further, α _m represents an initial phase difference between the desired wave and the m-th delayed wave.

The equalized frequency domain desired wave (C _eq (f)) can be obtained by dividing C _r (f) by H (f). expressed.

The desired wave (v _cq (t)) in the time domain after equalization can be obtained by IDFT of Expression (23) and is expressed by Expression (24).

Here, n _w (t) is a noise voltage included in the desired wave (v _cq (t)) in the time domain after equalization. As shown in Expression (24), noise n _w (t) exists even in the time domain after equalization. However, if the influence of the noise n _w (t) on the bit error rate is the same as that of the Gaussian noise, if the signal power ratio (S / N ratio) is 21 [dB] or more, the determination process and error correction can be applied to the bit. It is known that error-free reception is possible. Here, the S / N ratio is expressed by “power of v (t) / power of n _w (t)”. The condition where the S / N ratio is 21 [dB] or higher indicates a CNR condition where the bit error rate of Viterbi decoding is 2 × 10 ⁻⁴ during 64QAM modulation used for high-definition broadcasting (NHK Reception Technology Center). (See page 48, “Digital Terrestrial Broadcasts to Know”).

However, as apparent from the equation (22), in order to obtain the transfer function H (f) when M delayed waves arrive, R _m , t _dm , α _{m for} each delayed wave (identification number m). Need to know. However, in the delay profile measurement using SP, a delay time exceeding ± 168 [μs] cannot be measured. On the other hand, in the delay profile measurement of the power spectrum method, the absolute value of the delay time and the delay wave level can be measured even when the delay time exceeds the range of ± 1 [ms]. Therefore, among parameters that need to be known, R _m and | t _dm | are obtained by an algorithm of the power spectrum method, and the remaining α _m and delay polarity (delay or advance) are described later. I asked for it.

[Detection method of initial phase difference between desired wave and delayed wave]
Next, as an analysis principle of the initial phase difference detection unit 61 of the multipath distortion equalization apparatus 1A, an initial phase difference detection method for an arbitrary m-th delayed wave among M delayed waves is expressed by a formula (formula (formula ( 25) to formula (34)).

<Detection method of initial phase difference>
The initial phase difference (α _m ) is obtained using SP. As shown in FIG. 6, the SPs are intermittently arranged at a rate of one for every 12 symbols in the frequency direction and one for every 4 symbols in the time symbol direction. Here, K is 5616 (total number of carriers (5617) −1), and −K / 2 ≦ k ≦ K / 2 represents a band. N _u is the total number of data and at the same time the sampling frequency / discrete frequency interval. 3 represents one segment, FIG. 6 represents 13 segments.

Here, it is assumed that the symbol number in the time direction is represented by n and the carrier number in the frequency direction is represented by k (discrete frequency interval = 0.92 [kHz]). In this case, the frequency domain signal (S _r (n, k)) in an arbitrary symbol of the received wave is expressed by Expression (25). In the above equation (21), f (discrete frequency interval = 0.92 [kHz] / 32) is used as the carrier number, but here, the short cycle cut-off time is the effective symbol length (1.008 [ms]). Therefore, the carrier number is set to k.

In Expression (25), for example, in mode 3, N _u = 8192. For symbol numbers n = 0, 1, 2,..., T ₀ = n × (9N _u / 8). Here, the coefficient 9/8 indicates that the symbol length including the guard interval is 9/8 times the effective symbol length. Further, since S _r (n, k) shown in Expression (25) is a frequency domain signal in an arbitrary symbol, the signal shown in Expression (25) includes a data symbol and an SP symbol. However, only SP is used to detect the initial phase difference (α _m ). Therefore, k in the following formula is handled only by k where SP exists. For example, the sum (sigma) for k means to do only for SP symbols.

  In order to obtain the transfer function, the SP of the desired wave is required in addition to the SP of the received wave. If the frequency domain signal of the SP symbol of the desired wave (main wave) is S (n, k), and the sum of the Gaussian noise voltage and the noise voltage caused by intersymbol interference is N (n, k), Equation (26) becomes It holds.

Here, R _rm is a delay wave level that decreases due to intersymbol interference caused by delay or advance of the delay time. T _dm is the delay time of the m-th delay wave that is normalized by the sampling interval.

S _r (n, k) / S (n, k) obtained by dividing S _r (n, k) in equation (26) by S (n, k) is a transfer function detected using SP, If this is _expressed as H _r (k), it is expressed by equation (27). This H _r (k) corresponds to the output of the first transfer function generation means 10 shown in FIG. Note that the frequency domain signal S (n, k) of the SP symbol of the desired wave is stored in the SP standard value storage unit 14.

When R _rm e ^jαm is ^obtained from Equation (27), there is a noise term (a function of n, k), and therefore it cannot be obtained with accuracy only by the value at k. Therefore, by averaging as follows, the error due to the noise term is removed and R _rm e ^jαm is obtained.

で除算した周波数（ｋ）および時間領域の関数を生成し、これを式（２８）に示す目的関数Ａ_m（ｋ）として定義する。 Specifically, a case where R _rq e ^jαq of an arbitrary q-th delayed wave among M delayed waves is ^obtained will be described. In this case, after subtracting 1 from the transfer function H _r (k) shown in Expression (27) using the relational expression of Expression (27),

A frequency (k) and time domain function divided by is generated, and this is defined as an objective function A _m (k) shown in Equation (28).

The total sum (frequency average) of A _m (k) obtained by adding all carriers is expressed by Expression (29).

は、ｋの変化に対して、複素平面上で半径１の円周上を回転する。したがって、この第１項のｋについての和のうち、ｔ_dm≠ｔ_dqの場合の平均（加算の結果）は０となる。一方、式（２９）の右辺カッコ内の第１項のｋについての和をとるときに、ｔ_dm＝ｔ_dqの場合の

は、すべてのｋにおいて１となる。また、式（２９）の右辺カッコ内の第２項においては、ｔ_dqは０ではない。そのため、第１項のｔ_dm≠ｔ_dqの場合と同様に、第２項のｋについての和は常に０となる。以上のことから、式（３０）の関係が得られる。なお、式（３０）において、ｍ＝ｑである。 In the first term in the right parenthesis of equation (29),

Rotates on the circumference of radius 1 on the complex plane with respect to the change of k. Therefore, the average (result of addition) in the case where t _dm ≠ t _dq is 0 among the sums of k in the first term. On the other hand, when taking the sum of k in the first term in the right parenthesis of Equation (29), the case of t _dm = t _dq

Becomes 1 at all k. Further, in the second term in the right parenthesis of Expression (29), t _dq is not 0. Therefore, as in the case of t _dm ≠ t _dq in the first term, the sum of k in the second term is always 0. From the above, the relationship of Expression (30) is obtained. In the formula (30), m = q.

Here, the sum coefficient is K / 12 on the rightmost side of Equation (30) because addition is performed only by SP, and one SP is arranged in 12 in the frequency direction. is there. If this equation (30) is modified, equation (31) is obtained. Since the objective function A _m (k) shown in the above equation (28) is obtained from the observed value and the specified value, Re (real part) and Im (imaginary part) are obtained as specific complex numbers.

From the equation (31), the initial phase difference α _m can be obtained from the equation (32).

Next, a delay time in which the initial phase difference α _m can be detected is obtained. Here, the time (t _i ) of intersymbol interference (mixed with different symbols) normalized by the sampling interval is expressed by Expression (33).

Here, t _g is the guard interval length normalized by the sampling interval, and t _dm is the delay time of the m-th delayed wave normalized by the sampling interval.

If N _u is the number of samplings in the effective symbol period, the m-th delayed wave level / desired wave level (R _rm ) is expressed by Expression (34) (see Non-Patent Document 1). Note that R _m is a measured value obtained by the delay profile measuring means 30, and the delayed wave level R _rm itself that decreases due to inter-symbol interference corresponds to a part of R _{m in} a region where inter-symbol interference occurs.

The range of the delay time in which the initial phase difference α _m can be detected is 1−t _i / N _u > 0 from the equation (34). At this time, when the delay time indicates “delay”, t _dm <N _u + t _g from the above equation (33) and equation (34). That is, the initial phase difference α _m can be detected if the delay time is less than N _u + t _g .

Similarly, when the delay time indicates “advance”, the range of the delay time in which the initial phase difference α _m can be detected is 1−t from the equation below equation (33) and equation (34). _dm / N _u > 0. From this condition, t _dm <N _u . That is, the initial phase difference α _m can be detected if the delay time is less than _Nu .

[Delay time polarity detection method]
Next, as a principle of analysis of the delay time polarity determination means 62 of the multipath distortion equalization apparatus 1A, a delay time polarity (delay, advance) detection method in an arbitrary m-th delay wave out of M delay waves Will be described.

The delay time polarity determining means 62 determines the polarity (delay / advance) of the delay time t _dm . The principle is as follows.
In the summation formula of A _m (k) shown in the above equation (29), when the delay time t _dq of the q-th delayed wave is true polarity (when t _dm = t _dq ), this A _m (k) Has a value shown in the above-described equation (30).
On the other hand, in the case of a delay time having a reverse polarity (t _dm = −t _dq ), the summation formula of A _m (k) is zero.
Therefore, for example, the calculation can be performed by performing the calculation of the above-described formula (29) for two cases of ± t _dq and comparing the calculation results.

[Operation flow of multipath distortion equalizer]
The operation flow of the multipath distortion equalization apparatus 1A of the second embodiment is the same as the operation flow of the multipath distortion equalization apparatus 1. However, the following operations (1) to (3) are mainly different.

(1) The initial phase difference detection means 61 subtracts the term representing the main wave from the transfer function at the SP frequency, and then divides the result by the time component indicating the absolute value of the delay time detected for the delayed wave. An objective function is generated for each frequency, a sum obtained by adding the objective function with respect to each SP frequency is obtained, and an initial phase difference is calculated as a high-frequency phase difference of the delayed wave from the obtained sum.

(2) The delay time polarity determination means 62 determines whether the value of the sum of the objective functions is 0. If the value does not become 0, the delay time polarity determination unit 62 determines that the delay time polarity is a true polarity, When the value is 0, it is determined that the polarity is not true.

(3) The transfer function calculating unit 40A determines the component (R) of the ratio between the level of the delayed wave and the level of the main wave detected when measuring the delay profile and the absolute value (| t of the delay time of the delayed wave _d |), the initial phase difference (α) calculated for the delay wave, and the determination result (true polarity) of the delay time polarity determination means 62, and the equalization target for the propagation path of the delay wave The transfer function at the frequency to be calculated is calculated.

  According to the second embodiment, the absolute value of the delayed wave level / desired wave level and delay time obtained by using the power spectrum method, the initial phase difference between the desired wave and the delayed wave, and the polarity (delay, advance) of the delay time. ) To generate a transfer function and divide the received wave, so that multipath distortion can be equalized.

  Further, in the conventional reception method, when the delay time exceeds the guard interval length or ± 168 [μs], reception may be difficult or impossible, but according to the second embodiment, as shown in the examples described later. In addition, since reception is possible even when a delay wave (plurality) having a delay time of about 1 [ms] and a desired wave level / delay wave level of about 3 [dB] is mixed, interference due to an increase in future SFN stations It is very effective in improving

(Modification)
As mentioned above, although each embodiment was described, this invention is not limited to these, In the range which does not change the meaning, it can implement variously. For example, the receiving apparatus 100 has been described as including a single multipath distortion equalization apparatus 1 (or 1A), but may be configured to include a plurality of multipath distortion equalization apparatuses. FIG. 7 shows a case where two multipath distortion equalization apparatuses 1A (1a, 1b) are provided in parallel, and the reception apparatus 100B includes these two multipath distortion equalization apparatuses 1a, 1b. . Here, the timing at which time-domain data is _cut out with the period T _cut (=) from the multipath wave V _r (t) input to the multipath distortion equalizers 1a and 1b is different. If the multipath distortion equalization apparatus 1a cuts out at the timing of “t = 0, 32, 64,...”, For example, the multipath distortion equalization apparatus 1b has “t = 16, 48, 80,. Cut out at the timing.

The reason for providing a plurality of units in this way is as follows. As described in each embodiment, when a multipath wave is cut out at a cycle that is an integral multiple of the effective symbol length, a signal in which effective symbols and guard intervals appear alternately is obtained. In the multipath wave received by the receiving apparatus, when the delay time is relatively long and the delay wave level is close to the main wave level, the frequency domain signal is timed by the IDFT 53A of the equalizing means 3A (see FIG. 5). When returning to the area signal, the CNR may deteriorate for the first and last symbols due to intersymbol interference. That is, among the periods (= T _u × p = about 32 [ms]) of the multipath distortion equalization signal V _eq (t) output from the multipath distortion equalization apparatus 1A, the first and last ones have a CNR. May deteriorate.

  Therefore, in this modification, the receiving device 100B includes a receivable signal extracting unit 150 and a receivable signal combining unit 160 between the multipath distortion equalizing device 1A (1a, 1b) and the receiver 130. I decided to prepare.

Receivable signal extraction section (receivable signal extracting means) 150, each multipath distortion equalizer 1A The generated signal V _eq (t) is acquired for each period T _cut, the period T _cut signal V _eq ( Multipath distortion equalization of a predetermined number of effective symbol length portions satisfying the receivable signal level located in the middle except the first portion and the end portion not satisfying the predetermined receivable signal level in t) Each device 1A is extracted and output to the receivable signal synthesizer 160. Here, the receivable signal level indicates a level where the CNR is larger than 20 [dB].

  The receivable signal synthesizer (receivable signal synthesizer) 160 synthesizes the signals satisfying the receivable signal levels extracted by the receivable signal extraction unit 150 so as to be continuous, and outputs them to the receiver 130. Specifically, when each of the multipath distortion equalization apparatuses 1a and 1b is referred to as an equalizer 1 and an equalizer 2 in the receiving apparatus, the receivable signal combining unit 160 outputs the signal from the equalizer 1. Next to the signal that satisfies the receivable signal level (the part indicated by “a” in FIG. 14), the signal that satisfies the receivable signal level output by the equalizer 2 (the part indicated by “b” in FIG. 14). Combining is performed continuously, and further, a portion indicated by “a” and a portion indicated by “b” are alternately synthesized. Thereby, it is possible to obtain a continuous signal in which CNR can be received for all symbols.

  Note that the number of multipath distortion equalization apparatuses 1A in the receiving apparatus 100B may be three or more as long as the multipath wave input to the receiving apparatus 100B can be synthesized by shifting the timing at which the multipath waves are cut in a long cycle. . Further, instead of the multipath distortion equalization apparatus 1A, a configuration may be adopted in which a plurality of multipath distortion equalization apparatuses 1 are provided in the receiving apparatus 100B.

In order to confirm the effect of the present invention, a computer simulation for verifying the performance of the multipath distortion equalizer of the present invention was performed. Here, three experiments (Experiment 1, Experiment 2, Experiment 3) were roughly performed on the multipath distortion equalization apparatus 1A according to the second embodiment. Experiment 1 is an experiment to verify the detection performance of the initial phase difference (α _m ), Experiment 2 is an experiment to measure the constellation before and after equalization, and Experiment 3 is an experiment to measure the CNR after equalization. In the graph of the experimental results, only representative examples of experimental results are shown in which the experiment was performed while appropriately changing the conditions for the case where the delayed wave was one wave and the case where the delayed wave was two waves.

(Experimental conditions assumed in each experiment)
The cutting time (pT _u ) was 32 times the effective symbol length (about 32 [ms]).
Since the length of one symbol is 9/8 times the effective symbol length, the number of symbols obtained from the cut time of 32 [ms] is 27.

(Experiment 1: Detection of initial phase difference (α _m ))
<Experimental conditions>
When the delay wave is 1 wave and the delay time is 20 [μs], when the carrier number (k) is changed, the instantaneous value of the objective function A _m (k) shown in the equation (28), An experiment was performed to obtain the complex delayed wave level R _rm e ^jαm (average value) shown in the above equation (31).
Further, the experiment was performed in the same manner by changing the delay time to a condition of 300 [μs].

At this time, the CNR of the input signal is 50 [dB].
Signal conditions when the delay wave is one wave are as follows.
R ₁ (delay wave level / desired wave level) = − 6 [dB]
α ₁ (initial phase difference between desired wave and delayed wave) = 45 °

<Experimental result>
FIG. 8A shows experimental results on a complex plane when the delay time is 20 [μs]. FIG. 8B shows the experimental results when the delay time is 300 [μs]. In each graph, the horizontal axis represents the real axis, and the vertical axis represents the imaginary axis.

When the delay time is 20 [μs], there is no noise due to inter-symbol interference. Therefore, the second noise component term in the parentheses on the right side of the equation (29) is almost O (input signal) for all k. Only Gaussian noise). Therefore, as shown in FIG. 8A, even if the carrier number (k) is changed, A _m (k) (instantaneous value) is almost the same value. In addition, A _m (k) (instantaneous value) substantially matches R _rm e ^jαm (average value).

On the other hand, when the delay time is 300 [μs] (exceeding the guard interval), the influence of noise due to intersymbol interference occurs. Therefore, as shown in FIG. 8B, the instantaneous value of A _m (k) is distributed around the average value (R _rm e ^jαm ), and the level of R _rm e ^jαm (average value) depends on the signal condition. R ₁ = −6 [dB]. However, as shown in FIG. 8B, the phase number of the complex number detected from R _rm e ^jαm (average value) of A _m (k) is the initial position of the desired wave and the delayed wave, which are given signal conditions. It is in good agreement with the phase difference (45 degrees).

(Experiment 2: Constellation before and after equalization)
Experiment 2-1: When the delay wave is one wave <Experimental conditions>
When the delay wave is 1 wave and the delay time is 20 [μs], the constellation in the output before the equalization by the multipath distortion equalizer 1A and the constellation in the output after the equalization are measured. did. Further, the experiment was performed in the same manner by changing the delay time to 600 [μs] (a value exceeding the guard interval length and ± 168 [μs]).

<Experimental result>
FIG. 9A shows the experimental results before equalization and FIG. 9B shows the experimental results after equalization when the delay time is 20 [μs]. FIG. 9A and FIG. 9B show the twelfth symbol that is substantially in the center of the cut out 27 symbols. Similarly, in the case where the delay time is 600 [μs] (a value exceeding the guard interval length and ± 168 [μs]), the experimental results before and after equalization are respectively shown in FIGS. 10 (a) and 10 (b). Show.
As shown in FIGS. 9 and 10, it can be seen that multipath distortion is equalized (no symbol error) at any delay time.

Experiment 2-2: When there are two delayed waves <Experimental conditions>
A similar experiment was conducted when the delay wave was two waves.
However, the condition of the first delayed wave of the two waves is as follows.
R ₁ (delay wave level / desired wave level) = − 6 [dB]
Delay time (T _d1 ) = 150 [μs]
The condition of the second delayed wave of the two waves is as follows.
R ₂ (delayed wave level / desired wave level) = − 10 [dB]
Delay time (T _d1 ) = 900 [μs]

<Experimental result>
FIG. 11 (a) shows the experimental results before equalization, and FIG. 11 (b) shows the experimental results after equalization for the constellation when the delay wave is two waves. Although the delay time of both delay waves exceeds the guard interval length or ± 168 μs, it can be seen that, as shown in FIG. 11, the delay waves are equally equalized as in the case of one delay wave.

(Experiment 3: CNR after equalization at each symbol number)
In order to obtain good equalization performance, it is necessary not to cause a symbol error in the symbol after the determination process. For this purpose, the unnecessary component (modulation error) caused by the delayed wave is required to be within each frame on the phase plane (1/2 of the distance between signals). In 64QAM used for high-definition broadcasting, The desired wave (the wave having the highest level among the incoming waves) needs to have a level of 20 [dB] or higher with respect to these values.

Experiment 3-1: When the delay time is delayed <Experimental conditions>
When the delay wave is one wave, an experiment was performed to obtain the equalized CNR for the symbol number (n) using the delay time (T _d ) as a parameter.
At this time, the CNR (C / N) of the input signal is 60 [dB].
Signal conditions when the delay wave is one wave are as follows.
R ₁ (delay wave level / desired wave level) = − 3 [dB]

<Experimental result>
The experimental result of CNR after equalization for symbol number (n) in the case of delay is shown in FIG. The horizontal axis of the graph in FIG. 12 indicates symbol number (n), and the vertical axis indicates output C / N. Since the number of symbols is 27, the symbol number (n) is 0 to 26. The experiment was performed by changing the delay time (here, T _d ) to various values. Among these, the results at T _d = 20 [μs], 100 [μs], 600 [μs], 900 [μs], and 1.1 [ms] are shown in the graph.

In the method of the present invention, from the consideration of the above-described equations (33) and (34), the range in which the initial phase difference α _m can be detected is N _u when the delay time indicates “delay”. + T _g . That is, in actual time, T _u (effective symbol length) + T _g (guard interval length) is a theoretical limit. This sum is in the ISDB-T system, it is (9/8) T _u. At T _d = 1.1 [ms] close to this value, as shown in FIG. 12, a CNR of 30 [dB] or more is secured even after symbol number 11. In addition, when T _d = 1.1 [ms], the CNR is 21 [dB] (CNR at which the bit error rate of Viterbi decoding is 2 × 10 ^{−4 in} 64QAM modulation) or more in the seventh to tenth symbols. And reception is possible.

Experiment 3-2: When the delay time is advanced <Experimental conditions>
The signal conditions were the same as when the delay time was delayed.

<Experimental result>
FIG. 13 shows the experimental results of CNR after equalization for the symbol number (n) in the case of advance. The vertical axis and horizontal axis of the graph of FIG. 13 are the same as those of the graph of FIG. The experiment was performed by changing the delay time (here, T _d ) to various values. Among these, the results at T _d = −20 [μs], −100 [μs], −600 [μs], −900 [μs], and −1.1 [ms] are shown in the graph.

In scheme of the present invention, the equation (33) and from consideration of the formula (34), the initial phase difference alpha _m is detectable range is theoretical limit value when indicating the delay time "advances" in N _u Met. That is, in actual time, T _u (effective symbol length) is a theoretical limit. At -900 [μs], which is a delay time close to this value, as shown in FIG. 13, the 0th to 21st symbols can be used (CNR ≧ 21 [dB]). Note that this experiment is an experiment for a certain kind of variation phenomenon, and although not shown in the figure, it was found that the 0th to 21st symbols can be used in the same manner even at about −950 [μs]. .

  From the results of FIGS. 12 and 13, the yield may be lowered. However, as shown in FIG. 14, a plurality of equalizers (equalizer 1 and equalizer 2) are used, and 27 symbols out of each of the equalizers, except for the symbols positioned before and after, are intermediate. If the 15 symbols satisfying the receivable signal level located at are extracted and combined so as to be continuous, it is possible to generate a continuous signal from which symbols of low CNR are removed. In this case, the yield can be improved.

  Specifically, equalizer 1 and equalizer 2 remove the 0th to 6th symbols based on the result of FIG. 12, and also remove the 22nd to 26th symbols based on the result of FIG. Only the 7th to 21st symbols are output. Then, for example, after the 7th to 21st symbol groups output by the equalizer 1 (the part indicated by “a” in FIG. 14), the 7th to 21st symbol groups output from the equalizer 2 (see FIG. 14). 14), a continuous signal having a CNR of 21 [dB] or more (receivable) can be obtained for all symbols.

DESCRIPTION OF SYMBOLS 1,1A Multipath distortion equalization apparatus 2,2A transfer function production | generation means 3,3A equalization means 10 1st transfer function production | generation means 11 Cut-out part 12 FFT
13 SP Extraction Unit 14 SP Standard Value Storage Unit 15 Division Unit 16 Transfer Function Storage Unit 20, 20A Second Transfer Function Generation Unit 21 Complex Delay Wave Level Generation Unit 22 Delay Time Determination Unit 23 Subtraction Value Storage Unit 24 Rotor Storage Unit 25 Division unit 26 Average calculation unit 27 Deviation calculation unit 28 Comparison unit 30 Delay profile measurement means 40, 40A Transfer function calculation means 51 FFT
52 Division 53 IFFT
51A DFT
52A Division 53A IDFT
61 Initial Phase Difference Calculation Unit 62 Delay Time Polarity Determination Unit 70 Long-Period Cutout Unit 100, 100B Receiver 110 Frequency Conversion Unit 120 Synchronous Detection Unit 130 Receiver 140 Antenna 150 Receivable Signal Extraction Unit (Receivable Signal Extraction Unit)
160 Receivable signal combining unit (Receivable signal combining means)

Claims (5)

Transfer function generating means for generating a transfer function at a frequency to be equalized from a multipath wave indicating a received wave including a main wave and a delayed wave arriving as an OFDM radio wave, and a predetermined function from the input multipath wave By transforming the time domain data cut out by the period into a frequency domain signal by Fourier transform for each period, and by inverse Fourier transforming the result of dividing the transformed frequency domain signal by the transfer function, A multipath distortion equalization apparatus in OFDM signal reception comprising an equalization means for generating a signal in the time domain of the main wave,
The transfer function generating means includes
A transfer function at the frequency of the scattered pilot for the propagation path of the delayed wave is generated from the scattered pilot data extracted from the input multipath wave and the standard value of the scattered pilot stored in advance. First transfer function generating means;
Second transfer function generating means having delay profile measuring means,
The delay profile measurement means measures a delay profile by a power spectrum method for each period from time domain data cut out from the input multipath wave at a predetermined period, and determines the level of the delay wave and the main wave. Find the ratio and delay time
The second transfer function generating means includes
A high-frequency phase difference between the main wave and the delayed wave is obtained from the delay profile, and the polarity obtained from the measurement of the delay profile is different to a frequency and time domain function including a transfer function at the frequency of the scattered pilot. When the two delay times are respectively substituted, the delay time of the frequency average of the frequency and time domain functions that theoretically becomes zero and the one that does not become zero is defined as the true delay time. Identify,
Using the specified true delay time, the ratio between the level of the delayed wave and the level of the main wave, and the high-frequency phase difference, at the frequency to be equalized for the propagation path of the delayed wave A multipath distortion equalization apparatus characterized by generating a transfer function. The transfer function at the frequency of the scattered pilot and the transfer function at the frequency to be equalized include a term representing the main wave, a component of a ratio between the level of the delayed wave and the level of the main wave, A high-frequency phase difference component between the main wave and the delayed wave, and a term representing the delayed wave by the product of the time component indicating the delay time,
The second transfer function generating means includes
By dividing the term representing the delayed wave included in the transfer function at the frequency of the scattered pilot by the time component indicating the delay time for each delay time detected when measuring the delay profile, Complex delayed wave level generation means for generating a complex delayed wave level indicating a level drop caused by delay as a function of the frequency and time domain, respectively;
A statistic including at least an average of the generated complex delay wave levels with respect to the frequency to be equalized is calculated, and a complex delay wave level of which the frequency average of the calculated complex delay wave levels does not become zero is calculated. A delay time determining means for specifying the generated delay time as a true delay time;
Transfer function calculation for calculating a transfer function at the frequency to be equalized with respect to the propagation path of the delayed wave, using the specified true delay time and the complex delay wave level that generated the true delay time Means,
The multipath distortion equalization apparatus according to claim 1, comprising: The delay time determining means includes
Average calculating means for calculating an average of complex delay wave levels with respect to a change in frequency to be equalized for each complex delay wave level;
Deviation calculating means for calculating a deviation according to the delay time for generating the complex delayed wave level using the average of the complex delayed wave level calculated by the average calculating means,
Comparing means for comparing the magnitudes of the two deviations respectively obtained by the deviation calculating means, and specifying the delay time that generated the complex delay wave level with the smaller deviation as a true delay time;
The multipath distortion equalization apparatus according to claim 2, comprising: The transfer function at the frequency of the scattered pilot and the transfer function at the frequency to be equalized are the term representing the main wave, and the delay function with respect to the main wave for each of the plurality of delayed waves. A term representing each delayed wave by the product of the component of the ratio between the level of the delayed wave and the level of the main wave, the high-frequency phase difference component of the main wave and the delayed wave, and the time component indicating the delay time; Including
The second transfer function generating means includes
The frequency of each scattered pilot is obtained by subtracting the term representing the main wave from the transfer function at the frequency of the scattered pilot and then dividing by the time component indicating the absolute value of the delay time detected for the delayed wave. Each time an objective function is generated as a function of the frequency and time domain, the sum obtained by adding the objective function with respect to the frequency of each scattered pilot is obtained as a frequency average, and the high-frequency level of the delayed wave is obtained from the obtained sum. An initial phase difference calculating means for calculating an initial phase difference indicating a phase of the delayed wave with reference to a phase of a carrier center frequency of the main wave as a phase difference;
It is determined whether the value of the sum of the objective functions is 0. If the value does not become 0, it is determined that the polarity of the delay time is true. If the value is 0, the value is true. A delay time polarity determining means for determining that the polarity is not;
The component of the ratio between the level of the delayed wave and the level of the main wave detected when measuring the delay profile, the absolute value of the delay time of the delayed wave, the initial phase difference calculated for the delayed wave, and Transfer function calculation means for calculating a transfer function at the frequency to be equalized with respect to the propagation path of the delayed wave using the determination result of the delay time polarity determination means;
The multipath distortion equalization apparatus according to claim 1, comprising: A receiving apparatus comprising a plurality of multipath distortion equalization apparatuses according to any one of claims 1 to 4,
The period of extracting time domain data from the multipath wave input to each multipath distortion equalizer is an integer multiple of 3 or more of the effective symbol length, and the timing of extraction for each multipath distortion equalizer is different.
The receiving device is:
A signal in the time domain of the main wave generated by the equalization means of each multipath distortion equalization apparatus is acquired for each period, and is determined in advance from the main wave signal having a period that is an integral multiple of the effective symbol length. A predetermined number of effective symbol length portions satisfying the receivable signal level located in the middle except for the beginning and end portions of the period that do not satisfy the received receivable signal level, for each of the multipath distortion equalizers Receivable signal extracting means for extracting;
Receivable signal synthesizing means for recovering the main wave signal of the period by continuing a predetermined number of effective symbol length portions of the main wave signal extracted for each multipath distortion equalizer by the receivable signal extracting means. And a receiving device.

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