JP5298367B2 - Induction motor control device - Google Patents

Description

  The present invention relates to a control device for stably vector controlling an induction motor without a speed sensor.

  Vector control is widely known as a high-performance and high-accuracy control method for induction motors. Since vector control requires speed information of the motor, the speed information is usually obtained using a speed sensor such as a pulse generator. On the other hand, there are many requests to apply vector control to applications where a speed sensor cannot be installed, and various methods have been conventionally used for speed sensorless vector control including means for estimating speed and magnetic flux from applied voltage and current to an electric motor. Proposed.

  However, when the frequency of the voltage and current applied to the electric motor is zero, the voltage applied to the inductance of the electric motor is always zero, so that the magnetic flux cannot be estimated from the voltage. This indicates that the error between the estimated magnetic flux and the actual magnetic flux value does not converge. As a result, when the frequency is 0, speed estimation is impossible and the electric motor cannot be operated.

In view of such points, Patent Document 1 discloses a technique for operating a motor by enabling speed estimation even when the frequency of a voltage applied to the motor becomes zero.
FIG. 7 is a block diagram of the prior art described in Patent Document 1. In FIG. 7, 101 is a magnetizing current calculating means, 102 and 103 are torque current adjusting means and magnetizing current adjusting means, and 104 and 110 are coordinates. Conversion means, 105 is voltage detection means, 106 is current detection means, 107 and 108 are three-phase to two-phase conversion means, 109 is current / magnetic flux estimation means, 111 is speed estimation means, 200 is an inverter, M is an induction motor Show.

In this prior art, the current / magnetic flux estimation means 109 estimates the magnetic flux and current of the induction motor M from the voltage and current output from the inverter 200 using a model based on the state equation of the induction motor M.
Here, the state equation of the induction motor can be expressed as Equation 1. In Equation 1, T on the shoulder of the matrix represents a transposed matrix.

The current / magnetic flux estimation means 109 estimates the magnetic flux and current by the following mathematical formula 2. Here, a symbol “^” is added to the estimated value. In addition, the matrix A ^ is obtained by replacing the angular velocity ω _r in the matrix A to the estimated angular velocity ω _{r ^.}

The speed estimation unit 111, estimates of the current and the magnetic flux, the torque current command, the magnetic flux command, from the current detection value i _s etc. 2 Airyo, estimates the speed of the motor M using Equation 3 below.

Formula 3 is characterized by having an amplitude deviation between the magnetic flux command and the estimated magnetic flux, and a term calculated by the torque current command, and feedback control is performed so that the estimated magnetic flux amplitude does not diverge with respect to the magnetic flux command. For example, attention is paid to the fact that the estimated speed stably converges to 0 even at a frequency of 0.
By applying Formula 3, even if the frequency is 0, the magnetic flux vector of the motor M can be set to a desired value by correcting the estimated speed in a direction that eliminates the amplitude deviation between the magnetic flux command and the estimated magnetic flux according to the sign of the torque current. It converges to a state, and the estimated speed can be matched with the actual angular speed.

  In FIG. 7, the torque current command and the magnetizing current command are calculated from the estimated speed and the estimated magnetic flux obtained as described above. The current adjusting means 102 and 103 calculate biaxial voltage commands such that the torque current command and the torque current detection value, and the magnetizing current command and the magnetizing current detection value coincide with each other. The biaxial voltage command is converted into a three-phase voltage command by the coordinate conversion means 104 and is given to the inverter 200 as a primary voltage command.

On the other hand, as one of the operating conditions in which the output frequency is near 0 Hz, there is a case where the electric motor rotates at a low speed and is in a regenerative state.
Patent Document 2 discloses a technique for performing stable operation by speed sensorless vector control by avoiding an operating point where the output frequency is 0 Hz in such a case.

FIG. 8 is a block diagram of the prior art described in Patent Document 2, wherein 201 is a voltage-type PWM inverter, 300 is a vector controller, 301 is a command generator, 302 and 313 are adders, and 303 is speed control. , 304 is a magnetic flux command generator, 305 is a magnetic flux controller, 306 and 307 are torque current regulators and magnetizing current regulators, 308 is a voltage command calculator, 309 is a voltage detector, 310 is a current detector, and 311 is A speed estimator, 312 is a slip frequency command calculator, 314 is an integrator, 315 is a speed command control circuit, and 316 is a dq converter that converts a three-phase current into a biaxial quantity.
This prior art is characterized in that a speed command control circuit 315 is added to a general speed sensorless vector control device.

FIG. 9 is a block diagram of the speed command control circuit 315, in which 321 is a speed command increase discriminator, 322 is a switch, 323 is an integrator, 324 is a limiter, 325 is a coefficient unit, and 326 is a multiplier.
In this speed command control circuit 315, the speed command increase discriminator 321 determines whether or not the regenerative state is based on the product of the speed command value ω _r ^* and the torque current command value I _q ^*. The switch 322 is switched based on the magnitude of the next frequency command ω ₁ ^* . Specifically, when the primary frequency command ω ₁ ^* is not more than a predetermined value in the regenerative state, it is determined that the primary frequency is approaching 0 Hz, and the switch 322 is switched to the terminal b.
As a result, the integrator 323 calculates a positive speed command correction value, and the speed command control circuit 315 uses the speed command correction value to calculate the speed command value ω _r ^* from the command generator 301 in FIG. It operates as a speed command changing means for correcting the absolute value to a value larger than the predetermined value.

The effect of the above operation will be described using mathematical expressions. The relationship of Formula 4 is established among the primary frequency ω ₁ , the rotational speed ω _r , and the slip frequency ω _s of the electric motor.

In the regenerative state, ω _s is negative, and ω _s changes in proportion to the braking torque. Therefore, when the rotational speed is low, the primary frequency decreases as the braking torque increases, and finally reaches 0 Hz.
In the prior art described in Patent Document 2, when the primary frequency ω ₁ decreases as the braking torque increases, and the primary frequency ω ₁ becomes a predetermined value or less, the speed command control circuit 315 described above performs. Addition of the correction value to the speed command value is started to increase the rotation speed. This is equivalent to increasing the speed command value by the amount that the slip frequency has decreased due to the increase in braking torque.
Therefore, the sum of the rotation speed and the slip frequency is always constant, and the primary frequency command value is controlled to a predetermined constant value in view of the relationship of Equation 4. As a result, unstable driving in the vicinity of 0 Hz can be avoided.

JP 2000-224898 A (paragraphs [0010] to [0013], FIG. 1) JP 2004-40842 A (paragraphs [0023] to [0040], FIGS. 1 and 2)

In Patent Document 1, by correcting the estimated speed, the estimated speed can be made to match the actual angular speed even when the output frequency of the inverter becomes zero.
However, at the operating point where the output frequency of the inverter is near 0, the output voltage value is small because the operation is performed at a low speed, the output voltage detection error, the state equation used for the magnetic flux estimation value calculation and the actual motor model The effect of errors between the two becomes significant. Depending on the generation amount of these errors, a large speed estimation error may occur, resulting in unstable control, and normal operation becomes impossible. In the worst case, the motor is operated in a direction opposite to the rotational direction based on the rotational speed command. May be accelerated rapidly. In Patent Document 1, no mention is made of the influence of these errors.

Further, in Patent Document 2, the primary frequency decreases as the braking torque increases, and when the primary frequency falls below a predetermined value, the speed command value is increased if the braking torque further increases thereafter. Thus, the primary frequency is controlled to be constant. This is equivalent to controlling the primary frequency to be constant by increasing the speed command value by the amount that the slip frequency has decreased due to the increase in braking torque. Since the manipulated variable of the speed command value is calculated by an integrator, the speed of increase of the speed command value is determined by the integration time constant. On the other hand, the slip frequency varies depending on the load state.
Here, considering that the braking torque has a time constant sufficiently faster than the integration time constant of the integrator and the load fluctuates so as to increase rapidly, the slip frequency decreases rapidly, but the operation of the speed command value The amount cannot follow the change of the slip frequency. As a result, the decrease amount of the slip frequency becomes larger than the increase amount of the rotation speed.
Considering the relationship of Equation 4, this indicates that the primary frequency cannot be controlled to a constant value and the primary frequency becomes transiently below a predetermined value. In the worst case, the primary frequency is 0 Hz. There is also a risk that the control may become unstable by reaching the vicinity.

  Accordingly, the problem to be solved by the present invention is to solve the above-mentioned problems, and in any case, an induction motor capable of stably realizing speed sensorless vector control without rotating the motor in the direction opposite to the rotation speed command. It is to provide a control device.

In order to solve the above-mentioned problem, the invention according to claim 1 estimates and calculates the magnetic flux, current and speed of the motor using the current and voltage of the induction motor, and uses the estimated speed to convert power by vector control. In a control device for an induction motor that performs variable speed control of the electric motor via a heater,
Means for adding a speed command correction amount to the speed command value of the motor;
Means for validating the speed command correction amount when the primary frequency decreases as the braking torque of the motor increases and the absolute value thereof reaches the absolute value of the lower limit value of the primary frequency command greater than zero. ,
The speed command correction amount, subtracting the absolute value of the speed command value from the absolute value of the lower limit value of the primary frequency command, and you characterized in that it is a value obtained by adding the absolute value of the slip frequency command value .

The invention according to claim 2 is the control apparatus for the induction motor according to claim 1,
Wherein an addition signal and a speed command value and the slip frequency provided with means for comparing the lower limit value as a reference signal, characterized that you and enable or disable the speed command correction amount according to a comparison result by the means.

  According to the present invention, in speed sensorless vector control of an induction motor, stable operation is possible without rotating in the direction opposite to the direction of the rotation command even in the extremely low speed region of the motor.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing an embodiment of the present invention. Although an embodiment of the induction motor at the time of forward rotation will be described here, the present invention can be applied by the same principle at the time of reverse rotation.

  In FIG. 1, components having the same functions as those shown in FIG. In FIG. 1, 114 is a primary frequency lower limit limit execution determination means, 115 is a speed command correction amount calculation means to which a limit execution flag is input from the determination means 114, and 116 is a speed command correction from the calculation means 115. A torque current command calculating means for inputting the quantity together with the speed command value and the speed estimated value.

In the present embodiment, the primary frequency lower limit limit execution determination unit 114 determines whether or not to perform an operation on the primary frequency and speed command from information used to calculate a primary voltage command to be given to the inverter 200. Then, the limit execution flag is enabled according to the determination result, and is output to the primary frequency calculation means 113 and the speed command correction amount calculation means 115.
In this embodiment, the motor M can be stably driven by speed sensorless vector control even in an extremely low speed range by enabling the limit execution flag and operating the primary frequency and speed command when a specific condition is satisfied. It is a thing. Since the basic configuration of the control device is the same as that of the prior art and general vector control shown in FIGS. 7 and 8, the following description will focus on the differences.

FIG. 2 shows the configuration of the primary frequency calculation means 113 in FIG.
The primary frequency calculation means 113 first adds the estimated speed value and the slip frequency. The above part is the same as the prior art, but in the present invention, the speed estimation value and the slip frequency are added by the adding means 113a, and the lower limit value of the added signal is limited by the lower limit limit means 113b when the limit execution flag is valid. Output as a primary frequency command.

Also, the lower limit means 113b lowers the above limit signal with the lower limit value only when the limit execution flag output from the primary frequency lower limit limit execution determination means 114 in FIG. 1 is valid. Process. As a result, the value of the primary frequency command is maintained at the minimum value of the lower limit means 113b at the minimum, so that the operation where the primary frequency is in the vicinity of 0 Hz can be avoided.
When the limit execution flag is invalid, the addition signal output from the adding means 113a is output as it is as a primary frequency command, which is exactly the same as the adder 313 in the prior art of FIG. To work.

FIG. 3 shows the configuration of the speed command correction amount calculation means 115 in FIG.
In the speed command correction amount calculation means 115, the primary frequency lower limit value, the speed command value, and the slip frequency are added and subtracted by the subtraction means 115a and 115b based on Equation 6 described later, and the result is given to the correction amount selection means 115c. The correction amount selection unit 115c selects a correction amount according to the limit execution flag. That is, when the limit execution flag is invalid, the speed command correction amount is set to 0, and only when the limit execution flag is valid, the calculation result according to Equation 6 is output as the speed command correction amount. That is, the calculation result of the speed command correction amount according to Equation 6 is validated or invalidated according to the limit execution flag.

FIG. 4 shows the configuration of the torque current command calculation means 116 in FIG.
In the torque current command calculating means 116, first, the speed command correction amount calculated by the speed command correction amount calculating means 115 is added to the speed command value set in the inverter 200 by the adding means 116a. Then, the output signal of the adding means 116a is used as the true speed command value, the difference from the estimated speed value is calculated by the subtracting means 116b, and the result of the proportional integration calculation by the PI control means 116c is output as the torque current command value.
Note that when the limit execution flag is invalid, the speed command correction amount is 0, so the same operation as the speed controller 303 in FIG. 8 is performed. It should be noted that hereinafter, a speed command value simply represents a speed command value to which no speed correction amount is added.

Here, the principle and effect of the present invention will be described with reference to FIG. FIG. 5 shows the operation of this embodiment when the braking torque is gradually increased while the induction motor M is operated at a constant speed.
First, in the speed sensorless vector control, the primary frequency command ω ₁ to the inverter is calculated by Equation 5.

In Equation 5, ω _r ^ is a speed estimation value and ω _s is a slip frequency. Since the slip frequency ω _s is proportional to the output torque, the primary frequency decreases as the braking torque increases. In particular, when the speed command value is low, depending on the value of the braking torque, the primary frequency reaches 0 Hz, the control becomes unstable, and the motor cannot be operated.

Therefore, in the present embodiment, the primary frequency decreases as the braking torque increases, and approaches a lower limit value (hereinafter also simply referred to as a primary frequency lower limit value or a lower limit value) _ωth of a preset primary frequency command. If it is discriminated, the limit execution flag is switched to valid as shown in FIG. The limit execution flag is enabled, in order to perform the lower limit processing to the primary frequency which are calculated as described above, thereafter, the primary frequency command braking torque be increased in certain remains lower limit omega _th Be controlled. In this embodiment, since the lower limit value is provided for the primary frequency command, the primary frequency is not transiently lower than the lower limit value as in Patent Document 2, and the operation in which the primary frequency is in the vicinity of 0 Hz is completely performed. Can be avoided.

  On the other hand, in vector control, Formula 5 should always be satisfied, but since the lower limit process forcibly changes the primary frequency, the relationship of Formula 5 cannot be established simply by performing the lower limit process. In particular, as in the conventional technique described in Patent Document 1, in the case of a control method for estimating magnetic flux or the like by calculation based on the voltage-current equation of the motor, providing a lower limit value for the primary frequency is Since this is not taken into account, it is impossible to perform a correct estimation simply by performing the lower limit process, and an unexpected operation may occur.

Therefore, in the present embodiment, as shown in FIG. 5, the change in the primary frequency command due to the lower limit processing is added to the speed command value as a speed command correction amount ω _cmp . The speed command correction amount ω _cmp is calculated by, for example, Equation 6, but can be calculated using other control variables, and is not limited to Equation 6.

In Equation 6, ω _r ^* is a speed command value before the correction amount is added.
As a result, since the speed estimated value follows the speed command value obtained by adding the correction amount ω _cmp , the lower limit process is performed on the primary frequency, and even when the primary frequency is fixed to the lower limit value, Always holds. Therefore, the magnetic flux can be estimated stably even during the limiter operation.
Note that Formula 6 is an arithmetic expression of the speed command correction amount ω _cmp corresponding only to the forward rotation (speed command value> 0). In order to support the reverse rotation (speed command value <0), Formula 7 It is sufficient to calculate by

By performing vector control using the value obtained by adding the speed command correction amount ω _cmp calculated by Expression 7 to the absolute value of the speed command value as the true speed command absolute value, the present invention can be applied regardless of whether the rotation is normal or reverse. An effect can be obtained.
However, in this embodiment, since the correction amount is added to the speed command value, the actual rotation speed of the motor follows the speed command with the correction amount added and rotates at a rotation speed different from the original speed command. become. However, since it is possible to avoid an operation in which the primary frequency that becomes unstable in principle in sensorless control is near 0 Hz, the control becomes unstable and the motor is not rotated in the direction opposite to the command. Also, the higher the output voltage primary frequency becomes lower decreases, the influence of the error of the error and the detection system of the motor constant is large, in the present embodiment, since maintaining such primary frequency is not lower than the lower limit value omega _th Thus, the influence of these errors can be reduced. As described above, according to the present embodiment, it is possible to stably control the electric motor even in an extremely low speed region.

  In speed sensorless control, there are various methods for estimating magnetic flux and speed. However, in this embodiment, since only the primary frequency command and the speed command are corrected, the present invention is applicable to speed sensorless control using any magnetic flux and speed estimation method. Further, the present embodiment can be realized only by modifying a program such as a CPU that calculates a voltage command to the inverter 200, and the realization thereof is very easy.

  Next, FIG. 6 shows a specific configuration of the primary frequency lower limit limit execution determination unit 114 that determines whether or not to operate the lower limiter for the primary frequency command. Here, the embodiment at the time of forward rotation of the electric motor is shown, but it can be applied by the same principle at the time of reverse rotation.

The primary frequency lower limit limit execution determination means 114 first calculates a reference signal by adding the speed command value and the slip frequency by the addition means 114a. In comparison means 114b, and comparing the reference signal and the primary frequency lower limit omega _th, the reference signal is to enable the limit execution flag only if the lower limit value omega _th.
In vector control, control is performed so that the speed command value matches the speed estimation value, so that the speed command value and the speed estimation value can be regarded as matching. Therefore, the reference signal is a primary frequency command (corresponding to a primary frequency command in the prior art) when the limiter of the primary frequency does not operate at all. If this reference signal is used, it can be determined whether or not the actual primary frequency is approaching the lower limit value. Therefore, it can be said that it is suitable for creating a limit execution flag.

1 is a block diagram showing a first embodiment of the present invention. It is a block diagram of the primary frequency calculating means in FIG. It is a block diagram of the speed command correction amount calculating means in FIG. It is a block diagram of the torque current command calculating means in FIG. It is a figure for demonstrating the principle and effect of this invention. It is a block diagram of the primary frequency lower limit limit execution determination means in FIG. It is a block diagram of the prior art described in patent document 1. It is a block diagram of the prior art described in patent document 2. It is a block diagram of the speed command control circuit in FIG.

Explanation of symbols

101: Magnetizing current command calculating means 102: Torque current adjusting means 103: Magnetizing current adjusting means 104, 110: Coordinate converting means 105: Voltage detecting means 106: Current detecting means 107, 108: Three-phase to two-phase converting means 109: Current Magnetic flux estimation means 111: Speed estimation means 112: Slip frequency calculation means 113: Primary frequency calculation means 113a: Addition means 113b: Lower limit means 114: Primary frequency lower limit limit execution determination means 114a: Addition means 114b: Comparison means 115 : Speed command correction amount calculation means 115a, 115b: Subtraction means 115c: Correction amount selection means 116: Torque current command calculation means 116a: Addition means 116b: Subtraction means 116c: PI control means 200: Inverter M: Induction motor

Claims (2)

Induction motor control that estimates and calculates the magnetic flux, current, and speed of the motor using the current and voltage of the induction motor, and uses the estimated speed value to control the speed of the motor through a power converter by vector control. In the device
Means for adding a speed command correction amount to the speed command value of the motor;
Means for validating the speed command correction amount when the primary frequency decreases as the braking torque of the motor increases and the absolute value thereof reaches the absolute value of the lower limit value of the primary frequency command greater than 0;
Equipped with a,
The speed command correction amount is a value obtained by subtracting the absolute value of the speed command value from the absolute value of the lower limit value of the primary frequency command and adding the absolute value of the slip frequency command value. Electric motor control device. In the induction motor control device according to claim 1,
Inducing said addition signal and a speed command value and the slip frequency provided with means for comparing the lower limit value as a reference signal, characterized that you and enable or disable the speed command correction amount according to a comparison result by the means Electric motor control device.

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