JP5236057B2 - AC / DC converter, control method for AC / DC converter, heat pump type water heater and air conditioner - Google Patents

Description

  The present invention relates to an AC / DC converter that suppresses harmonic current and controls a DC voltage, a control method for the AC / DC converter, a heat pump water heater and an air conditioner using the AC / DC converter.

  As a conventional AC / DC converter, the power supply is short-circuited through the reactor only once every half cycle while being synchronized with the power supply using a zero-cross signal of the power supply voltage, etc. It is known that a charging current is caused to flow by energy to suppress a harmonic current and improve a power factor (see, for example, Patent Document 1).

  Further, the reactor becomes large when the power supply is short-circuited only once in a half cycle of the power supply. In view of this, there has been known one that reduces the size of a reactor by performing a short-circuit operation twice or more in a power supply half cycle (for example, see Patent Document 2).

  Furthermore, a switch means for switching between full-wave rectification and voltage doubler rectification, a switch means for performing a power supply short circuit, and two switch means are known to suppress harmonic current and improve power factor. (For example, see Patent Documents 3 and 4).

  Furthermore, there are known ones that reduce the capacitor capacity by alternately switching two switch means, and those that improve the power factor by shifting the on-timing of switching and switching for a predetermined on-time (for example, patents) References 5 and 6).

  Further, it is known that the switch means is operated by high-frequency PWM to control the input current in a substantially sine wave shape to suppress harmonics and improve the power factor. (For example, refer to Patent Document 7).

  Furthermore, what is trying to suppress a harmonic current by operating two switching elements is known (for example, refer patent document 8, nonpatent literature 1).

Japanese Patent No. 2763479 (pages 9 to 10, FIGS. 1 to 7) Japanese Patent No. 3485047 (pages 3 to 6, FIGS. 1 to 7) Japanese Unexamined Patent Publication No. 2003-9535 (pages 4 to 7, FIGS. 1 to 13) Japanese Patent No. 3687641 (pages 7-9, FIGS. 1-5) Japanese Patent Laying-Open No. 2005-110491 (pages 17 to 25, FIGS. 1 to 4) JP 2008-99512 A (Pages 5 to 10, FIGS. 1 to 15) Japanese Patent No. 2140103 (second page, FIG. 4) JP 2008-22625 A (Pages 4 to 5, FIGS. 1 to 4)

Shinichi Hoshi, Kuniomi Oguchi, “Switching pattern determination method for single-phase multi-level rectifier circuit”, H17 Annual Conference of the Institute of Electrical Engineers of Japan, No. 1-61 (2nd page, FIG. 3)

  The conventional technique shown in Patent Document 1 is very simple control, and the operation of the switch means in the half cycle of the power supply is switching at a relatively low frequency at 100 Hz or 120 Hz, generating less noise, and inexpensive. Widely used as a method that can realize harmonic current suppression.

  On the other hand, a limit value is determined for the harmonic current included in the input current flowing from the power source, and it is necessary to suppress it to the limit value or less. However, when the harmonic current is to be suppressed below the limit value by the technique described in Patent Document 1, the impedance needs to be larger than a predetermined value. However, since the switching frequency is relatively low, the inductance value is increased. Therefore, there is a problem that the size of the reactor is increased.

  Therefore, Patent Document 2 discloses a technique for downsizing the reactor without changing the harmonic suppression performance by increasing the number of short-circuit operations of the switch means, but the power consumption increases and the input current increases. However, there is a problem that the reactor becomes large even if the inductance value is the same.

  Therefore, according to the high frequency PWM shown in Patent Document 7 (in this document, there is no particular description about the frequency, but in general, a technique for operating the switching means at a switching frequency of 15 to 20 kHz or more) The current becomes a substantially sine wave, and the harmonic current decreases drastically. In addition, it is theoretically possible to boost the output DC voltage higher than the reference voltage (DC voltage when the switch means is off) by adjusting the PWM duty factor. The pressure can be increased until

  However, this high-frequency PWM is current control that detects an input current and converts the input current into a substantially sine wave, and therefore requires high-speed control processing and high-frequency PWM control. Therefore, there are many generated noises, and the cost for noise countermeasures is enormous. In addition, since high-speed control is required, it is necessary to use a complicated peripheral circuit for analog control by a microcomputer having high processing performance or a dedicated IC, and there is a problem that it is expensive.

  Moreover, in the prior art shown by patent document 3 and patent document 4, the variable range of direct-current voltage becomes wide by providing the switch means which switches full wave rectification and voltage doubler rectification, and the switch means which performs a power supply short circuit. However, since the switching is relatively low frequency, the problem of increasing the size of the reactor cannot be solved.

  Furthermore, in the prior art shown in Patent Document 5, it is possible to reduce the capacitance of the capacitor by performing complementary switching at a frequency higher than the power supply frequency (for example, 1/1000 second in the literature) This is switching for reducing the capacity of the capacitor, and complementary switching has a problem in reducing the power supply harmonic current.

  In addition, Patent Document 6 shows a technique in which two switching elements are used with respect to the technique shown in Patent Documents 1 and 2, and the magnetic sound generated from the reactor is suppressed by a combination of the ON times. Even with this technology, there was a limit to miniaturization of the reactor.

  Further, Patent Document 8 discloses a technique for performing high-frequency PWM shown in Patent Document 7 using two switching elements. However, this is also current control in which current is detected and taken into the control. High frequency PWM control is required, and the problem of high cost cannot be solved.

  Non-Patent Document 1 discloses a technique for suppressing the harmonics of the input current by increasing the level of the input voltage of the rectifier using two switch means. According to this technique, the reactor can be operated even at relatively low frequency switching. Can be miniaturized. This technology calculates the on / off timing of switch means by GA (genetic algorithm) on the assumption that operating conditions change, such as controlling DC voltage or changing power consumption. Is. However, GA has a problem in mounting it in a control CPU such as a microcomputer in that a parameter is not determined unless it is repeated complicated calculation and generation change, and the previously calculated parameter is stored in a memory or the like. Therefore, it takes a long time to develop a product with a large number of models, and it has a large amount of memory, which causes practical difficulties.

  The present invention has been made to solve the above-described problems, and a first object of the present invention is to suppress the harmonic current at a lower cost than the high-frequency PWM and to improve the power factor. And a method for controlling the AC / DC converter, and a heat pump type water heater and an air conditioner using the AC / DC converter.

  A second object is to obtain an AC / DC converter, a control method for the AC / DC converter, and a heat pump water heater and an air conditioner using the AC / DC converter that can reduce the size of the reactor. .

  The third object is to obtain an AC / DC converter, a control method for the AC / DC converter, and a heat pump water heater and an air conditioner using the AC / DC converter that can be applied to a plurality of models having different operating conditions. There is.

An AC / DC converter according to the present invention includes a first rectifier and a second rectifier connected to an AC power source via a reactor, two capacitors connected in series between output terminals of the first rectifier, A first switch and a second switch connected in series between output terminals of the second rectifier, a connection point between the capacitors, a connection point of the first switch and the second switch, and a first switch Control means for PWM-controlling a switch group consisting of the first switch and the second switch so that the voltage between the output terminals of the rectifier becomes a predetermined value, and the control means is a combination of ON / OFF of the switch group. Each time ratio and generation order are controlled to generate a three-level substantially sinusoidal voltage between the input terminals of the rectifier .

  According to the AC / DC converter of the present invention, the control means performs PWM control on the switch group including the first switch and the second switch so that the voltage between the output terminals of the first rectifier becomes a predetermined value. By controlling the voltage between the output terminals of the first rectifier, the current flowing through the reactor can be converted into a sine wave. As a result, the reactor can be made smaller than the conventional method in which the switch means is operated once or several times in the half cycle of the power supply.

  Further, it is possible to operate with a PWM signal with a low frequency of about 1 kHz to 5 kHz, and there is no cost increase due to noise countermeasures with the high frequency PWM, and it can be put into practical use at a low cost.

1 is a circuit block diagram showing a first embodiment of the present invention. It is a circuit block diagram in the ideal state explaining Embodiment 1 of this invention. It is a principle operation | movement waveform diagram explaining Embodiment 1 of this invention. It is a circuit block diagram which shows the operation mode explaining Embodiment 1 of this invention. It is a vector diagram of the alternating voltage and electric current explaining Embodiment 1 of this invention. It is a feedback control block diagram explaining Embodiment 1 of this invention. It is a modulation waveform diagram of the operation signal for explaining the first embodiment of the present invention. It is a PWM waveform diagram of the operation signal explaining Embodiment 1 of the present invention. It is a modulation waveform diagram of PWM of the operation signal for explaining the first embodiment of the present invention. It is a PWM carrier waveform figure explaining Embodiment 2 of this invention. It is a structural diagram of the reactor explaining Embodiment 3 of this invention. It is another circuit block diagram explaining Embodiment 4 of this invention. It is a block diagram of the other bidirectional | two-way switch means in embodiment of this invention. It is a circuit block diagram which shows Embodiment 5 of this invention.

Embodiment 1 FIG.
FIG. 1 is a circuit block diagram showing Embodiment 1 of the present invention. In FIG. 1, an AC / DC power supply apparatus includes a rectifier 2 for rectifying an alternating current of an AC power supply 1, a first switch means 3 connected to one input terminal of the rectifier 2, and the other input terminal of the rectifier 2. Are connected to the second switch means 4 to which one is connected, the reactor 5 inserted between the AC power source 1 and the first switch means 3 or the second switch means 4, and one to the output terminal of the rectifier 2. The first capacitor 6 is connected, the second capacitor 7 is connected to the other output terminal of the rectifier 2, and the DC load 8 is connected to the output of the rectifier 2. The first switch means 3, the second switch means 4, the first capacitor 6, the second capacitor 7, and the other end of each of them are connected together. A diode 10 and a resistor 12 are connected in parallel with the first capacitor 6, and a diode 11 and a resistor 13 are connected in parallel with the second capacitor 7, and the diode 10 and the diode 11 are connected to the first capacitor 6 and the second capacitor. 7 and the polarity is reversed. The first capacitor 6 may be configured by connecting a plurality of capacitors in parallel or in series. The same applies to the second capacitor 7. The first switch means 3 may be configured by connecting a plurality of switches in parallel or in series. The same applies to the second switch means 4.

Further, the first switch means is a bidirectional switch means composed of, for example, an IGBT 3a and a diode rectifier 3b, and the second switch means is also a bidirectional switch means composed of, for example, an IGBT 4a and a diode rectifier 4b. Control means 20 for operating these switch means based on the output of the voltage detector 21 to detect and the output of the zero cross detector 22 for detecting the phase angle of the AC power supply 1 is provided.
The AC / DC converter of the present invention is realized by the two switch means 3 and 4 operating in a mutual manner as represented by a virtual AC power supply as shown in FIG.

  The AC / DC converter of the present invention will be described with reference to FIG. The AC power source 1 and the reactor 5 are the same as those shown in FIG. 1, and the AC / DC converter is a virtual AC power source 9. The voltage across the AC power supply 1 is Vs, the voltage across the virtual AC power supply 9 is Vc, and the current flowing through the reactor 5 is I as in FIG.

The current I flowing through the reactor 5 is determined by the voltage difference between the AC power supply 1 and the virtual AC power supply 9. Since the reactor current I is an AC amount, if the voltage across the reactor 5 is set to jωLI, the voltage difference between the AC power source 1 and the virtual AC power source 9 is expressed by the following equation (1).
jωLI = Vs−Vc (1)
Here, ω is an angular frequency, L is the inductance of the reactor 5, and j is an imaginary number.

When the voltage effective value of the AC power supply 1 is set to V1, the voltage Vs of the AC power supply 1 is expressed by the following formula (2).
Vs = √2 · V1 · sin (ωt) (2)
When the voltage effective value of the virtual AC power supply 9 is set to V2, the voltage Vc of the virtual AC power supply 9 is represented by the following formula (3).
Vc = √2 · V2 · sin (ωt−φ) (3)
Where φ is the phase difference between Vs and Vc.
Here, assuming that V1 = V2, the current I flowing through the reactor 5 can be expressed by the following formula (4) by calculation from formulas (1) to (3).
I = √2 · V1 / (jωL) · 2 · sin (φ / 2) · cos (ωt−φ / 2)
.... (4)
If the phase difference between Vs and Vc does not fluctuate, sin (φ / 2) becomes a constant. Therefore, if the constant is set to K all together, the current I is expressed by the following equation (5).
I = −j · K · cos (ωt−φ / 2) (5)
Here, since cos (ωt−φ / 2) can be expressed as sin (ωt−φ / 2 + π / 2), it can be easily understood that the waveform of the current I is a sine wave.

  Thus, if the voltage Vc output from the virtual AC power supply 9 is output in a sine wave shape, the current flowing in the reactor 5, in other words, the input current is converted into a sine wave having the same frequency as the frequency of the AC power supply 1. Current flows and harmonic current is suppressed. Further, when the phase difference between this current and the AC power source 1 becomes zero, the power source power factor becomes 100%. Therefore, the phase difference φ between the amplitude V2 in the virtual AC power source 9 and the AC power source 1 is appropriately controlled. If a sine wave voltage is output, the harmonics of the input current can be suppressed and power factor improvement can be realized.

  Therefore, the first switch means 3 and the second switch means 4 for outputting the three-level substantially sinusoidal voltage as shown in FIG. 3 as the converter voltage Vc between the input terminals of the rectifier 2 in FIG. 1 are operated. Will be described. In FIG. 3, Vo is an output DC voltage applied to the load 8.

  The operation of the first switch means 3 and the second switch means 4 for outputting the substantially sinusoidal voltage shown in FIG. 3 between the input terminals of the rectifier 2 will be described with reference to FIG. For the two switches of the first switch means 3 and the second switch means 4, there are 22 = 4 combinations of on and off. When the two switch means 3 and 4 are simultaneously turned on, the input terminals of the rectifier 2 are short-circuited. The circuit operation at this time is shown in FIG. In FIG. 4, the first switch means 3 and the second switch means 4 are described as ideal switches Sa and Sb, respectively. When the first switch means 3 and the second switch means 4 are turned on simultaneously, the input terminals of the rectifier 2 are short-circuited, so the converter voltage Vc becomes Vc = 0. It corresponds to.

  When the first switch means 3 is on and the second switch means 4 is off, the output voltage between the input terminals of the rectifier 2 is equal to the voltage across the second capacitor 7 as shown in FIG. 1/2 of the DC voltage Vo is output, and the converter voltage Vc is Vc = Vo / 2, which corresponds to the area A in FIG.

  Conversely, when the first switch means 3 is off and the second switch means is on, the voltage between the input terminals of the rectifier 2 is equal to the voltage across the capacitor 6 as shown in FIG. Therefore, it is ½ of the output DC voltage Vo. In this case, the converter voltage Vc is Vc = Vo / 2. Therefore, it corresponds to the area a in FIG.

  Next, when the first switch means 3 is off and the second switch means 4 is off, as shown in FIG. 4 (d), a full-wave rectification state occurs, so the voltage between the input terminals of the rectifier 2 is The voltage across the capacitor 6 and the capacitor 7 is equal to Vo, and Vc = Vo. This is the area C in FIG.

  When the control means 20 appropriately controls the time ratio and order of occurrence of the areas a to c in FIG. 3, it is possible to output a three-level sine wave voltage as the converter voltage Vc.

(E) to (h) in FIG. 4 are also the same operations as described above, except that the polarity of the AC power supply 1 is different. In FIG. 4, the direction of the arrow Vc in FIG. Since it is in the same direction as d), it is a negative value in FIG.
Accordingly, the region in FIG. 3 can also generate a region a ′ having a reverse polarity of Vc = −Vo / 2 and a region c ′ in which Vc = −Vo.

  Further, in the states of (b), (c), (f), and (g) in FIG. 4, the connection point of the first capacitor 6 and the second capacitor 7 and one end of the AC power supply 1 are connected. The circuit has the same configuration as so-called voltage doubler rectification. By appropriately controlling the appearance rate of the state where only one of the two switch means is turned on, in other words, the ratio at which Vo / 2 is output as the converter voltage Vc, the value of the output DC voltage Vo is set. This means that it can be controlled to a value that is greater than or equal to the DC voltage obtained by full-wave rectification.

  Next, a method for determining the on / off timing of the two switch means will be described. In the present invention, the on / off timing is not searched by calculation in advance, but the on / off timing of the two switch means is determined by feedback control.

  The feedback control of the present invention will be described with reference to FIG. FIG. 5A is a vector diagram showing a vector relationship derived from the principle circuit configuration in FIG. 2, and is generally described in textbooks. The current I is delayed in the reactor 5 with respect to the voltage Vs of the AC power supply 1. A voltage drop jωLI occurs in the reactor 5 so as to be orthogonal to the current I, and coincides with the voltage Vs of the AC power supply 1 by vector addition with Vc that is a converter voltage between the input terminals of the rectifier 2.

Therefore, in order to output the converter voltage Vc so that the power factor becomes 1, the delay phase of the current I may be 0. That is, the state shown in FIG. 5B may be a state in which the triangle in FIG. 5 is a right triangle in which Vs and jωLI are orthogonal to each other. Therefore, the phase difference φ may be controlled so that the delay phase difference φ of the converter voltage Vc with respect to the voltage Vs of the AC power supply 1 becomes the following formula (6).
φ = tan-1 (ωLI / V1) (6)

Further, the converter voltage Vc may be output so that the amplitude V2 of the converter voltage Vc becomes the following expression (7).

  If a feedback control system is constructed so that the phase difference φ and the amplitude V2 between the converter voltage Vc and the voltage Vs of the AC power supply 1 are uniquely determined, known sawtooth wave modulation, triangular wave modulation, space vector modulation, dipolar modulation, etc. By applying the modulation method, an operation signal for operating the switch means 3 and 4 can be generated.

Based on the above operating principle, the control means 20 operates as follows.
First, the control means 20 derives the phase difference φ between the voltage Vs of the AC power source 1 and the converter voltage Vc by DC voltage feedback control in order to control the output DC voltage of the AC / DC converter.

  FIG. 6 shows an example of a control block for controlling the phase difference φ. The DC voltage command value and the DC voltage detection value (here, Vo) are compared, and the difference is input to the PI controller. The output of the PI controller is configured to be a current command, and a current is obtained as an output value from the PI controller. Although this PI controller is realized by software, it goes without saying that it may be realized by hardware. The control means 20 obtains the phase angle φ by inputting the current command I *, which is the output of the PI controller, into the current I in the equation (6) of the phase difference φ. Further, the control means 20 calculates the amplitude V2 of the converter voltage Vc simultaneously with the expression (7) (that is, the right triangle Pythagorean theorem in FIG. 5B) when the phase difference φ is derived.

  Next, the control means 20 generates the converter voltage Vc according to the equation (3). Distribution from the converter voltage Vc to the first switch means 3 and the second switch means 4 can be realized by general unipolar modulation. A waveform diagram of the unipolar modulation is shown in FIG. The sine wave waveforms shown in FIGS. 7A and 7B are the converter voltage Vc generated by the equation (3). FIG. 7A shows a modulation signal for the first switch means 3 and FIG. 7B shows a modulation signal for the second switch means 4.

  The waveform in FIG. 7A will be described. The control means 20 compares the triangular wave (carrier wave) inverted by the positive polarity and the negative polarity with the converter voltage Vc generated by the equation (3). If the absolute value on the negative electrode side is taken, it corresponds to the positive electrode side, so unipolar modulation is performed. By turning off when the converter voltage Vc is larger than the triangular wave as the carrier wave, the operation signal of the first switch means 3, the waveform of FIG. 7C (Hi side is on) is obtained.

  Similarly, although the waveform of FIG. 7B is the second switch means 4 is on the negative side with respect to the converter voltage Vc, the modulation waveform is a sine whose phase is inverted by 180 degrees with respect to FIG. It becomes -Vc of the wave. Further, the triangular wave as a carrier wave also has a phase inverted by 180 degrees with respect to FIG. The control means 20 compares the modulated wave and the carrier wave in the same manner as described above, and the operation signal of the second switch means 4 and the waveform of FIG.

  Next, the control means 20 operates the first switch means 3 and the second switch means 4 with the waveforms of (c) and (d) of FIG. Thereby, the three-level converter voltage Vc is obtained by adding the waveforms of (c) and (d) of FIG. However, since (c) and (d) in FIG. 7 are Hi and the switch means is ON, for easy explanation, when Hi is set to 0 and Lo is set to 1, as shown in FIG. A chopped converter voltage Vc is obtained. The control means 20 outputs the three-level converter voltage Vc obtained by this unipolar modulation to the first switch means 3 and the second switch means 4 as a PWM signal.

  Here, (b) and (c) in FIG. 4 are the same circuit form (so-called voltage doubler rectification configuration) in which the AC power supply 1 has the same polarity and outputs Vo / 2, but different Vos in the same polarity. It is necessary to provide a circuit configuration for outputting / 2. Two capacitors are provided in series to output ½ of the output voltage Vo. However, when Vo / 2 is output, voltage rectification is performed, so the first capacitor 6 or the second capacitor 7 is output. Any of these will be charged. If only one capacitor is charged, it does not become half of the output voltage across the capacitor, and the converter voltage Vc is distorted, whereby the input current is also distorted and the harmonic current cannot be suppressed.

  Therefore, the first switch 6 and the second capacitor 7 are charged in the same polarity of the AC power supply 1 and the first switch means 3 and the second switch 7 are maintained so that a balance of 1/2 of the output voltage Vo is maintained. It is necessary to operate the switch means 4 in a balanced manner.

  In the unipolar modulation of FIG. 7, only the first switch means 3 is on, only the second switch means 4 is on, and these two operation modes Vc = Vo / 2 occur alternately. It can be said that this is a modulation system that is very suitable for the circuit configuration of the present invention because it operates in a balanced manner.

  In the first embodiment of the present invention, the unipolar modulation is described. However, even if the unipolar modulation is not used, if it is possible to distribute the converter voltage Vc in a balanced manner so that the converter voltage Vc is output by two switch means, for example, bipolar modulation or It goes without saying that any modulation method such as dipolar modulation or sawtooth wave modulation has the same effect.

  As described above, the first switch means 3 and the second switch means 4 are operated in a well-balanced manner, and a three-level sine wave voltage is output as the converter voltage Vc between the input terminals of the rectifier 2. The flowing current can be converted into a sine wave. As a result, the reactor 5 can be made smaller than the conventional system in which the switch means is operated only once or several times in a half cycle of the power source to short-circuit the power source.

  Further, for example, it is possible to operate at a low frequency PWM of about 1 kHz to 5 kHz, and there is no cost increase due to noise countermeasures by the high frequency PWM, and it can be put into practical use at a low cost.

  This is because the input current can be realized in a substantially sine wave without the control of the input current only by outputting the converter voltage so as to be a sine wave, thereby enabling operation with a low-frequency PWM.

  In addition, the converter voltage is controlled by using a modulation method such as unipolar modulation, and the DC voltage is feedback controlled, so that it is possible to create a product group with a different number of models and specifications without searching for the parameters necessary to obtain the desired output voltage. It can be easily applied.

  Although the modulation is based on the PWM carrier frequency in the present embodiment, in the case of the unipolar modulation described above, the two triangular wave carriers that are carrier waves have a phase difference of 180 degrees from each other, and thus equivalently twice the carrier frequency. A frequency component (hereinafter sometimes referred to as a double component) is superimposed on the input current.

  The input current on which the double component of the carrier frequency is superimposed flows to the reactor 5. The reactor 5 generates a magnetic flux by the flowing current, but the magnetic flux having a component twice the carrier frequency becomes an electromagnetic excitation force in the reactor 5, and the vibration generated from the reactor 5 generates an electromagnetic sound due to the component twice the carrier frequency. Occur.

  The AC / DC converter according to the present invention can operate at a low PWM frequency. For example, even when 5 kHz is used, 10 kHz, which is twice as high, is a frequency in the audible region, and the higher the frequency, the higher the frequency. It becomes the sound that remains.

Therefore, a method for suppressing the electromagnetic noise of the reactor 5 caused by the PWM frequency will be described.
The waveform diagram of FIG. 8 is obtained by extracting the modulation type triangular wave carrier and two switching signals shown in FIG. 8A shows the waveform near the power supply zero cross, and FIG. 8B shows the waveform near the power supply peak.

  In FIG. 8 (a), the switching state of A is that the other switch is on and the other switch is off. Therefore, when this state corresponds to FIG. 4 (c), the switching state of B is as shown in FIG. Corresponding to 4 (a), the switching state of C corresponds to FIG. 4 (b). Thus, as the converter voltage Vc, the charging of the two capacitors 6 and 7 can be realized in a well-balanced manner when the state of Vo / 2 appears twice. Furthermore, since switching is performed symmetrically with half of the triangular wave, a double component of the carrier frequency is generated.

  Similarly, in FIG. 8B, the switching state of C corresponding to FIG. 4B, the switching state of D corresponding to FIG. 4D, and the switching state of A corresponding to FIG. As the converter voltage Vc changes, the state of Vo / 2 appears twice, whereby the two capacitors 6 and 7 can be charged in a well-balanced manner.

  In addition, since half the triangular wave has symmetry, if the double component of the carrier frequency is suppressed, the PWM is corrected to be asymmetric with the appearance rate of one period of the triangular wave being constant. By doing so, it is possible to suppress the electromagnetic sound peak at a component twice the carrier frequency. For example, if the appearance rate of A is 30% of one period of the triangular wave, it is distributed to each half of the triangular wave by 15% at the time of symmetry, but it is redistributed to 20% and 10%.

  However, when making it asymmetrical, it becomes necessary to generate a PWM for each half-carrier of the triangular wave, so that the processing load related to control increases accordingly. Since the present invention is a control that can reduce the processing load of the control, in order to take advantage of the feature, PWM generation for each carrier period, in other words, a component twice the carrier frequency while maintaining symmetry The purpose is to suppress the electromagnetic sound peak.

  In FIGS. 8A and 8B, three switching states appear, but two of them are in a state where voltage rectification, in other words, Vo / 2 is output in either case. Therefore, by manipulating the appearance ratio of the two voltage doubler rectification states, the double component of the carrier frequency is suppressed while maintaining symmetry. On the other hand, as described above, unless the two voltage doubler rectification states appear in a balanced manner, the voltage balance between the two capacitors is lost, the input current is also distorted, and the harmonic current cannot be suppressed. Therefore, the appearance ratio of the two voltage doubler rectification states is manipulated in a balanced manner.

As a method for controlling the appearance ratio of the voltage doubler rectification in a balanced manner, for example, with even number of carriers as a unit, the pulse width is changed to be different from the conventional one in half of the carriers, and the other half In the carrier, a pulse width corrected so that the pulse width in the entire even-numbered carrier becomes substantially the same as that in the conventional carrier is generated. For example, with the two carriers as a unit, the first carrier is operated so as to increase the appearance ratio in the double voltage rectification, and the next carrier, that is, the remaining carriers, is reduced in the appearance ratio of the one voltage doubler rectification.
A specific operation method is shown in FIG. FIGS. 9A and 9B are waveform diagrams for comparison with FIGS. 8A and 8B, and the alternate long and short dash line switching signal in FIG. 9 represents the switching signal shown in FIG. The triangular wave carrier in FIG. 9 is a waveform in which square waves that become Hi and Lo are superimposed for each carrier period. Since this square wave changes every carrier period, it becomes a waveform in which symmetry is maintained at half the frequency of the carrier. Since the switching signal is compared with the triangular wave, symmetry in one carrier is ensured, and when one carrier is considered as one group, the switching signal has a waveform different from that of the adjacent switching signal.

  Therefore, in the input current waveform, a side band of 1/2 component of the carrier frequency generated due to the square wave with respect to the double component and the double component of the carrier frequency is generated. For example, assuming that the carrier is 5 kHz, a side band of 2.5 kHz, which is half of 5 kHz, is centered on 10 kHz, which is twice that, and peaks appear at 7.5 kHz and 12.5 kHz in addition to 10 kHz. The so-called peak frequency disperses energy to other frequencies, and the peak of the double component of the carrier frequency is reduced rather than simple PWM control.

  In addition, in the switching condition other than Vo / 2, in FIG. 9A, the switching state of B (the switching state of D in FIG. 9B) is maintained at the same ratio in FIG. 8 and FIG. It does not affect the output converter voltage Vc, and the appearance rate of Vo / 2 is canceled by the adjacent switching signal (that is, the width of A in FIG. 8A is larger than the width of A in FIG. 8A). Although the width of A is larger, the width of C in FIG. 9A is correspondingly smaller than the width of C in FIG. Since the sum of the widths is equal in FIGS. 8 and 9, the capacitor voltage is not unbalanced, the input current waveform is improved, and the harmonic current can be suppressed.

  In addition, although a square wave having a frequency that is ½ of the carrier is superimposed on the triangular wave, the control processing capability is not limited to ½ of the carrier frequency. Needless to say, any method that generates PWM for each carrier frequency without increasing the frequency has the same effect. Furthermore, it can be considered that the appearance rate is manipulated in a random manner, but if the random number is not generated so that the average value of the random numbers generated from 0 to 1 becomes 0.5, the balance of the capacitor voltage will be lost. The processing load can be reduced compared to random numbers.

  In addition, although the description is made so that the square wave is superimposed, it is needless to say that a sine wave or a triangular wave has the same effect as described above, not a square wave. Furthermore, although it has been described by superimposing it on a triangular wave carrier that is a carrier wave, it goes without saying that the same effect can be obtained even if it is superimposed on the modulated wave side.

  Next, effects of the diodes 10 and 11 in FIG. 1 will be described. The diodes 10 and 11 have the same meaning as not being connected because the capacitor 6 or the capacitor 7 connected in parallel is in a non-current state and is turned off if the capacitor 6 or the capacitor 7 is in a normal state having electric charges and a positive voltage. is there.

  However, when the voltage supply from the AC power supply 1 is lost and the power consumption at the load 8 is not zero, the capacitors 6 and 7 are discharged until there is no charge (that is, the DC voltage becomes zero). At this time, the load 8 uniformly consumes charges from the capacitors 6 and 7 connected in series. However, if the capacitances of the capacitors 6 and 7 vary, the other capacitor is consumed even if the charge of the other capacitor is consumed. The charge remains in the capacitor, and the DC voltage does not become zero.

  Since the load 8 is consumed until the DC voltage becomes 0, the capacitor in which the charge is consumed has a negative charge amount, and a negative voltage is applied.

  Therefore, by connecting the diodes in antiparallel, it is possible to suppress the application of the negative voltage from being less than the forward voltage drop of the diode. Thereby, even when an electrolytic capacitor having a voltage polarity is used, it is possible to prevent the failure of the capacitor and improve the reliability of the capacitor.

Embodiment 2. FIG.
An example of another embodiment of the method for suppressing electromagnetic noise from the reactor 5 will be described with reference to FIG. The waveform of FIG. 10A is a diagram using the above-described constant frequency as a carrier wave, and the waveform of FIG. 10B is a carrier wave having a frequency that changes in synchronization with the power source described in the second embodiment. It is the figure used as.

  When a general carrier frequency as shown in FIG. 10A is constant, an electromagnetic sound based on the carrier frequency is generated. In the AC / DC converter of the present invention, a component twice the carrier frequency is generated. In addition, since the power supply voltage is approximately 0 near the zero cross of the AC power supply 1, no current flows regardless of how much switching is performed, and switching is performed wastefully.

  In the vicinity of the zero crossing, even if switching is reduced, the current waveform improvement effect is not impaired, and the harmonic current does not increase. Therefore, as shown in FIG. 10B, the carrier frequency is lowered near the zero cross. Further, the carrier frequency is increased toward the peak of the power supply, and the carrier frequency is decreased again by shifting to zero crossing after the peak.

  What is important here is that the carrier frequency and the power supply frequency are synchronized. In the case of non-synchronization, it can be assumed that the deviation increases with time and the lowered carrier frequency is near the peak of the power supply.

  When a carrier wave as shown in FIG. 10B is used, a plurality of carrier frequencies are generated for each half cycle of the power source, and electromagnetic sound is dispersed in a plurality of peak frequencies. Furthermore, unnecessary switching near the zero crossing is eliminated, and the circuit efficiency can be improved by reducing the switching loss. In addition, since the carrier frequency is managed, the peak frequency generated in the input current can be set intentionally. For example, the peak is not overlapped with the power supply harmonic current (for example, n times the power supply frequency). The frequency can be set by shifting, and an increase in high-order harmonic current can also be suppressed.

  Further, the PWM carrier frequency is synchronized with the AC power source 1 as shown in FIG. 10B, and a signal having a period lower than the carrier frequency is superimposed as shown in FIG. Needless to say, it has the same effect as described above.

Embodiment 3 FIG.
In the second embodiment, the suppression by the control of the peak electromagnetic sound having the double component of the carrier frequency has been described. In the third embodiment, the mechanical peak electromagnetic noise suppression method of the reactor 5 will be described.

FIG. 11 is a structural diagram of a reactor used in the AC / DC converter of the present invention.
In a reactor having a shape in which a winding is provided in the central part, the core in the central part becomes an electromagnet, attracting the core disposed horizontally above the head, and this vibrates, causing electromagnetic noise. An air gap (so-called gap) is interposed between the central electromagnet and the upper core to form an air layer. Therefore, by inserting, for example, a resin 41 having flexibility of a non-magnetic substance into this gap, the bending vibration of the upper core can be suppressed, and electromagnetic noise from the reactor 40 can be suppressed. Since the electromagnetic noise is generated from the core, it goes without saying that even if the winding is made of aluminum wire, copper wire, or other material, the same effect is obtained.

  It is said that a resin substance is inserted into the gap part, but it does not have to be made of resin, and can suppress electromagnetic noise from the reactor or a material that can suppress electromagnetic noise from the reactor. There is no problem using any structure that can be used.

  Furthermore, the repulsive force between the coils vibrates the side core of FIG. 11 and causes electromagnetic noise. Therefore, a structure is adopted in which the coil itself is fixed to the side core. For example, the coil itself may be impregnated with a resin material or fixed with resin. By combining such a mechanical suppression method and a control suppression method, it is possible to suppress unpleasant electromagnetic noise in the several kHz band.

Embodiment 4 FIG.
In the first embodiment, the configuration as shown in FIG. 1 has been described. However, the same control as in the first embodiment can be applied even in the circuit configuration as shown in FIG. In the basic circuit of FIG. 12, the operation of the IGBTs 3a and 4a which are switching elements is the same as in FIG. 1 and FIG. 12, and the diode rectifier 14 equivalently converts the IGBTs 3a and 4a of the unidirectional flow switching elements into bidirectional switching means. Therefore, it goes without saying that it has the same effect as described above.

  Further, by adopting the circuit configuration as shown in FIG. 12, the number of diodes in the path through which the current flows becomes half of that in FIG. , And 1/2. Thereby, it has the effect that the conversion efficiency of an AC / DC converter can be improved.

  Further, until now, the bidirectional switch means is composed of an IGBT and a diode rectifier. However, even with the configuration as shown in FIG. Needless to say, it has.

Embodiment 5 FIG.
FIG. 14 is a circuit block diagram showing the fifth embodiment of the present invention. 1 is an AC power source, 50 is a system impedance by a transmission line for supplying power, 51 is a product adopting the AC / DC converter of the present invention, 52 is another product connected to the power line, and 53 is an AC / DC converter. A voltage distortion suppression unit 54 that suppresses voltage distortion of the adopted product is a general household power supply wiring range.

  In the AC / DC converter according to the present invention, the double component of the carrier frequency is superimposed on the input current of the converter during the switching operation. Further, since the operation is possible with low frequency switching, the operating frequency is low and the energy of the low frequency is strong. Therefore, in a house or building where the system impedance 50 is large, switching by the AC / DC converter of the present invention is a carrier. Generates voltage distortion due to frequency.

  When the frequency of the voltage distortion is about several times the carrier frequency, other products 52 connected to the power supply line are affected by the voltage distortion. Specifically, when the other product 52 is, for example, an IH cooking heater, a sound of this voltage distortion frequency band can be heard from a pan cooked by the IH cooking heater, or in the case of a lighting fixture, the voltage distortion frequency. In the case of a ventilation fan, the sound of the frequency of this voltage distortion is heard during the operation of the ventilation fan.

  Such an effect varies depending on the system impedance 50, so it rarely occurs in general households, but very rarely, the system impedance is very high by passing from a high-voltage transmission line to a substation, pole transformer, etc. It may be large and the above-mentioned influence may occur.

  In that case, the voltage distortion suppression part 53 which suppresses voltage distortion is inserted. The insertion positions are at points A and B described in FIGS. 1 and 12. As described above, the voltage distortion in the AC / DC converter according to the present invention is a component twice the carrier frequency, and the peak is concentrated at that frequency. Therefore, the carrier frequency component of the input current that causes voltage distortion is specified by using a LC band-pass filter, π-type filter, resonance filter, or the like, as a component twice the carrier frequency.

  When the power consumption of the product 51 is large, a configuration such as a simple normal filter increases the size and is not practical. Therefore, a small and practical filter can be configured by specifying the double component of the carrier frequency.

  As an application example of the present invention, the present invention can be used for a power supply device for a load that consumes power by direct current. In particular, the heat pump water heater can be used as a power supply device for an inverter that is a DC / AC converter, and can be applied to an inverter that drives a permanent magnet motor, realizing energy savings, and a low-cost, low-noise AC / DC converter configuration. In addition to air conditioners, refrigerators, and washer / dryers, it can be applied to general household appliances such as refrigerators, dehumidifiers, showcases, vacuum cleaners, and can also be applied to fan motors, ventilation fans, hand dryers, etc. .

  1 power supply, 2 rectifier, 3 first switch means, 3a IGBT, 3b diode rectifier, 4 second switch means, 4a IGBT, 4b diode rectifier, 5 reactor, 6 first capacitor, 7 second capacitor, 8 Load, 9 Virtual AC power supply, 10 First diode, 11 Second diode, 12 Resistor, 13 Resistor, 20 Control means, 21 Voltage detector, 22 Zero-cross detector, 40 Reactor, 41 Resin, 50 Power transmission line system Impedance, 51 Products adopting the AC / DC power supply device of the present invention, 52 Other products, 53 Voltage distortion suppression unit, 54 General household power supply wiring range.

Claims (23)

A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors, the connection point of the first switch and the second switch are connected, and the first switch is set so that the voltage between the output terminals of the first rectifier becomes a predetermined value. And a control means for PWM controlling a switch group consisting of the second switches ,
The control means includes
3. An AC / DC converter characterized by generating a three-level substantially sinusoidal voltage between the input terminals of the rectifier by controlling the time ratio and generation order of each on / off combination of the switch group . A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors, the connection point of the first switch and the second switch are connected, and the first switch is set so that the voltage between the output terminals of the first rectifier becomes a predetermined value. And a control means for PWM controlling a switch group consisting of the second switches,
A diode is connected in antiparallel to each of the two capacitors ;
The control means includes
3. An AC / DC converter characterized by generating a three-level substantially sinusoidal voltage between the input terminals of the rectifier by controlling the time ratio and generation order of each on / off combination of the switch group . A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors, the connection point of the first switch and the second switch are connected, and the first switch is set so that the voltage between the output terminals of the first rectifier becomes a predetermined value. And a control means for PWM controlling a switch group consisting of the second switches,
Detecting means for detecting a DC voltage of the output terminal of the rectifier,
The control means includes
Based on the output of the detection means and a preset DC voltage command, the phase and amplitude of the voltage to be generated at the input terminal of the rectifier are calculated, and the switch group is turned on / off based on the calculated phase and amplitude. Determine timing
AC-DC converter characterized by this. A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors, the connection point of the first switch and the second switch are connected, and the first switch is set so that the voltage between the output terminals of the first rectifier becomes a predetermined value. And a control means for PWM controlling a switch group consisting of the second switches,
Detecting means for detecting a DC voltage of the output terminal of the rectifier,
A diode is connected in antiparallel to each of the two capacitors;
The control means includes
Based on the output of the detection means and a preset DC voltage command, the phase and amplitude of the voltage to be generated at the input terminal of the rectifier are calculated, and the switch group is turned on / off based on the calculated phase and amplitude. Determine timing
AC-DC converter characterized by this. The AC / DC converter according to claim 1, wherein each of the two capacitors is configured by connecting a plurality of capacitors in parallel or in series. AC-DC converter according to claim 1, characterized in that configured by connecting the first switch and the second respective switch plurality of switches in parallel or in series. The control means includes
AC-DC converter according to any one of claims 1 to 6, characterized in that PWM control the switches in the low frequency 1KHz～5kHz. The control means includes
By controlling the time ratio and the order of occurrence of each combination of on-off of the switches, and claim 3, 4, characterized in that to produce a substantially sinusoidal voltage of a three-level between the input terminals of the rectifier, wherein The AC / DC converter according to any one of claims 5 to 7, which is dependent on the items 3 and 4 . The switch group includes:
Operate to generate four rectification states: full wave rectification, first voltage rectification, second voltage rectification and power supply short circuit;
The control means includes
The AC / DC converter according to claim 8, wherein the switch group is operated by combining the four rectification states so that a voltage between the output terminals becomes a predetermined value. The control means includes
The AC / DC converter according to claim 8 or 9 , wherein the switch group is controlled so that an appearance rate of the first voltage doubler rectification and an appearance rate of the second voltage doubler rectification are the same. The control means includes
The AC / DC converter according to any one of claims 8 to 10 , wherein the substantially sine wave voltage is generated by applying a unipolar modulation or a dipolar modulation method to PWM control. Comprising detection means for detecting a DC voltage at the output terminal of the rectifier,
The control means includes
Based on the output of the detection means and a preset DC voltage command, the phase and amplitude of the voltage to be generated at the input terminal of the rectifier are calculated, and the switch group is turned on / off based on the calculated phase and amplitude. Timing is determined. The AC / DC converter according to any one of claims 1 and 2, and claims 5 to 11 subordinate to claims 1 and 2 . The control means includes
The PWM carrier, AC-DC converter according to any one of claims 1 to 12, characterized in providing an offset amount of preset voltage for each cycle of the PWM. The control means includes
AC-DC converter according to any one of claims 1 to 13, characterized in that to synchronize the period of the PWM cycle an AC power source. The control means includes
The AC / DC converter according to claim 14 , wherein the PWM cycle near the zero cross of the AC power supply is lower than the PWM cycle near the peak. AC-DC converter according to any one of claims 1 to 15, characterized in that by inserting a resin material in the gap of the reactor was fixed the reactor. A heat pump type water heater comprising the AC / DC converter according to any one of claims 1 to 16 . The heat pump type water heater according to claim 17, wherein a filter for attenuating a double component of a PWM frequency of the AC / DC converter is provided between the AC power supply and the AC / DC converter. An air conditioner comprising the AC / DC converter according to any one of claims 1 to 16 . The air conditioner according to claim 19, wherein a filter for attenuating a double component of a PWM frequency of the AC / DC converter is provided between the AC power supply and the AC / DC converter. A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors and the connection point of the first switch and the second switch are connected, and PWM control is performed so that the voltage between the output terminals of the first rectifier becomes a predetermined value. Control of an AC / DC converter including a switch group including a first switch that generates four rectification states of wave rectification, first voltage rectification, second voltage rectification, and power supply short-circuit, and the second switch In the method
The switch group is operated by combining the four rectification states so that the voltage between the output terminals of the AC / DC converter becomes a predetermined value ,
By controlling the time ratio and generation order of each on / off combination of the switch group, a three-level substantially sinusoidal voltage is generated between the input terminals of the rectifier.
A control method for an AC / DC converter characterized by the above. A first rectifier and a second rectifier connected to an AC power source via a reactor;
Two capacitors connected in series between the output terminals of the first rectifier;
A first switch and a second switch connected in series between output terminals of the second rectifier;
The connection point between the capacitors and the connection point of the first switch and the second switch are connected, and PWM control is performed so that the voltage between the output terminals of the first rectifier becomes a predetermined value. A first switch that generates four rectification states of a wave rectification, a first voltage doubler rectification, a second voltage doubler rectification, and a power supply short circuit, and a switch group including the second switch,
In a control method for an AC to DC converter comprising detection means for detecting a DC voltage at an output terminal of the first rectifier,
The switch group is operated by combining the four rectification states so that the voltage between the output terminals of the AC / DC converter becomes a predetermined value,
Based on the output of the detection means and a preset DC voltage command, the phase and amplitude of the voltage to be generated at the input terminal of the rectifier are calculated, and the switch group is turned on / off based on the calculated phase and amplitude. Determine timing
A control method for an AC / DC converter characterized by the above. 23. The method of controlling an AC / DC converter according to claim 21 , wherein the switch group is controlled so that the first voltage doubler rectification and the second voltage doubler rectification have the same generation ratio. .

Images