JP4930866B2 - Failure detection device for power element - Google Patents

Description

  The present invention relates to an apparatus for detecting a failure of a power semiconductor element used in an inverter circuit or the like.

  Japanese Laid-Open Patent Publication Nos. 4-185228 (Patent Document 1) and 2001-197724 (Patent Document 2) and the like are techniques for detecting an overcurrent of a power semiconductor element and preventing the destruction of the semiconductor element. The technique described in is known.

  Japanese Patent Laid-Open No. 4-185228 (Patent Document 1) discloses a short-circuit protection device in parallel with a switching element that constitutes an inverter, only when a low-voltage source for detecting a short circuit, a variable resistor, and a switching element are in conduction. A diode that applies a forward bias to the switch element by a low voltage source is connected in series. As a result of the large current flowing through the switch element, the voltage extracted from the voltage extraction terminal of the variable resistor becomes a value exceeding the threshold value. At this time, the switch element is forcibly turned off, and the switch element is prevented from being damaged.

IGBT (Insulated) described in Japanese Patent Laid-Open No. 2001-197724 (Patent Document 2)
In the gate drive circuit of a gate bipolar transistor, the voltage between the collector and emitter of the IGBT is detected through a resistor and a diode. When this detected value exceeds the voltage of the reference power supply built in the overcurrent determination circuit, it is determined that the IGBT is in an overcurrent state, the soft cutoff circuit operates, and the IGBT is gently turned off.

JP-A-4-185228 JP 2001-197724 A

  The present invention relates to a technique for detecting that a power semiconductor element is broken or deteriorated, unlike the technique disclosed in the above-described prior art documents.

IGBTs and MOSFETs (Metal Oxide Semiconductor) that make up power modules
When a power element such as a field effect transistor fails for some reason, there is a high possibility that the main electrodes of the power element are short-circuited. At this time, when the load connected to the power module is an inductive load such as a synchronous motor, the power module may be overloaded by a counter electromotive force generated in the load due to a short-circuit current between the main electrodes.

  Therefore, in order to protect the entire system using the power module, it is necessary to constantly monitor the withstand voltage of the power element and cut off the output current from the power module to the load when the power element loses the withstand voltage.

  An object of the present invention is to provide a failure detection device capable of detecting a failure of a power element used in a power module with a simple configuration.

  The present invention is a power element failure detection apparatus in a power circuit including a switching element in which a main current flowing from a first main electrode to a second main electrode changes according to a control signal. According to the power element failure detection apparatus of the present invention, the cathode is connected to the first main electrode, the first rectifying element, one end is connected to the anode of the first rectifying element, and the other end is connected to the second main rectifying element. A first resistance element to which a positive voltage is applied to the electrode; and a failure determination unit that determines whether or not a failure determination condition for detecting a failure of the switching element is satisfied. Here, the failure determination condition includes a condition that the monitoring voltage between the anode of the first rectifying element and the second main electrode is lower than a predetermined first reference voltage. The first reference voltage is lower than the sum of the voltage between the first and second main electrodes of the switching element in the on state and the forward voltage of the first rectifying element, and the first rectifying element Is set to a voltage higher than the forward voltage.

  According to the present invention, the failure of the power element can be detected by a simple configuration in which the voltage between the first and second main electrodes of the power element is monitored via the first rectifying element.

It is a figure for demonstrating the structure of the motor drive device 10 with which the failure detection apparatus of Embodiment 1 of this invention is applied. It is explanatory drawing for demonstrating the structure of 1 A of failure detection apparatuses used for IGBTQ1 of FIG. 3 is a flowchart showing a failure detection process in the gate control computer 20 of FIG. 2. FIG. 3 is a timing diagram schematically showing an example of a voltage / current waveform of each part in FIG. 2. It is a circuit diagram which shows the structure of the failure detection apparatus 1B of Embodiment 2 of this invention. FIG. 6 is a timing chart schematically showing an example of voltage / current waveforms at various parts in FIG. 5. It is a circuit diagram which shows the structure of 1 C of failure detection apparatuses of Embodiment 3 of this invention. FIG. 8 is a timing chart schematically showing an example of voltage / current waveforms at various parts in FIG. 7. It is a block diagram which shows the structure of the failure detection apparatus 1D of Embodiment 4 of this invention. It is a graph which shows the temperature dependence of the current-voltage characteristic of IGBT. FIG. 10 is a block diagram showing an example of the configuration of the sample and hold circuit 26 of FIG. 9. FIG. 10 is a timing diagram schematically illustrating an example of a voltage / current waveform of each unit in FIG. 9.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. The same or corresponding parts are denoted by the same reference numerals, and the description thereof will not be repeated.

  In each of the following embodiments, an IGBT is taken as an example of a power element, but the present invention is also applicable to other power elements such as a MOSFET and a bipolar translator.

[Embodiment 1]
In the first embodiment, a failure detection apparatus 1A that detects that a short circuit failure has occurred in an IGBT will be described.

  FIG. 1 is a diagram for explaining a configuration of a motor drive device 10 to which the failure detection apparatus according to the first embodiment of the present invention is applied.

  Referring to FIG. 1, motor drive device 10 supplies three-phase AC power to motor 18 from output nodes U, V, and W. The motor 18 is a three-phase AC motor, and is equivalently expressed using winding resistances R1, R2, and R3 and winding inductances L1, L2, and L3 of each phase.

  The motor drive device 10 includes a DC power supply V1, a smoothing capacitor C1 connected in parallel to the DC power supply V1, an inverter circuit 11, and connection breakers 12, 14, and 16. Instead of the DC power supply V1, an AC power supply and a rectifier circuit can be used.

  The inverter circuit 11 is a three-phase bridge circuit, and for each phase, two N-channel IGBTs connected in series between the high voltage node P and the low voltage node N (grounded GND) (the IGBTs Q1, Q2, IGBTQ3, Q4 for the V phase and IGBTQ5, Q6) for the W phase. Inverter circuit 11 converts the output of DC power supply V1 into AC power by switching on / off IGBTs Q1 to Q6. The converted AC power is output from the output nodes N1, N2, and N3 for each phase. For example, pulse width modulation (PWM) is used as a conversion method from direct current to alternating current.

  Inverter circuit 11 further includes flywheel diodes D1 to D6 connected in parallel to IGBTs Q1 to Q6 and gate drive circuits GD1 to GD6 for applying a gate voltage between the gates and emitters of IGBTs Q1 to Q6.

  The flywheel diodes D1 to D6 are provided to recirculate the induced electromotive force generated in the motor 18 when the corresponding IGBTs Q1 to Q6 are in the off state. The cathodes of flywheel diodes D1 to D6 are connected to collectors that are first main electrodes of IGBTs Q1 to Q6, respectively, and the anodes of flywheel diodes D1 to D6 are connected to emitters that are second main electrodes of IGBTs Q1 to Q6. Each is connected.

  Gate driving circuits GD1 to GD6 output high / low level gate voltages between the gates and emitters of IGBTs Q1 to Q6 in accordance with control signals input to input nodes S1 to S6. The control signal is supplied from a control computer (reference numeral 20 in FIG. 2) not shown.

  The connection breakers 12, 14, and 16 receive an output from a failure detection device, which will be described later, and block currents of respective phases supplied from the inverter circuit 11 to the motor 18 when the IGBTs Q1 to Q6 are short-circuited.

  Next, with reference to FIGS. 2 to 4, a failure detection device that detects a short-circuit failure of the IGBTs Q <b> 1 to Q <b> 6 used in the above-described inverter circuit 11 will be described. Since a failure detection device having the same configuration is provided corresponding to each of the IGBTs Q1 to Q6, hereinafter, the failure detection device 1A provided in the IGBT Q1 will be described as a representative.

  FIG. 2 is a circuit diagram showing a configuration of failure detection apparatus 1A used in IGBT Q1 in FIG. In the circuit diagram of FIG. 2, the common potential VN1 serving as a potential reference is the potential of the emitter (node N1) of the IGBT Q1.

  Referring to FIG. 2, failure detection apparatus 1A includes the aforementioned control computer 20 and failure detection circuit 3 provided in gate drive circuit GD1. The failure detection circuit 3 includes a high voltage diode D7, resistance elements R4 and R5, DC power supplies V2 and V3, and an open collector output comparator CA1. In addition to the failure detection circuit 3, the gate drive circuit GD1 includes a drive circuit UA that amplifies a control signal from the gate control computer 20 and outputs the amplified signal to the gate of the IGBT Q1. Here, as shown in FIG. 2, the comparator CA1 operates at the power supply voltage VCC1 (for example, 5 volts), and the drive circuit UA operates at the power supply voltage VCC2 (for example, 15 volts).

  In the failure detection circuit 3 of FIG. 2, the cathode of the diode D7 is connected to the collector of the IGBT Q1. The anode of the diode D7 is connected to one end of the resistance element R4 and the inverting input terminal of the comparator CA1.

  DC power supply V2 is connected between the other end of resistance element R4 and node N1. DC power supply V3 is connected between the non-inverting input terminal of comparator CA1 and node N1. DC power supply V3 provides a reference potential for comparison with the anode potential of diode D7.

  Here, the voltage of the DC power supply V2 is sufficiently smaller than the voltage of several hundred volts applied between the nodes P and N of the inverter circuit 11, and is set to 5 volts, for example. The voltage of DC power supply V3 is lower than the sum of the collector-emitter voltage (hereinafter also referred to as “on voltage”) of IGBT Q1 in the on state and the forward voltage of diode D7, and the forward direction of diode D7. The voltage is set higher than the voltage. For example, if the on-voltage of the IGBT Q1 is 0.6 volts and the forward voltage of the diode D7 is 0.6 volts, the voltage of the DC power supply V3 is a value between 0.6 and 1.2 volts, for example, 0.8 Set to bolt. In consideration of the fact that the input voltage is around 0 volts, the potential of the negative power supply node VEE connected to the comparator CA1 is set lower than the potential VN1 of the reference node N1.

  The output terminal of the comparator CA1 is connected to the power supply node VCC1 through the resistance element R5. The output of the comparator CA1 is open when the input voltage at the non-inverting input terminal is greater than the input voltage at the inverting output terminal. At this time, the potential of the output terminal of the comparator CA1 becomes a high level potential pulled up through the resistance element R5.

  The gate control computer 20 receives the output signal of the comparator CA1 through the photocoupler PC. The photocoupler PC is provided to electrically insulate the failure detection circuit 3 from the gate control computer 20. In the photocoupler PC, when the input signal is at a high level, the phototransistor PT that has received light emitted from the photodiode PD is turned on.

  The gate control computer 20 outputs a signal for interrupting the connection breaker 12 based on the control signal given to the gate of the IGBT Q1 and the output of the comparator CA1. The gate control computer 20 and the connection breaker 12 are connected via a photocoupler PC.

  Next, the operation of the failure detection apparatus 1A will be described. From a functional viewpoint, of the failure detection device 1A, the comparator CA1, the DC power supply V3, the resistance element R5, and the gate control computer 20 constitute a failure determination unit 2A for determining a failure of the IGBT Q1. Failure determination unit 2A monitors the potential of the anode of diode D7 with respect to node N1. Then, failure determination unit 2A determines a short circuit failure of IGBT Q1 based on the anode potential of diode D7 being monitored.

  The specific operation modes of the failure determination unit 2A will be described by dividing them into the following four operation modes (i) to (iv).

  (I) The IGBT Q1 is in a normal and on state with a high voltage (for example, several hundred volts) being applied between the nodes P and N in FIG.

  In this case, the sum (for example, 1.2 volts) of the on-voltage of the IGBT Q1 and the forward voltage of the diode D7 is input to the inverting input terminal of the comparator CA1 connected to the anode of the diode D7. As a result, the input voltage at the inverting input terminal of the comparator CA1 becomes larger than the voltage (for example, 0.8 volts) of the DC power supply V3, so that the output of the comparator CA1 is at a low level (common potential VN1). That is, the failure determination unit 2A determines that a short circuit failure has not occurred.

  (Ii) The IGBT Q1 or the flywheel diode D1 is short-circuited regardless of whether or not a voltage is applied between the nodes P and N.

  In this case, since the potential of the collector of IGBT Q1 is substantially 0 volts, the forward voltage (for example, 0.6 volts) of diode D7 is input to the inverting input terminal of comparator CA1. As a result, the input voltage at the inverting input terminal of the comparator CA1 becomes smaller than the voltage (for example, 0.8 volts) of the DC power supply V3, so that the output of the comparator CA1 is pulled up via the resistance element R5. Potential). That is, the failure determination unit 2A determines that a short circuit failure has occurred.

  (Iii) When a high voltage (for example, several hundred volts) is applied between the nodes P and N, the IGBT Q1 is normal and off, and the flywheel diode D1 is off.

  In this case, since the high voltage between the nodes P and N is blocked by the high breakdown voltage diode D7, the voltage of the DC power supply V2 (for example, 5 volts) is input to the inverting input terminal of the comparator CA1. As a result, the input voltage at the inverting input terminal becomes higher than the voltage of the DC power supply V3 (for example, 0.8 volts), so the output of the comparator CA1 is at a low level (common potential VN1). That is, the failure determination unit 2A determines that a short circuit failure has not occurred.

  (Iv) When a high voltage (for example, several hundred volts) is applied between the nodes P and N, the IGBT Q1 is normal and off, and the flywheel diode D1 is on.

  In this case, the potential of the collector of the IGBT Q1 becomes a negative potential with respect to the common potential VN1 by the forward voltage of the flywheel diode D1. As a result, the input voltage at the inverting input terminal of the comparator CA1 is a value obtained by subtracting the forward voltage of the flywheel diode D1 from the forward voltage of the diode D7 (less than 0 volts). Accordingly, since the input voltage at the inverting input terminal is smaller than the voltage of the DC power supply V3 (for example, 0.8 volts), the output of the comparator CA1 becomes high level. That is, only the detection result of the failure detection circuit 3 means that a short-circuit failure has occurred as in the operation mode (ii).

  Therefore, in the failure detection device 1A according to the first embodiment, the gate control computer 20 considers the logic level of the control signal supplied to the IGBT Q1 so that the short-circuit failure is not determined in the operation mode (iv) described above. Determine the failure.

  FIG. 3 is a flowchart showing a failure detection process in the gate control computer 20 of FIG.

  Referring to FIG. 3, in step S1, the computer 20 is in a waiting state until the output of the comparator CA1 becomes high level. When the output of the comparator CA1 becomes high level (YES in step S1), the process proceeds to step S2.

  In step S2, the computer 20 determines whether or not the control signal supplied to the gate of the IGBT Q1 is at a high level. The above-described operation mode (iv) occurs when the control signal is at a low level and the IGBT Q1 is in an off state, and therefore excludes the case where the control signal is at a low level (NO in step S2).

  When both steps S1 and S2 are YES, the process proceeds to step S3. In step S3, the computer 20 determines that the IGBT Q1 is a short circuit failure. And the computer 20 outputs the signal which interrupts | blocks the connection circuit breaker 12, and a failure detection process is complete | finished.

  In summary, failure determination unit 2A of the first embodiment determines whether or not the following failure determination condition is satisfied in order to detect a failure in IGBT Q1. The failure determination condition at this time is that the monitoring voltage between the anode of the diode D7 and the emitter of the IGBT Q1 is lower than the power supply voltage V3, and a control signal for turning on the IGBT Q1 is given to the IGBT Q1.

Next, the operation of the failure detection apparatus 1A will be described with reference to specific voltage / current waveforms.
FIG. 4 is a timing chart schematically showing an example of the voltage / current waveform of each part in FIG. In FIG. 4, the vertical axis indicates, in order from the top, the gate voltage of the IGBT Q1, the collector voltage of the IGBT Q1, the anode voltage of the diode D7, the output voltage of the comparator CA1, the collector current (main current) of the IGBT Q1, and the current of the flywheel diode D1. . The horizontal axis is the elapsed time.

  Referring to FIG. 4, the gate voltage of IGBT Q1 is 0 volt in the period from time t0 to t1, t2 to t3, t4 to t5, t6 to t7, and t8 to t11 (hereinafter referred to as off period). IGBTQ1 is in an off state. In the case of FIG. 4, since the flywheel diode D1 is always in an off state, when the IGBT Q1 is normal, the collector voltage in the off section is a high voltage of 600 volts. Since this high voltage is blocked by the diode D7, the anode voltage of the diode D7 is equal to 5 volts which is the voltage of the DC power supply V2.

  On the other hand, in the period from time t1 to t2, t3 to t4, t5 to t6, t7 to t8, t11 to t12 (hereinafter referred to as the on period), the gate voltage of the IGBT Q1 is 15 volts, and the IGBT Q1 is in the on state. is there. When the IGBT Q1 is normal in the ON section of FIG. 4, the collector voltage is equal to the ON voltage of the IGBT Q1. Therefore, the anode voltage of the diode D7 is about 1.2 to 2 volts which is the sum of the ON voltage of the IGBT Q1 and the forward voltage of the diode D7. Further, in the ON section of FIG. 4, the collector current of the IGBT Q1 is gradually increased by PWM control. The magnitude of the collector current is about 10 to 300 amperes.

  At time t9 in the middle of times t8 to t11 in the off section, the IGBT Q1 is short-circuited. As a result, the collector voltage of the IGBT Q1 gradually decreases to 0 volts, and the anode voltage of the diode D7 gradually decreases to the forward voltage (0.6 volts). At the same time, a collector current (short circuit current) is generated. The short-circuit current reaches several thousand amperes at maximum.

  At time t10, when the anode voltage of the diode D7 becomes lower than the power supply voltage V3 (0.8 volts) which is the reference voltage, the output voltage of the comparator CA1 is switched from the low level (0 volts) to the high level (5 volts).

  At time t11, the gate control computer 20 outputs a control signal for switching the IGBT Q1 from the off state to the on state. At this time, the gate control computer 20 detects that the output voltage of the comparator CA1 is at a high level, and outputs a signal for cutting off the connection breaker 12. Eventually, the collector current of the IGBT Q1 is cut off and becomes 0 amperes.

  As described above, according to the failure detection apparatus 1A of the first embodiment, the voltage between the collector and the emitter of the IGBT Q1 is monitored via the diode D7. At this time, the high voltage applied between the collector and the emitter when the IGBT Q1 is in the off state is blocked by the high voltage diode D7. Therefore, it is possible to monitor the voltage between the collector and emitter of the IGBT Q1 by a simple method using the comparator CA1.

  Further, failure detection apparatus 1A determines the failure of IGBT Q1 by combining the condition that the anode voltage of diode D7 is lower than the reference voltage (V3) and the condition that the logic level of the gate control signal is high. . Therefore, since the flywheel diode D1 is in the ON state, it is possible to determine whether the IGBT Q1 is short-circuited, as distinguished from the normal case where the collector-emitter voltage of the IGBT Q1 is low.

[Embodiment 2]
FIG. 5 is a circuit diagram showing a configuration of failure detection apparatus 1B according to the second embodiment of the present invention. The failure detection device 1B of FIG. 5 is a modification of the failure detection device 1A of the first embodiment, and detects that the IGBT Q1 has a short circuit failure. In the circuit diagram of FIG. 5, the common potential VN1 that serves as a reference for the potential is the potential of the emitter (node N1) of the IGBT Q1. Since the gate drive circuit and the failure detection device having the same configuration are provided corresponding to each of the IGBTs Q1 to Q6, hereinafter, the gate drive circuit GD1B and the failure detection device 1B provided in the IGBT Q1 will be described as representatives.

  Referring to FIG. 5, gate drive circuit GD1B includes failure detection device 1B and drive circuit UA. Here, failure detection apparatus 1B includes high-voltage diode D7, resistance element R4, DC power supply V2, and failure determination unit 2B. Fault determination unit 2B includes open collector output comparators CA1 and CA2, DC power supplies V3 and V4, and resistance element R5. Drive circuit UA is similar to that of the first embodiment, and operates at power supply voltage VCC2 (for example, 15 volts).

  In the failure detection apparatus 1B of FIG. 5, the cathode of the diode D7 is connected to the collector of the IGBT Q1. The anode of the diode D7 is connected to one end of the resistance element R4, the inverting input terminal of the comparator CA1, and the non-inverting input terminal of the comparator CA2.

  DC power supply V2 is connected between the other end of resistance element R4 and node N1. DC power supply V3 is connected between the non-inverting input terminal of comparator CA1 and node N1. DC power supply V4 is connected between the inverting input terminal of comparator CA2 and node N1. The DC power supplies V3 and V4 are reference potentials for comparison with the anode potential of the diode D7.

  Here, the voltage of the DC power supply V2 is sufficiently smaller than the voltage of several hundred volts applied between the nodes P and N of the inverter circuit 11, and is set to 5 volts, for example. The voltage of DC power supply V3 is set to a voltage lower than the sum of the on-voltage of IGBT Q1 and the forward voltage of diode D7 and higher than the forward voltage of diode D7. For example, if the on-voltage of the IGBT Q1 is 0.6 volts and the forward voltage of the diode D7 is 0.6 volts, the voltage of the DC power supply V3 is a value between 0.6 and 1.2 volts, for example, 0.8 Set to bolt. The voltage of the DC power supply V4 is set to a value higher than the value obtained by subtracting the forward voltage of the flywheel diode D1 from the forward voltage of the diode D7 and lower than the forward voltage of the diode D7. For example, instead of the DC power supply V4, the inverting input terminal of the comparator CA2 is connected to the node N1, and the voltage of the inverting input terminal is set to 0 volts. In consideration of the fact that the input voltage is around 0 volts, the potential of the negative power supply node VEE connected to the comparators CA1 and CA2 is set to a potential lower than the potential VN1 of the reference node N1. . The positive power supply voltage VCC1 supplied to the comparators CA1 and CA2 is, for example, 5 volts.

  Output terminals of comparators CA1 and CA2 are connected to power supply node VCC1 through resistance element R5. Comparators CA1 and CA2 constitute a window comparator. Therefore, when the voltage of the anode of the diode D7 is larger than the voltage of the DC power supply V4 and smaller than the voltage of the DC power supply V3, the voltages at the output terminals of the comparators CA1 and CA2 are pulled up via the resistance element R5. It becomes a high level voltage.

  The output terminals of the comparators CA1 and CA2 are further connected to the connection breaker 12 via the photocoupler PC. When the potentials at the output terminals of the comparators CA1 and CA2 become high level, the connection breaker 12 is cut off.

  Next, the operation of the failure detection apparatus 1B will be described. Failure detection device 1B differs from failure detection device 1A in FIG. 2 in that it further includes a comparator CA2 and a DC power supply V4. This makes it possible to determine the failure of the IGBT Q1 when the flywheel diode D1 is in the ON state without using the gate control computer 20.

  A specific operation mode of the failure determination unit 2B will be described by dividing it into the following four operation modes (i) to (iv).

  (I) The IGBT Q1 is normal and in the on state in a state where a high voltage (for example, several hundred volts) is applied between the nodes P and N in FIG.

  In this case, the voltage of the anode of the diode D7 is the sum of the ON voltage of the IGBT Q1 and the forward voltage of the diode D7 (for example, 1.2 volts). Since the voltage at the inverting input terminal of the comparator CA1 is greater than the voltage of the DC power supply V3 (for example, 0.8 volts), the output of the comparator CA1 is at a low level (common potential VN1). On the other hand, the voltage at the non-inverting input terminal of the comparator CA2 is higher than the voltage of the power supply voltage V4 (for example, 0 volts), so the output of the comparator CA2 is open. Therefore, the output voltages of the comparators CA1 and CA2 by the wired AND are at a low level. That is, the failure determination unit 2A determines that a short circuit failure has not occurred.

  (Ii) The IGBT Q1 or the flywheel diode D1 is short-circuited regardless of whether or not a voltage is applied between the nodes P and N.

  In this case, since the potential of the collector of the IGBT Q1 is approximately 0 volts, the anode voltage of the diode D7 is equal to the forward voltage (eg, 0.6 volts) of the diode D7. Since the voltage at the inverting input terminal of the comparator CA1 is smaller than the voltage of the DC power supply V3 (for example, 0.8 volts), the output of the comparator CA1 is open. In addition, since the voltage at the non-inverting input terminal of the comparator CA2 is larger than the voltage of the power supply voltage V4 (for example, 0 volts), the output of the comparator CA2 is also opened. As a result, the voltage at the output terminals of the comparators CA1 and CA2 by the wired AND becomes high level (voltage pulled up through the resistance element R5). That is, the failure determination unit 2A determines that a short circuit failure has occurred.

  (Iii) When a high voltage (for example, several hundred volts) is applied between the nodes P and N, the IGBT Q1 is normal and off, and the flywheel diode D1 is off.

  In this case, since the high voltage between the nodes P and N is blocked by the high breakdown voltage diode D7, the anode voltage of the diode D7 is equal to the voltage of the DC power supply V2 (for example, 5 volts). Since the voltage at the inverting input terminal of the comparator CA1 is greater than the voltage of the DC power supply V3 (for example, 0.8 volts), the output of the comparator CA1 is at a low level (common potential VN1). On the other hand, since the voltage at the non-inverting input terminal of the comparator CA2 is larger than the voltage of the power supply voltage V4 (for example, 0 volts), the output of the comparator CA2 is opened. As a result, the output voltages of the comparators CA1 and CA2 due to the wired AND become low level. That is, the failure determination unit 2A determines that a short circuit failure has not occurred.

  (Iv) When a high voltage (for example, several hundred volts) is applied between the nodes P and N, the IGBT Q1 is normal and off, and the flywheel diode D1 is on.

  In this case, the potential of the collector of the IGBT Q1 becomes a negative potential with respect to the common potential VN1 by the forward voltage of the flywheel diode D1. As a result, the anode voltage of the diode D7 becomes a value obtained by subtracting the forward voltage of the flywheel diode D1 from the forward voltage of the diode D7 (less than 0 volts). Therefore, the input voltage at the inverting input terminal of the comparator CA1 is smaller than the voltage of the DC power supply V3 (for example, 0.8 volts), so the output of the comparator CA1 is open. On the other hand, the voltage at the non-inverting input terminal of the comparator CA2 is lower than the voltage of the power supply voltage V4 (for example, 0 volts), and thus becomes low level (common potential VN1). As a result, the output voltages of the comparators CA1 and CA2 due to the wired AND become low level. That is, the failure determination unit 2A determines that a short circuit failure has not occurred.

  In summary, failure determination unit 2B of the second embodiment determines that the monitoring voltage between the anode of diode D7 and the emitter of IGBT Q1 is lower than power supply voltage V3 and higher than power supply voltage V4. As a result, it is detected that a short circuit failure has occurred in at least one of IGBT Q1 and flywheel diode D1.

  FIG. 6 is a timing chart schematically showing an example of the voltage / current waveform of each part of FIG. In FIG. 6, the vertical axis represents the gate voltage of the IGBT Q1, the collector voltage of the IGBT Q1, the anode voltage of the diode D7, the output voltage of the comparator CA1, the output voltage of the comparator CA2, the collector current of the IGBT Q1, and the current of the flywheel diode D1 in order from the top. Indicates. However, although the outputs of the comparators CA1 and CA2 are actually ANDed by a wired AND, they are shown separately for convenience of explanation in FIG. The horizontal axis is the elapsed time.

  Referring to FIG. 6, the gate voltage of IGBT Q <b> 1 is 0 volt in the period of time t <b> 0 to t <b> 1, t <b> 2 to t <b> 3, t <b> 4 to t <b> 5, t <b> 6 to t <b> 7, t8 to t <b> 11. IGBTQ1 is in an off state. On the other hand, in the period from time t1 to t2, t3 to t4, t5 to t6, t7 to t8, t11 to t12 (hereinafter referred to as the on period), the gate voltage of the IGBT Q1 is 15 volts, and the IGBT Q1 is in the on state. is there.

  In the case of FIG. 6, the collector current of the IGBT Q1 gradually increases by PWM control at times t1 to t2, t3 to t4, t5 to t6, and t7 to t8 in the on period. The collector current of IGBTQ1 is about 10 to 300 amperes. At this time, since the collector voltage of the IGBT Q1 is equal to the ON voltage, the anode voltage of the diode D7 has a value of about 1.2 to 2 volts.

  The flywheel diode D1 is in the off state at times t0 to t1 in the off section of FIG. At this time, the high voltage of 600 volts applied between the collector and emitter of the IGBT Q1 is blocked by the diode D7, so that the anode voltage of the diode D7 is equal to 5 volts which is the voltage value of the DC power supply V2.

  On the other hand, at times t2 to t3, t4 to t5, t6 to t7, and t8 to t11 in the off section, current flows through the flywheel diode D1 due to the induced electromotive force generated in the load. At this time, the collector potential of the IGBT Q1 is lower than the emitter potential by the forward voltage of the flywheel diode D1. As a result, the anode voltage of the diode D7 is approximately -2 to 0 volts.

  As described above, the anode voltage of the diode D7 does not become a value between the reference voltage V4 (0 volt) and the reference voltage V3 (0.6 volt) during the period from time t0 to t8.

  The flywheel diode D1 is short-circuited at time t9 in the middle of the time t8 to t11 in the off section. As a result, a short circuit current flows through the flywheel diode D1. The collector voltage of the IGBT Q1 is a negative voltage at times t8 to t9, but changes to 0 volts due to a short-circuit failure of the flywheel diode D1. As a result, the anode voltage of the diode D7 changes from −2 to 0 volts to 0.6 volts, which is the forward voltage of the diode D7.

  At time t10, the anode voltage of the diode D7 becomes higher than the reference voltage V4 (0 volt), and the output of the comparator CA2 is switched from the low level (0 volt) to the high level (5 volt). As a result, the outputs of the comparators CA1 and CA2 both become high level, and a short circuit failure is detected.

  As described above, according to failure detection apparatus 1B of the second embodiment, the voltage between the collector and the emitter of IGBT Q1 is monitored via diode D7, as in the first embodiment. As a result, it is possible to determine a short circuit failure of the IGBT Q1 with a simple configuration using the comparators CA1 and CA2.

  Further, failure detection apparatus 1B determines the failure of IGBT Q1 under the determination condition that the anode voltage of diode D7 is greater than reference voltage (V4) and less than reference voltage (V3). Therefore, it is possible to determine the failure of the IGBT Q1 in distinction from the normal case where the flywheel diode D1 is in the ON state. Further, since the failure determination is performed using the comparators CA1 and CA2, the failure determination can be performed more quickly than in the first embodiment using the control computer 20.

[Embodiment 3]
FIG. 7 is a circuit diagram showing a configuration of failure detection apparatus 1C according to the third embodiment of the present invention. The failure detection device 1C of FIG. 7 is a modification of the failure detection device 1B of the second embodiment, and detects that the IGBT Q1 has a short circuit failure. Since the gate drive circuit and the failure detection device having the same configuration are provided corresponding to each of the IGBTs Q1 to Q6, hereinafter, the gate drive circuit GD1C and the failure detection device 1C provided in the IGBT Q1 will be described as representatives.

  Referring to FIG. 7, gate drive circuit GD1C includes failure detection device 1C and drive circuit UA. Here, the failure detection device 1C of FIG. 7 differs from the failure detection device 1B of FIG. 5 in that it further includes a logic circuit LA1. Since the failure detection device 1C in FIG. 7 is common to the failure detection device 1B in FIG. 5 for other points, the description of the common points will not be repeated. Drive circuit UA is the same as in the first and second embodiments, and operates at power supply voltage VCC2 (for example, 15 volts).

  Referring to FIG. 7, logic circuit LA1 is an AND circuit that outputs a logical product of two input signals. One input terminal of the logic circuit LA1 is connected to the output terminals of the comparators CA1 and CA2. The other input terminal of the logic circuit LA1 is connected to the input node S1 of the control signal of the IGBT Q1. The output terminal of the logic circuit LA1 is connected to the photocoupler PC. Therefore, the logic circuit LA1 outputs a high-level output signal when the output signals of the comparators CA1 and CA2 are at a high level and the control signal of the IGBT Q1 is at a high level. At this time, failure determination unit 2C determines that IGBT Q1 has a short circuit failure. Then, the failure determination unit 2C blocks the connection breaker 12 connected via the photocoupler PC. Logic circuit LA1 operates at power supply voltage VCC1 (for example, 5 volts).

  The on-voltage of the IGBT Q1 varies depending on the collector current and also varies depending on the temperature. Therefore, it is difficult to set the voltage value of the power supply voltage V3, which is a threshold for determining a short circuit failure. Therefore, the failure detection apparatus 1C of FIG. 7 can detect the short-circuit failure of the IGBT Q1 with higher accuracy by taking the logical product of the outputs of the comparators CA1 and CA2 and the control signal of the IGBT Q1.

  FIG. 8 is a timing chart schematically showing an example of the voltage / current waveform of each part of FIG. In FIG. 8, the vertical axis indicates the gate voltage of the IGBT Q1, the collector voltage of the IGBT Q1, the anode voltage of the diode D7, the output voltage of the comparator CA1, the output voltage of the comparator CA2, the output voltage of the logic circuit LA1, and the collector current of the IGBT Q1, in order from the top. The current of the flywheel diode D1 is shown. However, although the outputs of the comparators CA1 and CA2 are actually ANDed by a wired AND, they are shown separately for convenience of explanation in FIG. The horizontal axis is the elapsed time.

  Referring to FIG. 8, the gate voltage of IGBT Q <b> 1 is 0 volt in the period of time t <b> 0 to t <b> 1, t <b> 2 to t <b> 3, t <b> 4 to t <b> 5, t <b> 6 to t <b> 7, t8 to t <b> 11. IGBTQ1 is in an off state. In the case of FIG. 8, since the flywheel diode D1 is always in an off state, when the IGBT Q1 is normal, the collector voltage in the off section is a high voltage of 600 volts. Since this high voltage is blocked by the diode D7, the anode voltage of the diode D7 is equal to 5 volts which is the voltage of the DC power supply V2.

  On the other hand, in the period from time t1 to t2, t3 to t4, t5 to t6, t7 to t8, t11 to t12 (hereinafter referred to as the on period), the gate voltage of the IGBT Q1 is 15 volts, and the IGBT Q1 is in the on state. is there. When the IGBT Q1 is normal in the ON section of FIG. 8, the collector voltage is equal to the ON voltage of the IGBT Q1. Therefore, the anode voltage of the diode D7 is about 1.2 to 2 volts which is the sum of the ON voltage of the IGBT Q1 and the forward voltage of the diode D7. Further, in the ON section of FIG. 8, the collector current of the IGBT Q1 is gradually increased by PWM control. The magnitude of the collector current is about 10 to 300 amperes.

  From the above, it can be seen that the anode voltage of the diode D7 does not become a value between the reference voltage V4 (0 volt) and the reference voltage V3 (0.6 volt) at the times t0 to t8. Accordingly, at time t0 to t8, the output of the comparator CA1 is at a low level (0 volt), and the output of the comparator CA2 is at a high level (5 volt).

  At time t9 in the middle of times t8 to t11 in the off section, the IGBT Q1 is short-circuited. As a result, the collector voltage of the IGBT Q1 gradually decreases to 0 volts, and the anode voltage of the diode D7 gradually decreases to the forward voltage (0.6 volts). At the same time, a collector current (short circuit current) is generated. The short-circuit current reaches several thousand amperes at maximum.

  At time t10, when the anode voltage of the diode D7 becomes lower than the power supply voltage V3 (0.8 volts) which is the reference voltage, the output voltage of the comparator CA1 is switched from the low level (0 volts) to the high level (5 volts). As a result, the logical product of the output of the comparator CA1 and the output of the comparator CA2 becomes high level.

  At time t11, the gate control computer 20 outputs a high-level control signal for switching the IGBT Q1 from the off state to the on state, and the gate voltage of the IGBT Q1 becomes 15 volts. As a result, the logic circuit LA1 is switched to the high level at time t11. The connection breaker 12 receiving the high level output of the logic circuit LA1 cuts off the current to the load, so that the collector current of the IGBT Q1 finally becomes 0 amperes.

  As described above, according to the failure detection device 1C of the third embodiment, the logical circuit LA1 is used to calculate the logical product of the outputs of the comparators CA1 and CA2 and the control signal of the IGBT Q1, and thus the effect of the second embodiment. In addition, it is possible to detect a short-circuit fault of the IGBT Q1 with higher accuracy. Further, unlike the first embodiment, since the logic operation is performed by the logic circuit LA1 without using the gate control computer 20, the failure detection can be performed at a higher speed than in the first embodiment.

[Embodiment 4]
In the fourth embodiment, a failure detection apparatus that detects that the life of the power element has deteriorated will be described.

  In general, power elements used in power modules are often soldered. Since the solder joint portion deteriorates with the aging of the power module, the on-voltage of the power element increases with the same energization current. Furthermore, since the metal wire that electrically connects the power elements deteriorates with age, it similarly causes an increase in the on-voltage for the same energization current. The failure detection apparatus 1D according to the fourth embodiment determines the life deterioration of the power element by detecting an increase in the on-voltage with respect to the same energization current. Failure detection device 1D can be used alone or in combination with failure detection devices 1A, 1B, and 1C of the first to third embodiments.

  FIG. 9 is a block diagram showing a configuration of failure detection apparatus 1D according to the fourth embodiment of the present invention. In the block diagram of FIG. 9, the common potential VN1 serving as a potential reference is the potential of the emitter (node N1) of the IGBT Q1. Since the gate drive circuit and the failure detection device having the same configuration are provided corresponding to each of the IGBTs Q1 to Q6, hereinafter, the gate drive circuit GD1D and the failure detection device 1D provided in the IGBT Q1 will be described as representatives.

  Referring to FIG. 9, gate drive circuit GD1D includes each component of failure detection apparatus 1D excluding drive circuit UA and resistance element R7. Here, drive circuit UA is the same as in the first to third embodiments, and operates at power supply voltage VCC2 (for example, 15 volts).

  The failure detection device 1D includes a high voltage diode D7, a resistance element R4, a DC power source V2, and a deterioration determination unit 2D. Further, the deterioration determination unit 2D includes open collector output comparators CA3 and CA4, a logic circuit LA2, DC power supplies V5 and V6, a one-shot pulse generation circuit 24, a sample and hold circuit 26, and an element life alarm 28. And resistive elements R6, R7, and R9. Resistor elements R6 and R9 are used as pull-up resistors connected to the output terminals of comparators CA3 and CA4, respectively. One ends of resistance elements R6 and R9 are connected to power supply node VCC1 (for example, 5 volts). The comparators CA3 and CA4 and the logic circuit LA2 operate with the power supply voltage VCC1.

  In FIG. 9, the connection of diode D7, resistance element R4, and DC power supply V2 is the same as in the first to third embodiments. That is, the cathode of the diode D7 is connected to the collector of the IGBT Q1, and one end of the resistance element R4 is connected to the anode of the diode D7. A positive voltage is applied to the other end of the resistance element with respect to the node N1 by the DC power supply V2.

  In order to measure the collector current, IGBTs with current detection electrodes (hereinafter referred to as sense IGBTs) are used as the IGBTs Q1 to Q6 used in the fourth embodiment. The sense IGBT has a current detection electrode (sense electrode) through which a detection current flows according to a collector current (main current). A detection resistor is provided between the sense electrode and the emitter electrode, and a voltage generated at both ends of the detection resistor is detected. In FIG. 9, a resistance element R7 is provided between the sense electrode of IGBTQ1 and the node N1, and a resistance element R8 is provided between the sense electrode of IGBTQ2 and the low voltage node N.

  As described above, failure detection apparatus 1D of the fourth embodiment detects an increase in the on-voltage of the power element under the same energization current. Since the on-voltage of the power element depends on the temperature, it is necessary to consider the temperature dependence of the on-voltage for accurate degradation determination.

  FIG. 10 is a graph showing the temperature dependence of the current-voltage characteristics of the IGBT. In FIG. 10, the horizontal axis indicates the collector-emitter voltage, and the vertical axis indicates the collector current.

  Referring to FIG. 10, there is a cross point CP in the current / voltage characteristic curve of the IGBT for different temperatures. At this time, in the region where the current is smaller than the current value IX at the cross point CP, the on-voltage decreases as the temperature increases. That is, the ON voltage has a negative temperature dependency. On the other hand, in the region where the current is larger than the current value IX, the ON voltage increases as the temperature increases. That is, the ON voltage has a positive temperature dependency. At the current value IX at the boundary between the two regions, there is almost no temperature dependence of the ON voltage. Accordingly, the current value IX at the boundary between the negative voltage region and the positive region where the temperature dependency of the on voltage is a reference current, and the secular change of the on voltage VX at the reference current IX is measured. As a result, the influence of temperature in determining the deterioration of the power element can be minimized.

  Although the current value IX at the cross point CP differs depending on the individual IGBT elements, it is generally 0.8 to 1.2 times the collector current rating. On the other hand, the IGBT is used under the condition that the collector current is 1.5 times the rating at the maximum. Therefore, when a current close to the maximum value flows through the IGBT, the collector current reaches the reference current IX, so that the life determination is performed by the failure detection device 1D. On the other hand, when the collector current does not reach the reference current IX, the life detection cannot be performed by the failure detection device 1D. However, in this case, since the heat generation of the IGBT is also small, it is considered that the possibility that the IGBT falls into a short circuit failure is small.

  Referring to FIG. 9 again, the comparator CA4 is provided for detecting the coincidence between the collector current and the reference current IX. The sense electrode of the sense IGBT Q1 is connected to the non-inverting input terminal of the comparator CA4, and the voltage generated at both ends of the resistance element R7 is input. A DC power supply V6 is connected to the inverting input terminal of the comparator CA4. The voltage value of the DC power supply V6 is set based on the magnitude of the reference current IX so that the comparator CA4 outputs a high level signal when the collector current of the sense IGBT Q1 exceeds the reference current IX.

  The logic circuit LA2 outputs a logical product of the control signal of the IGBT Q1 and the output signal of the comparator CA4. The one-shot pulse generation circuit 24 outputs a one-shot pulse to the sample and hold circuit 26 using the rising edge of the output of the logic circuit LA2 as a trigger.

FIG. 11 is a block diagram showing an example of the configuration of the sample and hold circuit 26 of FIG.
Referring to FIG. 11, sample and hold circuit 26 includes operational amplifiers 34 and 36, switches SW1 and SW2, and a capacitor C2 for holding voltage. The operational amplifiers 34 and 36 are used as voltage followers by directly connecting the output terminal and the inverting input terminal. An operational amplifier 34, a switch SW1, and an operational amplifier 36 are connected in series in this order between the input terminal 30 and the output terminal 32 of the sample and hold circuit 26. A capacitor C2 and a switch SW2 are connected in parallel to each other between the non-inverting input terminal of the operational amplifier 36 and the node N1.

  While the one-shot pulse generation circuit 24 outputs a high level signal, the switch SW1 is closed and the voltage at the input terminal 30 is charged in the capacitor C2. The voltage charged in the capacitor C2 is output from the output terminal 32. After a certain time has elapsed since the one-shot pulse generation circuit 24 outputs a high level signal, the switch SW2 connected to the one-shot pulse generation circuit 24 via the delay circuit 38 is closed. As a result, the voltage charged in the capacitor C2 is discharged through the switch SW2.

  Referring to FIG. 9 again, the sample and hold circuit 26 in the failure detection apparatus 1D holds the anode voltage of the diode D7 when the output of the one-shot pulse generation circuit 24 is at a high level. The held anode voltage is output to the non-inverting input terminal of the comparator CA3.

  A DC power source V5 is connected to the inverting input terminal of the comparator CA3. The comparator CA3 outputs a high level signal when the anode voltage of the diode D7 exceeds the voltage value of the DC power supply V5. As a result, the element life alarm 28 notifies that the life of the IGBT Q1 has deteriorated.

  Here, the voltage value of DC power supply V5 is set to about 1.2 times the ON voltage corresponding to reference current IX when IGBT Q1 is new, for example. Since the on-voltage of the power element at the end of the life increases compared to when it is new, the power element is determined to have a life when the on-voltage with respect to the reference current IX exceeds a predetermined reference voltage V5.

  FIG. 12 is a timing chart schematically showing an example of the voltage / current waveform of each part of FIG. In FIG. 12, the vertical axis indicates the gate voltage of the IGBT Q1, the collector voltage of the IGBT Q1, the anode voltage of the diode D7, the output voltage of the sample and hold circuit 26, the output voltage of the comparator CA4, the output voltage of the logic circuit LA2, and the IGBT Q1. Current of the flywheel diode D1. The horizontal axis is the elapsed time.

  Referring to FIG. 12, the gate voltage of IGBT Q1 is 0 volt in the period of time t0 to t1, t2 to t3, t4 to t5, t6 to t7, and t8 to t9 (hereinafter referred to as the off period). IGBTQ1 is in an off state. In the case of FIG. 12, since the flywheel diode D1 is always in an off state, when the IGBT Q1 is normal, the collector voltage in the off section is a high voltage of 600 volts. Since this high voltage is blocked by the diode D7, the anode voltage of the diode D7 is equal to 5 volts which is the voltage of the DC power supply V2.

  On the other hand, in the period from time t1 to t2, t3 to t4, t5 to t6, t7 to t8, t9 to t11 (hereinafter referred to as the on period), the gate voltage of the IGBT Q1 is 15 volts, and the IGBT Q1 is in the on state. is there. In the ON period, the anode voltage of the diode D7 is the sum of the ON voltage of the IGBT Q1 and the forward voltage of the diode D7.

  In the ON section, the collector current of the IGBT Q1 gradually increases due to the PWM control. Then, at time t10 in the middle of the interval from time t9 to t11, the magnitude of the collector current reaches the reference current IX. At this time, the output of the comparator CA4 and the output of the logic circuit LA2 are at a high level (for example, 5 volts). As a result, the sample and hold circuit 26 holds the anode voltage of the diode D7 at time t10.

  Here, when the IGBT Q1 is new, the anode voltage of the diode D7 corresponding to the reference current IX is VD1 (for example, 1.2 to 2 volts). When the IGBT Q1 is at the end of its life, the anode voltage of the diode D7 is higher than VD1. Suppose that it is large VD2 (for example, 2.8 volts). At this time, if the DC power supply V5 is set to a value between VD1 and VD2 (for example, 2.5 volts), the output of the comparator CA3 becomes high level at time t10, and the life deterioration of the IGBT Q1 is determined. The

  As described above, according to the failure detection device 1D of the fourth embodiment, the secular change of the on-voltage is detected on the basis of the current value at the cross point CP, so that the power element can be used under the condition that the influence of temperature is minimized. It is possible to determine the deterioration of the life. If the life of the power element used in the power module is found, an exchange signal can be issued before the power module breaks at the life limit, and the power module can be prevented from being broken.

  In the first to fourth embodiments described above, the IGBT Q1 failure detection device 1A of the inverter circuit 11 has been described. However, the same configuration is used for the failure determination devices of the IGBTs Q2 to Q6 except for the common potential setting method. Thus, the same effect can be obtained. Here, in the failure detection device for IGBTQ3 and IGBTQ5, the potentials of nodes N2 and N3 are used as the common potential instead of the common potential VN1 of FIG. For the failure detection devices of IGBTQ2, IGBTQ4, and IGBTQ6, the potential of the ground node GND is used as the common potential instead of the common potential VN1 of FIG.

  The embodiment disclosed this time must be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  Q1-Q6 IGBT, 1A-1D Failure detection device, 2A-2C Failure determination unit, 2D Degradation determination unit, 11 Inverter circuit, 20 Gate control computer, 26 Sample hold circuit, CA1-CA4 comparator, D1-D6 flywheel Diode, D7 diode, IX reference current, LA1, LA2 logic circuit, PC photocoupler, R4 to R9 resistance element, V1 to V6 DC power supply.

Claims (2)

A power element failure detection apparatus in a power circuit including a switching element in which a main current flowing from a first main electrode to a second main electrode changes according to a control signal,
A first rectifying element having a cathode connected to the first main electrode;
A first resistance element having one end connected to the anode of the first rectifying element and the other end applied with a positive voltage with respect to the second main electrode;
A deterioration determining unit for determining deterioration of the switching element;
The degradation determination unit detects and detects a monitoring voltage between the anode of the first rectifying element and the second main electrode when the magnitude of the main current matches a predetermined reference current. Determining whether the monitoring voltage exceeds a predetermined reference voltage;
The reference current is set such that the magnitude of the main current is a boundary between the first region and the second region,
The first region is a region in which the voltage between the first and second main electrodes of the switching element has a negative temperature dependency,
The power device failure detection apparatus, wherein the second region is a region in which a voltage between the first and second main electrodes of the switching element has a positive temperature dependency. The switching element includes a sense electrode for detecting the main current,
The deterioration determination unit
A second resistance element connected between the sense electrode and the second main electrode for measuring the main current;
A first comparator for comparing a voltage generated at both ends of the second resistance element with a voltage corresponding to the reference current;
A sample and hold circuit for holding the monitoring voltage when the main current matches the reference current;
2. The power element failure detection device according to claim 1, further comprising a second comparator that compares the monitoring voltage held by the sample and hold circuit with the reference voltage.

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