JP4797074B2 - Vector control device for permanent magnet motor, vector control system for permanent magnet motor, and screw compressor - Google Patents

Description

  The present invention relates to a vector control device, a control system, and a screw compressor for a permanent magnet motor without a position sensor, and more particularly to a technique for realizing a highly responsive control characteristic.

  In order to reduce the size of a motor control system, a technique for controlling a motor by omitting a position sensor that detects a motor rotation position (rotation angle) is required. This position sensorless vector control system motor drive technology uses vector command output voltage command value, current detection value, and speed command value to determine the motor rotation angle reference axis (motor flux axis) and control axis. A technique for performing an estimation calculation for estimating an axis error is disclosed in Patent Document 1. Patent Document 1 describes a control method in which the axial error command value is basically set to “zero”.

JP 2001-251889 A

The method described in Patent Document 1 can realize high response and highly stable control for a motor having a small electrical time constant (product of inductance value and resistance value), that is, a small to medium-sized motor.
However, when applied to a motor having a relatively large electric time constant, the control response of the frequency estimation calculation is limited, so that a large axis error may occur during acceleration / deceleration operation.
As a result, there is a problem that excessive current flows through the motor and the operation efficiency deteriorates.

  The present invention has been made in order to solve the above-described problem, and even in the case of a permanent magnet motor having a large electric time constant, a vector control device for a permanent magnet motor capable of suppressing an axis error during acceleration / deceleration operation, It is an object to provide a control system and a screw compressor.

  In order to achieve the above object, the vector control apparatus of the present invention uses a change (differentiation) of a detected current value (d-axis current detected value or q-axis current detected value) flowing in the permanent magnet motor to perform a vector control reference. A reference axis correction unit that corrects the control axis (for example, phase command value).

  According to the present invention, a shaft error during acceleration / deceleration operation can be suppressed even in the case of a permanent magnet motor having a large electric time constant.

It is a block diagram of the control system of the permanent magnet motor which is 1st Embodiment of this invention. It is a characteristic view which shows the driving | running characteristic of control response frequency _FPLL = 5Hz. It is a characteristic view which shows the driving | running characteristic of control response frequency _FPLL = 16Hz. It is a gain-frequency characteristic diagram and a phase-frequency characteristic diagram. It is a Lous table. It is a block diagram of a stabilization compensation part. It is a gain-frequency characteristic diagram and a phase-frequency characteristic diagram. It is a control characteristic figure which shows a driving | running characteristic ( _FPLL = 16Hz). It is a block diagram of the control system of the permanent magnet motor which is 2nd Embodiment of this invention. It is a block diagram of another stabilization compensation part. It is a block diagram of the control system of the permanent magnet motor which is 3rd Embodiment of this invention. It is a block diagram of the control system of the permanent magnet motor which is 4th Embodiment of this invention. It is a block diagram of the control system of the permanent magnet motor which is 5th Embodiment of this invention. It is a block diagram of the control system of the permanent magnet motor which is 6th Embodiment of this invention. It is a block diagram of the screw compressor provided with the control system of the permanent magnet motor of this invention.

(First embodiment)
FIG. 1 is a configuration diagram of a control system for a permanent magnet motor according to the first embodiment of the present invention.
The control system 200 includes a permanent magnet motor 1, a power converter 2 that drives the permanent magnet motor 1, a DC power source 21 that supplies DC power to the power converter 2, and a control signal Vu ^* that controls the power converter 2 ^. , Vv ^* , Vw ^* , and a current detector 3 for detecting AC currents iu, iv, iw flowing through the permanent magnet motor 1.
Here, the permanent magnet motor 1 generates a motor torque obtained by synthesizing the torque component due to the magnetic flux of the permanent magnet and the reluctance torque component due to the direction difference between the inductances of the armature windings, and rotates at the motor rotational frequency ωr. The power converter 2 generates a drive voltage that is proportional to the three-phase control signals Vu ^* , Vv ^* , and Vw ^* and that is PWM (Pulse Width Modulation) controlled, and supplies drive power to the permanent magnet motor 1.

  The control device 100 includes a CPU (Central Processing Unit), a ROM (Read Only Memory), a RAM (Random Access Memory), and the like. When the CPU executes a program, the vector control unit 150, the reference axis correction unit 155, Realize the function. The reference axis correction unit 155 includes the functions of the phase estimation unit 7, the stabilization compensation unit 8, and the addition unit 9.

The phase estimating unit 7 calculates the phase estimated value θdc by integrating the frequency estimated value ω1c using the equation (1), and estimates the reference axis (the magnetic flux axis of the motor) of the permanent magnet motor 1.

  The stabilization compensator 8 performs a differential operation on the detected d-axis current value Idc and outputs a stabilization signal (a stabilization compensation value) Δθd as a differential operation value. The adding unit 9 adds the phase estimation value θdc and the stabilization signal Δθd to calculate a new phase estimation value (other phase estimation value) Δθdc_a, and outputs the calculation result to the vector control unit 150.

The vector control unit 150 includes a coordinate conversion unit 4, an axis error estimation unit 5, a frequency estimation unit 6, a low-pass filter 10, a vector control calculation unit 11, a coordinate conversion unit 12, and an addition unit 22. Based on the frequency command value ωr ^* , the current detection values i _uc , i _vc , i _wc detected by the current detector 3, and other phase estimation values θdc_a, vector-controlled control signals Vu ^* , Vv ^* , Vw ^{* And} the frequency estimation value ω1c are generated.

The vector control calculation unit 11 sets the d-axis current command value Id ^* set to “zero”, the first-order lag value Iqctd of the q-axis current detection value Iqc, the frequency command value ωr ^* , and the electric constants of the permanent magnet motor 1 ( The d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* are output based on a preset value). That is, the vector control calculation unit 11 sets the d-axis current command value Id ^* , the first-order lag value Iqctd of the q-axis current detection value Iqc, the frequency command value ωr ^* , and the set value of the motor constant (R: resistance value, Ld: d The d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* shown in Equation (2) are calculated using the shaft inductance value, Lq: q-axis inductance value, and Ke: induced voltage coefficient.

The coordinate conversion unit 12 calculates the three-phase control signals Vu ^* , Vv ^* , Vw ^* from the d-axis voltage command value Vdc ^* , the q-axis voltage command value Vqc ^*, and the new phase estimation value θdc_a, and converts the power. To the device 2.

  The coordinate conversion unit 4 calculates a d-axis current detection value Idc and a q-axis current detection value Iqc from the current detection values iuc, ivc, iwc of the three-phase alternating currents iu, iv, iw and the phase estimation value θdc. . The low-pass filter 10 outputs a first-order lag value Iqctd of the q-axis current detection value Iqc, and this output signal is fed back to the vector control calculation unit 11.

Based on the d-axis voltage command value Vdc ^* , the q-axis voltage command value Vqc ^* , the frequency estimation value ω1c, the d-axis current detection value Idc, the q-axis current detection value Iqc, and the motor constant, An estimation calculation of an axis error Δθ which is a deviation between the phase estimation value θdc and the phase value θd of the rotation angle (electrical angle) of the permanent magnet motor 1 is performed, and an estimation value Δθc is output. That is, the axis error estimation unit 5 sets the d-axis voltage command value Vdc ^* , the q-axis voltage command value Vqc ^* , the d-axis current detection value Idc, the q-axis current detection value Iqc, the frequency estimation value ω1c, and the motor constant. Using the value, an estimation calculation of an axis error value Δθdc (= θdc−θd), which is a deviation between the phase estimation value θdc and the motor phase value θd, is performed according to Equation (3).

ここで、Ｋｐ は、比例ゲインであり、（５）式に示すように設定されている。

ここで、 Ｆ_ＰＬＬ は、周波数推定部６に設定する制御応答周波数［Ｈｚ］である。 The adder 22 calculates a deviation between the axial error command value Δθc ^* set to “zero” and the estimated axial error value Δθc.
The frequency estimation unit 6 performs a proportional calculation or (proportional + integration) calculation on the calculation result of the addition unit 22, adds the calculation result to the frequency command value ωr ^* , and outputs a frequency estimation value ω1c. The phase estimation value ω1c is fed back to the axis error estimation unit 5. Further, the frequency estimation unit 6 calculates the frequency estimated value ω1c by the equation (4) so that the estimated value Δθc of the axis error becomes “zero”.

Here, Kp is a proportional gain, and is set as shown in equation (5).

Here, F _PLL is a control response frequency [Hz] set in the frequency estimation unit 6.

Next, the “effect of correcting the phase estimation value (control axis) for adding the stabilization signal Δθd to the phase estimation value θdc”, which is a feature of the present embodiment, will be described. First, the control characteristics of “when no stabilization compensation Δθd is added”, which is a conventional method, will be described.
FIG. 2 is a characteristic diagram showing “operating characteristics when the control response frequency _FPLL is low (F _PLL = 5 Hz)” set in the frequency estimator 6, and FIG. 2A shows the motor rotation frequency ωr [rad / 2B shows the time characteristic of the primary current I1 [A], and FIG. 2C shows the time characteristic of the axis error estimated value Δθc [deg].

FIG. 2 (a) shows the A section in which the motor rotation frequency ωr accelerates from the lowest speed (point a) to the highest speed (point b), and rotates at the highest speed for a predetermined time (about 2.2 seconds), and further reaches the highest speed. It is distinguished from the B section that linearly decelerates from the speed (point b) to the minimum speed (point a).
FIG. 2 shows that in section A (FIG. 2A), the estimated value Δθc of the axis error Δθ is generated by 30 degrees when acceleration is started and −30 degrees when acceleration is stopped (FIG. 2C). 1 indicates that the primary current I1 is 17.4 A (FIG. 2B). A predetermined primary current I1 flows during motor acceleration, and a negative regenerative current flows during motor deceleration.

On the other hand, FIG. 3 is a characteristic diagram showing “operating characteristics when the control response frequency F _PLL is increased (F _PLL = 16 Hz)”.
FIG. 3 shows that in the section A (FIG. 3A), the estimated value Δθc of the axis error Δθ is as small as 20 deg (−20 deg) (FIG. 3C). As a result, the primary current I1 of the permanent magnet motor 1 15A (FIG. 3 (b)). However, in the section B, the estimated value Δθc of the axis error is divergent (FIG. 3 (c)). There is a problem that leads to “operation stop due to overcurrent”.

2 and 3, when the control response frequency F _PLL set in the frequency estimation unit 6 is as low as F _PLL = 5 Hz, the axis error estimation unit 5 generates a large axis error Δθ. At the same time, the primary current I1 of the permanent magnet motor 1 increases, and the motor efficiency decreases.
On the other hand, when the control response frequency _FPLL is high, the shaft error Δθ is suppressed and the primary current I1 of the permanent magnet motor 1 is decreasing, but the control may become unstable.

ここで、ｄ軸電圧値Ｖｄ、及びｑ軸電圧値Ｖｑは、（７）式で示すことができる。

（７）式より、ｄ軸電圧値Ｖｄは、ｑ軸電圧指令値Ｖｑｃ^＊の情報が含まれていることが分かる。このｄ軸電圧値Ｖｄの変動により、ｄ軸電流Ｉｄと、ｑ軸電圧値Ｖｑと、ｑ軸電流Ｉｑとが変動する。軸誤差推定部５では、このｄ軸電流検出値Ｉｄｃ、ｑ軸電流検出値Ｉｑｃを用いた演算を行っているため、軸誤差の推定値Δθｃに関係する一巡の不安定ループが発生してしまう。
ここで、「軸誤差の指令値Δθｃ^＊」から「軸誤差の推定値Δθｃ」までの、一巡伝達関数Ｇ_Δθｃ（ｓ）は、Ａｖ（ωｒ）：モータ回転周波数ωｒの平均値として、

である。 Next, the cause of the “axis error estimated value Δθc instability” that occurs when the control response frequency F _PLL = 16 Hz is set high.
When there is an axis error Δθ (= θdc−θd) that is a deviation between the phase estimation value θdc and the phase value θd of the permanent magnet motor 1, conversion from the control axis (dc−qc) to the motor axis (dq) The matrix is given by equation (6).

Here, the d-axis voltage value Vd and the q-axis voltage value Vq can be expressed by Equation (7).

From the equation (7), it can be seen that the d-axis voltage value Vd includes information on the q-axis voltage command value Vqc ^* . Due to the variation of the d-axis voltage value Vd, the d-axis current Id, the q-axis voltage value Vq, and the q-axis current Iq vary. Since the axis error estimation unit 5 performs calculations using the d-axis current detection value Idc and the q-axis current detection value Iqc, a round of unstable loops related to the axis error estimation value Δθc occurs. .
Here, the round transfer function G _Δθc (s) from the “axis error command value Δθc ^* ” to the “axis error estimated value Δθc” is Av (ωr): an average value of the motor rotation frequency ωr,

It is.

FIG. 4 shows a control response frequency F _PLL (==) set in the characteristics of FIGS. 2 and 3, with the motor rotation frequency ωr being the highest frequency (point b in FIG. 2, ωr = 2π · 400 rad / s) in the equation (8). 5H, 16Hz) is a gain-frequency characteristic diagram and a phase-frequency characteristic diagram when substituting (8) into equation (8). Here, the vertical axis of these characteristic diagrams is gain [dB] and phase [deg], and the horizontal axis is frequency [Hz].

In FIG. 4, “dashed line” indicates a characteristic using F _PLL = 5 Hz, and “solid line” indicates a characteristic using F _PLL = 16 Hz. According to the characteristic diagram of FIG. 4, when the phase characteristic is -180 [deg] at the point c, the characteristic (broken line) of the control response frequency F _PLL = 5 Hz has a gain of 0 [dB] (= 1) or less. Therefore, it can be seen that it is “stable”, and the characteristic (solid line) of the control response frequency F _PLL = 16 Hz is “unstable” because the gain is 0 [dB] or more. That is, when increasing the control response frequency F _PLL, the estimated value of the axis error Δθc becomes unstable.

ここで、（１０）式のすべての係数が同符号である「必要条件」を満たしているので、ラウス表を作成する。作成したラウス表を図５に示す。図５の第１列目がラウス数列であり、安定であるためには、（１１）式が成立する必要がある。

（１１）式において、比例ゲインＫ_ＰＬＬについて整理すると、

となる。ここで、（１３）式の関係が成立する高速域の範囲では、

となる。すると、（１２）式を（１４）式に示すように考えることができる。

制御応答周波数Ｆ_ＰＬＬについて整理すると、

となる。つまり、高速域において、安定に運転するためには、（１５）式の関係を満たした制御応答周波数Ｆ_ＰＬＬを設定する必要がある。大容量のモータになると、抵抗値Ｒは小さく、ｄ軸インダクタンス値Ｌｄ、及びｑ軸インダクタンス値Ｌｑは大きくなる傾向にあり、制御応答周波数Ｆ_ＰＬＬがモータ定数により制限を受けてしまう問題があった。 Here, when the stability condition of the estimated value Δθc of the axis error is obtained, the characteristic equation of the _cyclic transfer function G _Δθc (s) of the equation (8) becomes the equation (9), and the equation of the equation (10) may be considered. .

Here, since all the coefficients of the equation (10) satisfy the “requirement” having the same sign, a Rous table is created. The created Rouss table is shown in FIG. The first column in FIG. 5 is the Rouss number sequence, and in order to be stable, equation (11) needs to be satisfied.

In the equation (11), when the proportional gain K _PLL is arranged,

It becomes. Here, in the range of the high speed region where the relationship of equation (13) holds,

It becomes. Then, equation (12) can be considered as shown in equation (14).

The control response frequency F _PLL can be summarized as follows:

It becomes. That is, in order to operate stably in the high speed range, it is necessary to set the control response frequency _FPLL that satisfies the relationship of the equation (15). When the motor has a large capacity, the resistance value R is small, the d-axis inductance value Ld and the q-axis inductance value Lq tend to be large, and the control response frequency _FPLL is limited by the motor constant. .

ここで、
Ｋａ：微分ゲイン、Ｔａ：一次遅れ時定数
である。 Therefore, the “stabilization compensator 8 (FIG. 1)” that is a characteristic configuration of the present embodiment will be described. The stabilization compensator 8 shown in FIG. 6 calculates a differential value of the d-axis current detection value Idc by the equation (16), and outputs a stabilization signal Δθd as a calculation result. Then, the adding unit 9 corrects the phase estimation value θdc using the stabilization signal Δθd shown in the equation (16).

here,
Ka: differential gain, Ta: first-order lag time constant.

ここに、
ξｃ：任意に設定する二次系の減衰係数である。
ここで、安定化補償部８を入れた場合における「軸誤差の指令値Δθｃ^＊」から「軸誤差の推定値Δθｃ」までの、一巡伝達関数ＧΔ_{θｃｎ＿ｄ}（ｓ）を求めると、

となる。（１３）式の関係が成立する高速域の範囲では、
（１８）式を、（１９）式に示すＧΔ_{θｃｎ＿ｄ＿ａ}（ｓ）に近似することが可能となる。

ここで、Ｋｅ^＊＝Ｋｅ、Ｒ^＊＝Ｒ、Ｌｄ^＊＝ Ｌｄ、Ｌｑ^＊＝ Ｌｑと仮定すると、（２０）式に示すＧΔ_{θｃｎ＿ｄ＿ｂ}（ｓ）を得る。

（２０）式中の二次遅れ要素は、（２１）式で示すＧ（ｓ）となる。

二次遅れ要素の一般系は、（２２）式で示すＧａ（ｓ）ある。

ここで、
ωｎ：固有周波数、ξ：減衰係数
である。（２２）式に、（２１）式を当てはめると、固有周波数ωｎと減衰係数ζとは（２３）式となる。

（２３）式において、モータ自体の減衰係数ξ_０は、

となる。（２４）式より、高速域（ωｒ：大）では減衰係数ξ_０が小さくなることが分かる。これが、軸誤差の推定値Δθｃが不安定となる原因である。 When the differential gain Ka and the first-order lag time constant Ta are expressed by the equation (17),

here,
ξc: A damping coefficient of a secondary system set arbitrarily.
Here, when the one-round transfer function _{GΔθcn_d} (s) from “axis error command value Δθc ^* ” to “axis error estimated value Δθc” when the stabilization compensator 8 is inserted,

It becomes. In the range of the high speed range where the relationship of equation (13) holds,
The equation (18) can be approximated to _{GΔθcn_d_a} (s) shown in the equation (19).

Here, assuming that Ke ^* = Ke, R ^* = R, Ld ^* = Ld, and Lq ^* = Lq, _{GΔθcn_d_b} (s) shown in Equation (20) is obtained.

The second-order lag element in the equation (20) is G (s) shown in the equation (21).

A general system of second-order lag elements is Ga (s) represented by the equation (22).

here,
ωn: natural frequency, ξ: damping coefficient. When the equation (21) is applied to the equation (22), the natural frequency ωn and the attenuation coefficient ζ become the equation (23).

In the equation (23), the damping coefficient ξ ₀ of the motor itself is

It becomes. From the equation (24), it can be seen that the damping coefficient ξ ₀ is small in the high speed range (ωr: large). This is the reason why the estimated value Δθc of the axis error becomes unstable.

（２５）式に含まれる二次遅れ要素のゲインの大きさ｜Ｇ_＿ｂ（ｓ）｜は、

となる。 Therefore, by introducing a “new damping coefficient ξc” that is irrelevant to the motor rotational frequency ωr on the control side (ξc >> ξ ₀ setting), the equation (20) is expressed as _{GΔθcn_d_c} (s) shown in the equation (25). To change.

The magnitude of the gain of the second-order lag element included in the equation (25) | _{G_b} (s) |

It becomes.

減衰係数ξｃは０を超えた値で設定すれば、減衰の効果が現れる。
ここでは、ξｃ＝０．５で設定したが、ξｃ＞０ で設定すればよい。
（１８）式に示す一巡伝達関数において、ωｒ＝２π・４００［ｒａｄ／ｓ］、Ｆ_ＰＬＬ＝１６［Ｈｚ］、ξｃ＝０．５を代入した周波数特性を図７に示す。 Here, if ξc = 0.5, the gain component of the motor rotation frequency ωr can be set to “1” as shown in the equation (27).

If the attenuation coefficient ξc is set to a value exceeding 0, the effect of attenuation appears.
Here, ξc = 0.5, but ξc> 0 may be set.
FIG. 7 shows frequency characteristics obtained by substituting ωr = 2π · 400 [rad / s], F _PLL = 16 [Hz], and ξc = 0.5 in the round transfer function shown in the equation (18).

When the stabilization compensator 8 (FIG. 6) of the present embodiment is used, it can be seen that the gain characteristic of the point c component is sufficiently lowered.
FIG. 8 is a control characteristic diagram when the same value as F _PLL = 16 Hz used in FIG. 3 is set. Conventionally, it can be seen that in the unstable B section as shown in FIG. 3, the configuration of the present embodiment shows that the estimated value Δθc of the axis error operates stably without divergence. The maximum value of the primary current I1 is about 15 [A] (see FIG. 8B), and the maximum value of the axis error estimated value Δθc is 20 [deg] (see FIG. 8C).

  According to this embodiment, from a motor with a small electrical time constant (ratio of resistance value and inductance value) to a motor with a large electrical time constant, the shaft error during acceleration / deceleration operation is minimized, and high efficiency operation is performed. Can be realized.

ここで、Ｋｂ を（２９）式に示すようにすると、

となる。ここで、
ξｃ：任意に設定する二次系の減衰係数である。 (Second Embodiment)
FIG. 9 is a configuration diagram of a control system according to the second embodiment of the present invention.
In the control device 100 for the permanent magnet motor 1 according to the first embodiment, the phase command value θdc is corrected with the stabilization signal Δθd that is the d-axis differential calculation value. However, the control device 100a according to the second embodiment The phase command value θdc is modified using the differential operation value Δθq.
9, the control system 200a includes a control device 100a, a permanent magnet motor 1, and a power converter 2, and the control device 100a includes a vector control unit 150a and a reference axis correction unit 155a.
1 to 7, 10 to 12, and 22 are the same as the components shown in FIG. 1.
The stabilization compensator 8a shown in FIGS. 9 and 10 corrects the phase estimation value θdc using the differential value of the q-axis current detection value Iqc. Further, the stabilization signal Δθq is obtained by the calculation shown in the equation (28).

Here, when Kb is expressed by equation (29),

It becomes. here,
ξc: A damping coefficient of a secondary system set arbitrarily.

（２０）式に示す一巡伝達関数Ｇ_{Δθｃｎ＿ｄ＿ｂ}（ｓ）と同一になる。
つまり、第２実施形態の構成を用いても、第１実施形態と等価な効果を奏することができる。 The adding unit 9a adds the phase estimation value θdc and the stabilization signal Δθq, calculates a new phase estimation value θdc_b, and outputs the calculation result to the coordinate conversion unit 12 as a corrected axis.
Here, when the round transfer function G _{Δθcn_q} (s) from the “axis error command value Δθc ^* ” to the “axis error estimated value Δθc” when the stabilization compensator 8 a is inserted, the equation (30) is obtained. And

This is the same as the one-round transfer function G _{Δθcn_d_b} (s) shown in the equation (20).
That is, even if the configuration of the second embodiment is used, an effect equivalent to that of the first embodiment can be obtained.

(Third embodiment)
FIG. 11 is a configuration diagram of a control system according to the third embodiment of the present invention.
First embodiment, and in the second embodiment, regardless of the magnitude of the control response frequency F _PLL to be set to the frequency estimation unit 6 has been modified to phase estimate θdc by stability compensation, this embodiment, the control This is a method of performing on / off operation of stabilization compensation according to the magnitude of the response frequency F _PLL .
In FIG. 11, the control system 200b includes a control device 100b, a permanent magnet motor 1, and a power converter 2, and the control device 100b includes a vector control unit 150b and a reference axis correction unit 155b. The vector control unit 150b includes an F _PLL 61 and a frequency estimation unit 6a, and the reference axis estimation unit 155b includes a stabilization compensation unit 8b and an addition unit 9b. In FIG. 11, the constituent elements denoted by reference numerals 1 to 5, 7, 10 to 12, and 22 are the same as the constituent elements in FIG.
The F _PLL 61 outputs a control response frequency F _{PLL — set} to be set in the frequency estimation unit.

Frequency estimation unit 6a, using the control response frequency _F PLL _{_ The set,} is set to indicate the proportional gain Kp_set to (31) below.

Stability compensation unit 8b, only if the control response frequency F _PLL _ _{The set} is (15) is not satisfied expression conditions, and is configured to output a stabilized signal .DELTA..theta.d. That is, in the present embodiment, the phase command value θdc is corrected only in the region where the estimated value Δθc of the axis error is unstable. Even if this embodiment is used, an effect equivalent to that of the first embodiment can be obtained.

(Fourth embodiment)
FIG. 12 is a configuration diagram of a control system for the permanent magnet motor 1 according to the fourth embodiment of the present invention.
In the first embodiment and the second embodiment, the phase command value θdc is corrected by the stabilization compensation regardless of the magnitude of the frequency estimation value ω1c. However, in this embodiment, the phase estimation value ω1c is stabilized by the magnitude of the frequency estimation value ω1c. ON / OFF operation of compensation is performed.
The control system 200c includes a control device 100c, a permanent magnet motor 1, and a power converter 2, and the control device 100c includes a vector control unit 150c and a reference axis correction unit 155c.
In the figure, the reference axis correcting unit 155c includes a stabilization compensator 8c, but the components indicated by reference numerals 1 to 7, 9 to 12, and 22 are the same as those in FIG.

となる。
安定化補償部８ｃは、周波数推定値ω１ｃが（３２）式の条件を満たす場合のみ、安定化信号Δθｄを出力する。
つまり、本実施形態では高速領域のみ、安定化信号Δθｄを用いて、位相指令値θｄｃの修正を行っている。
本実施形態を用いても、第１実施形態と等価な効果を奏することができる。 The range of the high speed range is a range that satisfies the equation (13). If the estimated frequency value ω1c is used instead of the motor rotation frequency ωr,

It becomes.
The stabilization compensator 8c outputs the stabilization signal Δθd only when the frequency estimation value ω1c satisfies the condition of the equation (32).
That is, in this embodiment, the phase command value θdc is corrected using the stabilization signal Δθd only in the high-speed region.
Even if this embodiment is used, an effect equivalent to that of the first embodiment can be obtained.

(Fifth embodiment)
FIG. 13 is a configuration diagram of a permanent magnet motor control system according to the fifth embodiment of the present invention.
The control system 200d includes a control device 100d, a permanent magnet motor 1, and a power converter 2, and the control device 100d includes a vector control unit 150d.
The vector control unit 150d includes a d-axis current control calculation unit 13, a q-axis current control calculation unit 14, and addition units 23 and 24. The control device 100d is connected to the d-axis current control calculation unit 13 via the addition unit 23. A connected speed control calculation unit 15 is provided. In FIG. 13, the components indicated by reference numerals 1 to 5, 7 to 9, 12, and 22 are the same as those in FIG.

The speed control calculation unit 15 receives the deviation between the frequency command value ωr ^* and the frequency estimation value ω1c, and calculates the q-axis current command value Iq ^* . The frequency estimation unit 6 calculates (proportional + integral) the deviation between the axial error command value Δθc ^* set to “zero” and the axial error estimation value Δθc, and outputs the estimated frequency value ω1c. The speed control calculation unit 15 outputs the first q-axis current command value Iq ^* from the deviation between the frequency command value ωr ^* and the frequency estimated value ω1c. The d-axis current control calculation unit 13 outputs the second d-axis current command value Id ^** from the deviation between the first d-axis current command value Id ^* set to “zero” and the d-axis current detection value Idc. The q-axis current control calculation unit 14 outputs a second q-axis current command value Iq ^** from the deviation between the q-axis current command value Iq ^* and the q-axis current detection value Iqc.

The vector control calculation unit 11a uses the electrical constant of the permanent magnet motor 1, the second d-axis current command value Id ^** , the second q-axis current command value Iq ^** , and the frequency estimation value ω1c according to the equation (33), The d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* are output.

Thus, also in the vector control device provided with the d-axis current control calculation unit 13 and the q-axis current control calculation unit 14, the same effect as that of the first embodiment can be obtained.
Further, in the present embodiment, the method of FIG. 1 is used in which the phase command value θdc is corrected with the d-axis differential calculation value Δθd, but in the method of correcting the phase command value θdc with the q-axis differential calculation value Δθq. Similar effects can be obtained.

(Sixth embodiment)
FIG. 14 is a configuration diagram of a control system for a permanent magnet motor according to the sixth embodiment of the present invention.
In the present embodiment, the present invention is applied to a control device that performs field-weakening control and includes a d-axis current control calculation unit and a q-axis current control calculation unit. A control system 200e of the sixth embodiment includes a permanent magnet motor 1, a power converter 2, a position detector 25, and a control device 100e, and the control device 100e includes a vector control unit 150e.
The vector control unit 150e includes a frequency calculation unit 6b, a voltage vector calculation unit 11b, a d-axis current control calculation unit 13a, a q-axis current control calculation unit 14a, a q-axis current deviation switching unit 18, and a d-axis current deviation. A switching unit 17, a phase error command calculation unit 30, and an output voltage limit detection unit 35 are provided. In the figure, the constituent elements denoted by reference numerals 1 to 4, 9, 12, and 22 are the same as those in FIG.

加算部９ｂは、位置検出値θｄと安定化信号Δθｄとを加算して、新たな位相推定値θｄｃ＿ｅを座標変換部１２に出力する。 The position detector 25 includes a resolver, an encoder, a Hall IC, a Hall element, and the like, detects the rotational position (rotational angle) of the permanent magnet motor 1, and outputs a position detection value θd.
The frequency calculation unit 6b uses the position detection value θd to obtain the frequency calculation value ω1 of the primary frequency according to the equation (34).

The adder 9 b adds the position detection value θd and the stabilization signal Δθd, and outputs a new phase estimation value θdc_e to the coordinate converter 12.

The voltage vector calculation unit 11b is based on the electric constant of the permanent magnet motor 1, the second current command value Id ^** , the second q-axis current command value Iq ^** , the frequency calculation value ω1, and the phase difference command value Δθc ^*. The voltage command value Vdc ^** and the voltage command value Vqc ^** are calculated. The d-axis current control calculation unit 13a outputs the second d-axis current command value Id ^** from the output value ΔId1 of the d-axis current deviation switching unit 17. The q-axis current control calculation unit 14a outputs the second q-axis current command value Iq ^** from the output value ΔIq2 of the q-axis current deviation switching unit 18.

The phase error command calculation unit 30 outputs a phase error command value Δθc ^* from the output value ΔIq1 of the q-axis current deviation switching unit 18. The output voltage limit detection unit 35 calculates the output voltage value V1 ^* of the power converter 2 from the d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* ,
When the voltage value V1 ^* is smaller than the voltage limit value V1 ^* max,
Set the output voltage limit flag V1 ^* lmt_flg to “0”,
When the voltage value V1 ^* reaches V1 ^* max, the output voltage limit flag V1 ^* lmt_flg is set to “1”.

The d-axis current deviation switching unit 17 outputs a deviation ΔId1 or “zero” between the d-axis current command value Id ^* and the d-axis current detection value Idc using the output voltage limit flag V1 ^* lmt_flg. The q-axis current deviation switching unit 18 uses the output voltage limit flag V1 ^* lmt_flg to set the deviation between the q-axis current command value Iq ^* and the current detection value Iqc, or “zero”, to either ΔIq1, ΔIq2, respectively. Output as.

Here, the operation of the phase error command calculation unit 30 will be described.
In the phase error command calculation unit 30, when the output voltage limit flag V1 ^* lmt_flg is “1”, a deviation (= ΔIq) between the first q-axis current command value Iq ^* and the q-axis current detection value Iqc is calculated (proportional + integration). The calculated value is calculated as a command value Δθc ^* for the axis error.
At this time, in the vector calculation of the voltage vector calculation unit 11b, the input signals ΔId1, ΔIq2 of the d-axis current control calculation unit 13a and the q-axis current control calculation unit 14a are both “zero”, and the output values Id ^** , Iq The calculation of ^** is not updated and the previous value is retained.
It can be seen that the d-axis voltage command value Vdc ^* and the q-axis voltage command value Vqc ^* shown in the equation (31) are constant values.
Next, a new d-axis voltage command value Vdc ^** , q-axis voltage is obtained from the axis error command value Δθc ^* , the d-axis voltage command value Vdc ^* , and the q-axis voltage command value Vqc ^* as shown in the equation (35). Command value Vqc ^** is calculated.

That is, in the region where the output voltage value V1 ^* is limited, the axis error Δθ, which is a phase error between the control axis and the magnetic flux axis of the motor, so that the q-axis current command value Iq ^* matches the current detection value Iqc. The output voltage value V1 ^* is controlled via (= θdc−θ). Then, field weakening control can be realized without generating the d-axis current command value Id ^* (zero).
In the control device 100e that performs such field weakening control and includes the d-axis current control calculation unit 13a and the q-axis current control calculation unit 14a, the control response frequency set in the phase error command calculation unit 30 is set high. be able to.
In the present embodiment, a method of correcting the phase command value θdc with the d-axis differential calculation value Δθd is used. However, the same effect can be obtained with a method of correcting the phase command value θdc with the q-axis differential calculation value Δθq. Is obtained.

(Seventh embodiment)
FIG. 15 is a configuration diagram of the screw compressor of the seventh embodiment.
The screw compressor 300 includes a control device 150, a power converter 2, and a motor & compressor 19, and is configured to be controlled using the operation panel 20. Note that the components indicated by reference numerals 2 to 12 in FIG. 1 are implemented by software and hardware circuits.
If the present invention is applied to a screw compressor, it is possible to realize a control characteristic with high response and high accuracy.

また、第１実施形態乃至第３実施形態では、高価な電流検出器３で検出した３相の交流電流ｉｕ、ｉｖ、ｉｗを検出する方式であったが、電力変換器２の過電流検出用に取り付けているワンシャント抵抗器に流れる直流電流から、３相のモータ電流ｉｕ＾、ｉｖ＾、ｉｗ＾を再現し、この再現電流値を用いる「低コストなシステム」にも対応することができる。 So far
In the first embodiment and the second embodiment, the d-axis current control Id ^* (= 0) and the primary value of the q-axis current detection value Iqc are provided without the d-axis current control calculation unit and the q-axis current control calculation unit. Vector control calculation is performed using the delay signal Iqctd, the speed command value ωr ^* and the electrical constant of the permanent magnet motor 1,
In the fifth embodiment, the second d-axis current command value Id ^{* is} calculated from the first d-axis current command value Id ^* , the first q-axis current command value Iq ^* , the d-axis current detection value Idc, and the q-axis current detection value Iqc ^{. * The} second q-axis current command value Iq ^** was created and the vector control calculation was performed using this second current command value.
From the first d-axis current command value Id ^* , the first q-axis current command value Iq ^* , the d-axis current detection value Idc, and the q-axis current detection value Iqc, voltage correction values ΔVd ^* and ΔVq ^* are created. Using the voltage correction values ΔVd ^* and ΔVq ^* , the first d-axis current command value Id ^* , the first q-axis current command value Iq ^* , the estimated frequency value ω1c, and the electrical constant of the permanent magnet motor 1, according to the equation (36) , D-axis voltage command value Vdc ^* and q-axis voltage command value Vqc ^* can be applied to a vector control calculation method.

In the first to third embodiments, the three-phase AC currents iu, iv, and iw detected by the expensive current detector 3 are detected. However, the overcurrent detection for the power converter 2 is performed. The three-phase motor currents iu ^, iv ^, iw ^ can be reproduced from the direct current flowing through the one-shunt resistor attached to the, and the "low cost system" using this reproduced current value can also be supported. .

DESCRIPTION OF SYMBOLS 1 Permanent magnet motor 2 Power converter 3 Current detector 4 Coordinate conversion part 5 Axis error estimation part 6, 6a Frequency estimation part 6b Frequency calculation part 7 Phase estimation part 8, 8a, 8b, 8c Stabilization compensation part 9, 9a, 9b, 16, 22, 23, 24 Adder 10 Low-pass filter 11, 11a Vector control calculator 11b Voltage vector calculator 12 Coordinate converter 13, 13a d-axis current control calculator 14, 14a q-axis current control calculator 15 Speed Control calculation unit 17 d-axis current deviation switching unit 18 q-axis current deviation switching unit 19 Motor & compressor 20 Operation panel 21 DC power supply 25 Position detector 30 Phase error command calculation unit 35 Output voltage limit detection unit 100, 100a, 100b, 100c, 100d, 100e control device 150, 150a, 150b, 150c, 150d, 150e vector Control unit 155,155a, 155b, 155c, 155d, 155e reference axis correction portion 200,200a, 200b, 200c, 200d, 200e control system 300 screw compressor Idc d-axis current detection value Iqc q-axis current detection value Id ^* d-axis Current command value Iq ^* q-axis current command value Id ^** Second d-axis current command value Iq ^** Second q-axis current command value Vdc ^* d-axis voltage command value Vqc ^* q-axis voltage command value Vu ^* , Vv ^* , Vw ^* Control signal Iqctd Primary delay value I1 Primary current ωr Motor rotation frequency ω1c Frequency estimation value ω1 Frequency calculation value θdc Phase estimation value θdc_a, θdc_b, θdc_c, θdc_d New phase estimation value (other phase estimation values)
Δθd, Δθq Stabilization signal (differential calculation value, stabilization compensation value)
F _PLL control response frequency iu, iv, iw AC current (drive current)
iuc, ivc, iwc Current detection value Vd d-axis voltage value Vq q-axis voltage value Id d-axis current Iq q-axis current G loop transfer function

Claims (9)

A control device for a permanent magnet motor comprising a vector control unit for vector-controlling a permanent magnet motor via a power converter with a frequency command value as a target value,
A reference axis correction unit that corrects a control axis serving as a reference for the vector control using a differential operation value that is a change in a detected current value flowing through the permanent magnet motor ;
An axis error estimator for estimating an axis error that is a deviation between a phase estimated value obtained by integrating the frequency estimated value of the permanent magnet motor with respect to the control axis and a phase value of the permanent magnet motor;
A frequency estimator that performs a proportional operation or a (proportional + integral) operation so that an estimation result of the axial error estimator matches an axial error command value set to zero, and outputs a frequency estimated value; Prepared,
The reference axis correction unit is
When the control response frequency set for the proportional calculation or the (proportional + integral) calculation is equal to or greater than the value calculated by the following calculation formula FPLL_max , the correction is performed by calculating the differential calculation value ,

Ld: d-axis inductance value Lq: q-axis inductance value R: resistance value The vector control unit
Vector controller for a permanent magnet motor, characterized in that the front Kibe vector control using a modified shaft that is corrected by the reference axis correction portion. The current detection value is a d-axis current detection value or a q-axis current detection value,
The reference axis correction unit is
A phase estimator that integrates the frequency estimate and calculates the phase estimate;
A stability compensation unit for calculating the differential operation value using the d-axis current detection value or the q-axis current detection value,
The phase estimation value calculated by the phase estimation unit and the differential calculation value calculated by the stabilization compensation unit, and an addition unit for calculating another phase estimation value,
The vector control unit
Using the axis of the other phase estimation value as the correction axis, the d-axis voltage command value and the q-axis voltage command value are converted into a three-phase voltage command value, and this voltage command value is transferred to the power converter. The vector control device for a permanent magnet motor according to claim 1 , wherein the vector control device outputs the vector control device. The vector control unit
D-axis current command value of the rotational coordinate system, a q-axis current command value, the d-axis current detection value, and the q-axis current detection value, as well, by using the frequency estimate, the rotation frequency and the frequency command of the permanent magnet motor 2. The permanent magnet motor vector control device according to claim 1, wherein the control is performed so that the values coincide with each other. The stabilization compensator is
wherein a differential signal of the d-axis current detection value, any attenuation rate, and multiplies the gain calculated using the motor constant, by passing the multiplied value to the low-pass filter, to output the differential operation value The vector control device for a permanent magnet motor according to claim 2 . Differential operation value of the d-axis current detection value,
ξc: Arbitrary damping ratio (0 <ξc <1), s: Laplace operator Ld: d-axis inductance value, Lq: q-axis inductance value R: resistance value, Ke: induced voltage coefficient Id: d-axis current Formula Gd (s)

The vector control device for a permanent magnet motor according to claim 2 or 4 , wherein the vector control device is calculated by: The differential operation value of the q-axis current detection value is
the differential signal of the q-axis current detection value, as claimed in claim 2 or claim 4, characterized in that multiplying the gain calculated using any attenuation factor and the motor constant and a motor rotation frequency information of the permanent magnet motor Vector control device. Differential operation value of the q-axis current detection value,
ξc: Arbitrary damping rate 0 <ξc <1, s: Laplace operator Lq: q-axis inductance value,
R: resistance value, Ke: induced voltage coefficient, ω1c: estimated frequency value Iq: q-axis current, the following equation Gq (s)

Vector controller for a permanent magnet motor according to claim 2 or claim 4, characterized in that it is calculated according to. A permanent magnet motor;
A power converter for driving the permanent magnet motor;
A vector control system of a permanent magnet motor comprising a vector control device for vector controlling the permanent magnet motor via the power converter with a frequency command value as a target value,
The vector controller is
A reference axis correction unit that corrects a control axis serving as a reference for the vector control using a differential operation value that is a change in a detected current value flowing through the permanent magnet motor;
An axis error estimator for estimating an axis error that is a deviation between a phase estimated value obtained by integrating the frequency estimated value of the permanent magnet motor with respect to the control axis and a phase value of the permanent magnet motor;
A frequency estimator that performs a proportional operation or a (proportional + integral) operation so that an estimation result of the axial error estimator matches an axial error command value set to zero, and outputs a frequency estimated value; Prepared,
The reference axis correction unit is
When the control response frequency set for the proportional calculation or the (proportional + integral) calculation is equal to or greater than the value calculated by the following calculation formula FPLL_max , the correction is performed by calculating the differential calculation value ,

Ld: d-axis inductance value Lq: q-axis inductance value R: resistance value
The vector control unit
A vector control system for a permanent magnet motor, wherein the permanent magnet motor is vector-controlled using a correction shaft corrected by the reference axis correction unit . A screw compressor using the vector controller for a permanent magnet motor according to any one of claims 1 to 7 .

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