JP4556512B2 - Active converter and control method thereof - Google Patents

Description

  The present invention relates to an active converter and a control method therefor, and more particularly to charging / discharging a bootstrap capacitor.

  For example, Patent Document 1 discloses a technique for causing an active converter to function as a diode rectification mode by turning off a gate of a switching element of the active converter.

  For example, Patent Document 2 discloses a technique for providing a bootstrap capacitor and charging it for a driver that drives the high-arm side switching element of the multiphase inverter. This shows a technique for initially charging the bootstrap capacitor by turning on the low-arm side switching element in all phases.

Japanese Patent No. 3158212 JP 2001-275366 A

  Like an inverter, an active converter can be downsized by adopting a bootstrap capacitor. However, since the multiphase power supply is connected to the input of the active converter, if the low arm side switching element is turned on in all phases and the bootstrap capacitor is initially charged, the multiphase power supply is short-circuited. It will be.

  Accordingly, an object of the present invention is to provide a technique for charging a bootstrap capacitor without using a short circuit of a power supply while employing a bootstrap capacitor in an active converter. Another purpose is to reduce the capacitance of the bootstrap capacitor.

  A first aspect of the active converter (4) according to the present invention is between the positive power supply output terminal (P) and the negative power supply output terminal (H), and between the positive power supply output terminal and the negative power supply output terminal. Connected to the multi-phase power source (1) via a plurality of input lines (8R, 8S, 8T) corresponding to the respective phases (1R, 1S, 1T). A plurality of switching circuits (40R, 40S, 40T) provided and a switching signal generation circuit for outputting a switching signal (Sr, Ss, St, Sx, Sy, Sz) for controlling the operation of the switching circuit for each phase (43) and a smoothing capacitor (44) provided between the positive power output terminal and the negative power output terminal. Each of the switching circuits includes a high arm side switching element (41r; 41s; 41t) that controls whether or not a current flows from the positive power supply output terminal to the input line (8R; 8S; 8T) of the corresponding phase. A high-arm side diode (Dr; Ds; Dt) connected in parallel to the high-arm side switching element and connected to the input line of the corresponding phase and a cathode connected to the positive-side power supply output terminal ), A low-arm side switching element (41x; 41y; 41z) for controlling whether or not a current flows from the input line (8R; 8S; 8T) of the corresponding phase to the negative power supply output terminal, and the low arm A cathode connected in parallel with the side switching element and connected to the input line of the corresponding phase, and an anode connected to the negative power supply output terminal And (Dz Dx;; Dy), the switching control circuit for driving the high-arm side switching elements and said low-arm side switching element based on the switching signal low-arm side diode and a (42R; 42T; 42S). Each of the switching control circuits includes a low arm side driver element (422) that drives the low arm side switching element based on the first switching signal (Sx; Sy; Sz) of the corresponding phase, The high arm side driver element (421) that drives the high arm side switching element based on the two switching signals (Sr; Ss; St) and the input line (8R; 8S; 8T) of the corresponding phase A bootstrap capacitor having one end and the other end, and having a bootstrap capacitor (426) for supplying power to the high arm side driver element, a cathode connected to the other end of the bootstrap capacitor, and an anode Connected to a diode (424) and the anode of the bootstrap diode A positive electrode which has a negative electrode connected to the negative power output terminal, and a DC power source (423) supplies power to the low-arm side driver element.

And only for the phase taking the maximum value and the phase taking the minimum value, at least one of the low arm side switching element and the high arm side switching element is turned off, and the low arm side switching element and the high arm side switching element of the other phase are turned off. Are turned off to charge the bootstrap capacitor (426).

A second aspect of the active converter (4) according to the present invention is the active converter according to the first aspect, wherein the high-arm side switching element (41r; 41s; 41t) in all phases (R, S, T). ) And the low arm side switching element (41x; 41y; 41z) are both turned off, and the bootstrap capacitor (426) of each phase is charged.

A third aspect of the active converter (4) according to the present invention is the active converter according to the first aspect, wherein the multiphase power source (1) is a three-phase power source (1R, 1S, 1T), The high-arm side switching elements (41r; 41s; 41t) and the low-arm side switching elements (41x; 41y; 41z) are switched at an ON period of 120 degrees or less to charge the bootstrap capacitor (426) of each phase. To do.

A fourth aspect of the active converter according to the present invention (4), an active converter according to the third aspect, the high-arm side switching elements of all phases each (41r; 41t; 41s) is in the OFF state of Charge the bootstrap capacitor (426) of the phase.

A fifth aspect of the active converter (4) according to the present invention is the active converter according to any one of the third and fourth aspects, wherein the switching ON period is pulse width modulated.

A sixth aspect of the active converter (4) according to the present invention is the active converter according to any one of the first to fourth aspects, wherein at least one of the plurality of input lines (8R, 8S, 8T). One is provided with a current limiting circuit (2R, 2S, 2T), and further includes two or more main switches (5R, 5S, 5T).

  A first aspect of an active converter control method according to the present invention is a method of controlling an active converter (4), and the active converter includes a positive power supply output terminal (P) and a negative power supply output terminal (H). And a plurality of input lines (8R, 1T, 1T) corresponding to the respective phases (1R, 1S, 1T) to the multiphase power supply (1). 8S, 8T) and a plurality of switching circuits (40R, 40S, 40T) provided corresponding to each phase, and switching signals (Sr, Ss, St, A switching signal generation circuit (43) that outputs Sx, Sy, Sz) for each phase, and a smoothing capacitor (44) provided between the positive power output terminal and the negative power output terminal. Each of the switching circuits includes a high arm side switching element (41r; 41s; 41t) that controls whether or not a current flows from the positive power supply output terminal to the input line (8R; 8S; 8T) of the corresponding phase. A high-arm side diode (Dr; Ds; Dt) connected in parallel to the high-arm side switching element and connected to the input line of the corresponding phase and a cathode connected to the positive-side power supply output terminal ), A low-arm side switching element (41x; 41y; 41z) for controlling whether or not a current flows from the input line (8R; 8S; 8T) of the corresponding phase to the negative power supply output terminal, and the low arm A cathode connected in parallel with the side switching element and connected to the input line of the corresponding phase, and an anode connected to the negative power supply output terminal And (Dz Dx;; Dy), the switching control circuit for driving the high-arm side switching elements and said low-arm side switching element based on the switching signal low-arm side diode and a (42R; 42T; 42S). Each of the switching control circuits includes a high arm side driver element (421) for driving the high arm side switching element based on the first switching signal (Sr; Ss; St) of the corresponding phase, and the corresponding phase of the switching control circuit (Sr; Ss; St). A bootstrap capacitor (426) having one end connected to the input line (8R; 8S; 8T) and the other end and supplying power to the high arm side driver element; and the other end of the bootstrap capacitor A DC power supply having a bootstrap diode (424) having a connected cathode and an anode, a positive electrode connected to the anode of the bootstrap diode, and a negative electrode connected to the negative power supply output terminal (423). A parallel connection of the main switch (5R, 5S, 5T) and the current limiting circuit (2R, 2S, 2T) is inserted into at least one of the plurality of input lines (8R, 8S, 8T). The control method includes (a) turning off the main switch, passing a current between the active converter and the multiphase power source via the current limiting circuit, and charging the smoothing capacitor; ) After the step (a), turning on the main switch. In either of the steps (a) and (b), at least one of the low arm side switching element and the high arm side switching element is turned off only for the phase that takes the maximum value and the phase that takes the minimum value. Both the low arm side switching element and the high arm side switching element of the phase are turned off.

  A second aspect of the active converter control method according to the present invention is the active converter control method according to the first aspect, wherein, in the step (b), in all phases (R, S, T). The bootstrap capacitor (426) of each phase is charged with both the high arm side switching elements (41r; 41s; 41t) and the low arm side switching elements (41x; 41y; 41z) turned off.

  A third aspect of the active converter control method according to the present invention is the active converter control method according to the first aspect, wherein the multiphase power supply (1) is a three-phase power supply (1R, 1S, 1T). In the step (b), each of the high arm side switching elements (41r; 41s; 41t) and the low arm side switching elements (41x; 41y; 41z) is switched at an ON period of 120 degrees or less. Charge the bootstrap capacitor (426) of the phase.

  A fourth aspect of the active converter control method according to the present invention is the active converter control method according to the third aspect, wherein in the step (b), the high arm side switching elements (41r) of all phases are provided. 41s; 41t) is turned off, and the bootstrap capacitor (426) of each phase is charged.

A fifth aspect of the active converter control method according to the present invention is the active converter control method according to any one of the third aspect and the fourth aspect, wherein the on-period of the switching is pulse width modulation. Is done.

  In the first aspect of the active converter (4) according to the present invention, a smoothing capacitor (44) is obtained by current flowing through the plurality of input lines (8R, 8S, 8T) with the multiphase power supply (1). Thus, a DC voltage (Ed) is output between the positive power supply output terminal (P) and the negative power supply output terminal (H). The bootstrap capacitor (426) that supplies power to the high arm side driver element (421) is charged by the DC power supply (423) via the bootstrap diode (424). Therefore, the DC power supply (423) has a function of directly supplying power to the low arm side driver element (422) and a function of supplying power indirectly to the high arm side driver element (421) via the bootstrap capacitor (426). Can also be used.

The path for charging the bootstrap capacitor (426) includes the smoothing capacitor (44). Therefore, the bootstrap capacitor can be charged while avoiding a short circuit between the phase power supplies (1R, 1S, 1T).

In the second aspect of the active converter (4) according to the present invention, the bootstrap capacitor (426) is charged via the high arm side diode (Dr; Ds; Dt) and the low arm side diode (Dx; Dy; Dz). Thus, the bootstrap capacitor (426) can be charged while avoiding a short circuit between the phase power supplies (1R, 1S, 1T).

In the third aspect of the active converter (4) according to the present invention, the DC voltage (Ed) held by the smoothing capacitor (44) such as when the power supply voltage is unbalanced, the DC side load is extremely small, or when the converter is restarted. Is larger than the rectified voltage, the bootstrap capacitor (426) can be charged while avoiding a short circuit of the multiphase power supply (1) or the smoothing capacitor (44). In particular, by using a square wave with a width of 120 ° or less, it is possible to cope with a case where the phase accuracy with the power supply voltage is poor.

In the fourth aspect of the active converter (4) according to the present invention, the bootstrap capacitor (426) of each phase is charged with the high arm side switching elements (41r; 41s; 41t) turned off. The discharge current can be reduced and the required capacitance is small.

In the fifth aspect of the active converter (4) according to the present invention, since the duty can be varied by pulse modulation, it is possible to correspond to the pulse modulation after charging the bootstrap capacitor (426). Therefore, the circuit constants of the switching control circuits (42R, 42S, 42T) can be designed with a high degree of freedom.

  In the seventh aspect of the active converter (4) according to the present invention, the main switch (5R, 5S, 5T) is turned off to complete the charging of the smoothing capacitor (44) by the current limiting circuit (2R, 2S, 2T), The bootstrap capacitor (426) can then be charged by turning on the main switch. Therefore, it is possible to avoid a situation in which the voltage held by the bootstrap capacitor decreases during charging of the smoothing capacitor, and to reduce the capacitance of the bootstrap capacitor. Further, when charging the smoothing capacitor, the inrush current can be suppressed by the current limiting circuit (2R, 2S, 2T).

  In the first aspect of the active converter control method according to the present invention, the bootstrap capacitor (426) can be charged while avoiding a short circuit between the respective phase power supplies (1R, 1S, 1T) via the current limiting circuit. Thereby, the power capacity required for the current limiting circuit can be reduced. In addition, it is possible to charge the bootstrap capacitor while charging the smoothing capacitor in step (a), and it is possible to avoid a situation in which the voltage held by the bootstrap capacitor decreases during the charging of the smoothing capacitor. Therefore, it is possible to reduce the capacitance of the bootstrap capacitor, shorten the charging time, and shorten the activation time of the active converter.

  In the second aspect of the control method of the active converter according to the present invention, the bootstrap capacitor (426) is charged via the high arm side diode (Dr; Ds; Dt) and the low arm side diode (Dx; Dy; Dz). Thus, the bootstrap capacitor (426) can be charged while avoiding a short circuit between the phase power supplies (1R, 1S, 1T).

  In the third aspect of the control method of the active converter according to the present invention, the DC voltage (Ed) held by the smoothing capacitor (44), such as when the power supply voltage is unbalanced, the DC side load is extremely small, the converter is restarted, etc. Is larger than the rectified voltage, the bootstrap capacitor (426) can be charged while avoiding a short circuit of the multiphase power supply (1) or the smoothing capacitor (44). In particular, by using a square wave with a width of 120 ° or less, it is possible to cope with a case where the phase accuracy with the power supply voltage is poor.

  In the fourth aspect of the control method of the active converter according to the present invention, the bootstrap capacitor (426) of each phase is charged with the high arm side switching elements (41r; 41s; 41t) turned off. The discharge current can be reduced and the required capacitance is small.

  In the fifth aspect of the control method of the active converter according to the present invention, since the duty can be varied by pulse modulation, it is possible to correspond to the pulse modulation after charging the bootstrap capacitor (426). Therefore, the circuit constants of the switching control circuits (42R, 42S, 42T) can be designed with a high degree of freedom.

First embodiment.
FIG. 1 is a circuit diagram for explaining a first embodiment of the present invention. The multi-phase (here, three-phase) power source 1 has phase power sources 1R, 1S, and 1T, which are star-connected at a neutral point N. And on the opposite side to the neutral point N, they are connected to the input lines 8R, 8S, 8T, respectively. The input lines 8R, 8S, and 8T are provided with reactors 3R, 3S, and 3T, respectively, and the multiphase power source 1 is connected to the active converter 4 through the reactor group 3 that is constituted by the reactors 3R, 3S, and 3T. .

  However, main switches 5R and 5T are inserted in the input lines 8R and 8T, respectively, and constitute a main switch group 5. The phase power supplies 1R and 1T are connected to the reactors 3R and 3T via main switches 5R and 5T, respectively. The current limiting circuit 2R is connected in parallel with the main switch 5R. That is, in the input line 8R, a parallel connection between the main switch 5R and the current limiting circuit 2R is inserted. The current limiting circuit 2R has a function of preventing an inrush current.

  The active converter 4 includes a positive power supply output terminal P that outputs a DC voltage Ed and a negative power supply output terminal H, and a smoothing capacitor 44 is provided therebetween.

  The active converter 4 also includes switching circuits 40R, 40S, and 40T and a switching signal generation circuit 43 that are provided corresponding to each phase of the multiphase power supply 1. The switching signal generation circuit 43 generates switching signals Sr and Sx for controlling the operation of the switching circuit 40R, switching signals Ss and Sy for controlling the operation of the switching circuit 40S, and switching signals St and Sz for controlling the operation of the switching circuit 40T. And output.

  The switching circuits 40R, 40S, and 40T are all provided between the positive power supply output terminal P and the negative power supply output terminal H, and the input lines 8R, 8S, and 1T are connected to the phase power supplies 1R, 1S, and 1T, respectively. It is connected via 8T.

  The switching circuit 40R includes a high arm side switching element 41r, a low arm side switching element 41x, a high arm side diode Dr, a low arm side diode Dx, and a switching control circuit 42R.

  The high arm side switching element 41r controls whether or not current flows from the positive power supply output terminal P to the input line 8R. The low arm side switching element 41x controls whether or not a current flows from the input line 8R to the negative power source output terminal H. The high arm side diode Dr is connected in parallel with the high arm side switching element 41r, and has an anode connected to the input line 8R and a cathode connected to the positive power supply output terminal P. The low arm side switching element Dx is connected in parallel with the low arm side switching element 41x, and has a cathode connected to the input line 8R and an anode connected to the negative power supply output terminal H. The switching control circuit 42R drives the high arm side switching element 41r based on the switching signal Sr and the low arm side switching element 41x based on the switching signal Sx.

  The switching circuit 40S includes a high arm side switching element 41s, a low arm side switching element 41y, a high arm side diode Ds, a low arm side diode Dy, and a switching control circuit 42S.

  The high arm side switching element 41s controls whether or not a current flows from the positive power supply output terminal P to the input line 8S. The low arm side switching element 41y controls whether or not a current flows from the input line 8S to the negative power source output terminal H. The high arm side diode Ds is connected in parallel to the high arm side switching element 41s, and has an anode connected to the input line 8S and a cathode connected to the positive power supply output terminal P. The low arm side switching element Dy is connected in parallel to the low arm side switching element 41y, and has a cathode connected to the input line 8S and an anode connected to the negative power supply output terminal H. The switching control circuit 42S drives the high arm side switching element 41s based on the switching signal Ss and drives the low arm side switching element 41y based on the switching signal Sy.

  The switching circuit 40T includes a high arm side switching element 41t, a low arm side switching element 41z, a high arm side diode Dt, a low arm side diode Dz, and a switching control circuit 42T.

  The high arm side switching element 41t controls whether or not a current flows from the positive power supply output terminal P to the input line 8T. The low arm side switching element 41z controls whether or not a current flows from the input line 8T to the negative power source output terminal H. The high arm side diode Dt is connected in parallel with the high arm side switching element 41t, and has an anode connected to the input line 8T and a cathode connected to the positive power supply output terminal P. The low arm side switching element Dz is connected in parallel to the low arm side switching element 41z, and has a cathode connected to the input line 8T and an anode connected to the negative power supply output terminal H. The switching control circuit 42T drives the high arm side switching element 41t based on the switching signal St and drives the low arm side switching element 41z based on the switching signal Sz.

  As the high arm side switching elements 41r, 41s, 41t and the low arm side switching elements 41x, 41y, 41z, for example, IGBT (insulated gate bipolar transistor) elements can be adopted.

  The switching control circuits 42R, 42S, and 42T all have the same configuration, and FIG. 1 representatively shows the internal configuration circuit of the switching control circuit 42R. The switching control circuit 42R includes a high arm side driver element 421, a low arm side driver element 422, a DC power supply 423, a bootstrap diode 424, a bootstrap resistor 425, and a bootstrap capacitor 426.

  The high arm side driver element 421 drives the high arm side switching element 41r based on the switching signal Sr, and the low arm side driver element 422 drives the low arm side switching element 41x based on the switching signal Sx. The bootstrap capacitor 426 is connected to the input line 8R and the resistor 425, and is charged by being sandwiched between the two. Power is supplied to the high arm driver element 421 by the voltage across the bootstrap capacitor 426.

  More specifically, the input line 8R, the bootstrap capacitor 426, the bootstrap resistor 425, the bootstrap diode 424, the DC power source 423, and the negative power source output terminal H are connected in this order. However, the order of the bootstrap resistor 425 and the bootstrap diode 424 may be switched.

  In order to charge the bootstrap capacitor 426 by making the potential on the bootstrap resistor 425 side higher than the potential on the input line 8R side, the negative electrode of the DC power source 423 is at the negative power source output terminal H, and the positive electrode is at the bootstrap diode. The anode of 424 and the cathode of the bootstrap diode 424 are connected to the bootstrap capacitor 426 on the side opposite to the input line 8R.

  Note that the DC power supply 423 also has a function of supplying power to the low arm side driver element 422. Thus, the DC power supply 423 has a function of directly supplying power to the low arm side driver element 422 and a function of supplying power indirectly to the high arm side driver element 421 via the bootstrap capacitor 426.

  The DC power supply 423 can be obtained, for example, by stepping down and rectifying the line voltage of the multiphase power supply 1.

  In a state where the main switch group 5 is turned off and the power source 1 is disconnected from the active converter 4, similar to the technique disclosed in Patent Document 2, if all of the low arm side switching elements 41x, 41y, 41z are turned on, booting is performed. The strap capacitor 426 can be initially charged.

  However, if the power supply 1 is disconnected from the active converter 4 by the main switch group 5, the smoothing capacitor 44 cannot be charged. Therefore, it is necessary to connect the power source 1 to the active converter 4 in charging the smoothing capacitor 44.

  In general, a current limiting circuit is provided to avoid an inrush current in charging the smoothing capacitor. In order to reduce the power capacity of the current limiting resistor normally provided in the current limiting circuit, it is desirable to reduce the current flowing therethrough. However, this increases the time for charging the smoothing capacitor. If the bootstrap capacitor 426 cannot be charged while the smoothing capacitor 44 is being charged, the bootstrap capacitor 426 is statically charged so that the bootstrap capacitor 426 maintains the desired voltage during the smoothing capacitor 44 charging. The electric capacity must be increased.

  Although this is originally a bootstrap system for reducing the size of the switching control circuit 42R that drives the switching element at a carrier frequency of several kHz or more, the bootstrap capacitor 426 is increased in size.

  For example, the current limiting resistance provided in the current limiting circuit 2R has a rating of 180Ω / 10 W, the smoothing capacitor 44 has an electrostatic capacity of 2200 μF, a power supply voltage of 200 V, and the time for the DC voltage Ed to reach the rectified voltage is 8 seconds. It will be about. When the circuit current of the driver element 421 that drives the high-arm side switching element 41r is 500 μA, the drive power supply voltage is 15 V, and the voltage of the bootstrap capacitor 426 that decreases for about 8 seconds is allowed to 12 V (this is the high-arm side) This is a voltage at which the saturation voltage starts to increase when an IGBT element is used as the switching element 41r), and the capacitance required for the bootstrap capacitor 426 reaches 1300 μF.

  However, if the bootstrap capacitor 426 is charged after the smoothing capacitor 44 is charged, the capacitance required for the bootstrap capacitor 426 can be reduced.

  Therefore, first, the main switch group 5 is turned off, and the smoothing capacitor 44 is charged via the current limiting circuit 2R. Thereafter, the active converter 4 and the multiphase power supply 1 are connected via the main switch group 5. At this time, a current flows between the active converter 4 and the multiphase power supply 1 when there is a resistor loaded to discharge the electric charge of the smoothing capacitor or when a DC load is generated.

  In a normal converter operation, the high arm side switching element and the low arm side switching element are turned on / off in a complementary manner except for a so-called guard band (dead time) between different switching modes. Now, the value “1” is associated with turning on the high arm side switching element, the value “0” is associated with turning on the low arm side switching element, and the above values are weighted for each of the R, S, and T phases. They are 4, 2, and 1, respectively. For example, when the high arm side switching elements 41r, 41s, and 41t are on, off, and on, respectively, the low arm side switching elements 41x, 41y, and 41z are off, on, and off, respectively. The value obtained by multiplying the value indicating ON / OFF of the above by multiplying the above weighting over all phases is 4 · 1 + 2 · 0 + 1 · 1 = 5. This value is adopted as the switching mode number.

  Switching mode S0 shows a case where the low arm side switching element is turned on in all phases, and switching mode S7 shows a case where the high arm side switching element is turned on in all phases. These switching aspects must be avoided. This is because the phase power supplies 1R, 1S, and 1T are short-circuited via the input lines 8R, 8S, and 8T.

  However, a switching mode in which both the high arm side switching element and the low arm side switching element are turned off in a so-called intermediate phase is desired. An intermediate phase is a phase that is neither a phase that takes a maximum value or a phase that takes a minimum value at a certain point in time, and is a phase that switches between phases every 60 degrees.

  For example, when the phase voltage decreases in the order of S phase, T phase, and R phase, the intermediate phase is the T phase. When the high arm side switching element 41t is turned on, a current flows from the input line 8S to the input line 8T through the reactor 3S, the high arm side diode Ds, and the high arm side switching element 41t. When the low arm side switching element 41z is turned on, a current flows from the input line 8T to the input line 8R through the reactor 3T, the low arm side switching element 41z, and the low arm side diode Dx. These currents are not desirable because they will short-circuit the phases.

  Besides the switching modes S0 and S7, as described above, switching in the intermediate phase is excluded. Therefore, after all, at least one of the low arm side switching element and the high arm side switching element is turned off only for the phase that takes the maximum value and the phase that takes the minimum value, and the other phase is the low arm side switching element and the high arm side switching element. Both will be turned off. In such a switching mode, the smoothing capacitor 44 is always interposed in the path connecting the phase power supplies 1R, 1S, and 1T. Therefore, a short circuit between each phase power supply 1R, 1S, 1T can be avoided.

  As described above, since the intermediate phase is switched every time the phase is 60 degrees, the ON period of the low arm side switching element and the high arm side switching element is 120 degrees or less. As such a switching mode, for example, there is a diode rectification mode by so-called natural commutation in which a current flows in accordance with a voltage level between a phase having a maximum value and a phase having a minimum value.

  FIG. 2 is a circuit diagram illustrating the flow of current in such a switching mode. Since the switch group 5 is off, only the R phase and the S phase are shown.

  When the voltage (line voltage) of the S-phase power source 1S viewed from the R-phase power source 1R is positive, a current flows as indicated by a broken line, and the smoothing capacitor 44 is charged via the current limiting circuit 2R. At this time, there is a period during which the R-phase bootstrap capacitor 426 is also charged. FIG. 3 is an equivalent circuit diagram for explaining the charging of the bootstrap capacitor 426, and shows the case where the line voltage is positive, which is shown as a DC power supply E.

  When the charge amount of the bootstrap capacitor 426 is small, the R-phase low arm side diode Dx is reverse-biased, and the charging current of the smoothing capacitor 44 is a DC power source 423, a bootstrap diode 424, and a bootstrap diode as indicated by a chain line arrow. It flows through a resistor 425 and a bootstrap capacitor 426. Since the forward voltage drops of the low arm side diode Dx and the bootstrap diode 423 are substantially equal, the amount of charge of the bootstrap capacitor 426 increases, and if the voltage comparable to the voltage of the DC power supply 423 is maintained, the low arm side diode Dx A forward bias is applied to both ends. As a result, the charging current of the smoothing capacitor 44 flows as indicated by a broken line arrow.

Second embodiment.
FIG. 4 is a graph for explaining a period during which the R-phase bootstrap capacitor 426 can be charged in the diode rectification mode in a state where the switch group 5 is conducted. The voltages V _R0 , V _S0 , V _T0 , and V _N0 are all based on the voltage across the smoothing capacitor 44, that is, the midpoint of the potential between the positive power supply output terminal P and the negative power supply output terminal H. The DC potential applied by the smoothing capacitor 44 at the lines 8R, 8S, 8T and the neutral point N is shown. Further, the voltage V _RN is a DC voltage given by the smoothing capacitor 44 on the input line 8R when the neutral point N is used as a reference. In the same manner, if the subscript of the letter V is capitalized in the voltage notation, it indicates that the DC voltage is supplied by the smoothing capacitor 44. Since the high arm side diodes Dr, Ds, and Dt and the low arm side diodes Dx, Dy, and Dz are periodically turned on / off due to periodic fluctuations in the voltage applied by the polyphase power supply 1, these voltages are also based on the power supply frequency. fluctuate.

The voltages V _R0 , V _S0 , and V _T0 are the same as the phase voltages V _RN , V _SN , and V _TN because no current flows through the intermediate phase when the R phase, S phase, and T phase are the intermediate phases, respectively. It changes almost linearly. On the other hand, when these are not intermediate phases, the potential Ed / 2 of the positive power supply output terminal P or the potential (-Ed / 2) of the negative power supply output terminal H is taken.

The voltage V _RS appearing on the input line 8R with respect to the input line 8S as a reference by the DC voltage supplied from the smoothing capacitor 44 becomes negative at times t0 to t5 from the relationship between the voltages V _R0 and V _{S0 in} the figure. There is a drift of the voltage V _N0 between the midpoint of the potential of the positive power supply output terminal P and the negative power supply output terminal H and the potential of the neutral point N. Since _VRN = V _R0 −V _N0 , V _SN = V _S0 −V _N0 , V _TN = V _T0 −V _N0 , V _RN + V _SN + V _TN = 0, V _N0 = (V _R0 + V _S0 + V _T0 ) / 3, and the drift amount fluctuates between the voltage −Ed / 6 and Ed / 6.

The bootstrap capacitor 426 can be charged at a position where the phase voltage V _rN (broken line waveform) provided by the multiphase power supply 1 is lower than the voltage V _RN , and the S phase becomes an intermediate phase at times t3 to t5. since no flow current, out of the time t0 to t5, time t1~t3 bootstrap capacitor only by the R-phase and S-phase at (time t1, and the voltage _{V RN} and phase voltage _{V rN} at t3 matches) 426 can be charged.

Similarly, although the a DC voltage _{V TR} positive T-phase as seen from the R-phase is the time t1 to t6, R phase between times t3, t5 in which the voltage _{V RN} and phase voltage _{V rN} matches And the T-phase can be used to charge the bootstrap capacitor 426.

  From the above, the R-phase bootstrap capacitor 426 can be charged in the diode rectification mode between time t1 and time t5, and converted to 120 degrees. In other words, 240 degrees is sufficient for the period during which the bootstrap capacitor 426 is discharged.

  As described above, the bootstrap capacitor 426 is charged during the charging of the smoothing capacitor 44, and the discharge period after completion of the charging of the smoothing capacitor 44 can be shortened. Therefore, the capacitance required for the bootstrap capacitor 426 can be small. . If the capacitance required for the bootstrap capacitor 426 is calculated under the same conditions as described above, it becomes 2.2 μF, which can be significantly reduced compared to the above-described capacitance of 1300 μF. Therefore, the charging time of the bootstrap capacitor 426 can be shortened, and the starting time of the active converter 4 can be shortened.

FIG. 5 is a graph showing the behavior of the voltage V _boot across the bootstrap capacitor 426 after the main switch group 5 is turned on together with the voltage V _rN . However, the offset of the vertical axis is arbitrary, and the range of the voltage V _rN is 10 times the range of the both-end voltage V _boot . It is shown that the voltage V _{boot at} both ends is stabilized when about 10 cycles of the voltage V _rN have passed.

Third embodiment.
When the voltages V _R0 , V _S0 , and V _T0 in the second embodiment are seen, the high arm side or the low arm side is conducted except for the intermediate phase, so that the on-period is a square having a 120 ° width, not limited to the diode rectification mode. The high arm side switching elements 41r, 41s, 41t and the low arm side switching elements 41x, 41y, 41z may be switched so as to form a wave. In this case, switching modes other than the switching modes S0 and S7 are employed.

FIG. 6 is a graph illustrating values of the switching signals Sr, Ss, St, Sx, Sy, Sz and various voltages when the switching mode in which the ON period of the switching element is 120 degrees or less in the intermediate phase is repeatedly adopted. It is. The higher / lower waveform value indicates ON / OFF of the corresponding switching element. With such a switching mode, the line voltages V _RS , V _ST , and V _TR are energized with a width of 60 °, so the on period of each phase is 120 degrees.

For example switching mode in which the voltage V _RS is negative is a period of time t1~t3 when the natural commutation, since switching mode as the voltage V _TR positive is the time t3 to t5, the boot for a period of both The strap capacitor 426 can be charged.

Fourth embodiment.
The turning on of the high arm side switching element does not contribute to the enlargement of the conduction angle. Therefore, the bootstrap capacitor 426 can be charged by turning off the high arm side switching element. FIG. 7 is a graph showing the switching mode according to the present embodiment. Compared with the third embodiment, all the switching signals Sr, Ss, St are set to low, and the high arm side switching elements 41r, 41s, 41t are turned off. I am letting.

  By adopting such a switching mode, the discharge current of the bootstrap capacitor 426 can be reduced, and the required capacitance can be reduced.

  FIG. 8 is a graph showing a first modification of the present embodiment. In order to make the correspondence with FIG. 6 easier to understand, although the corresponding voltage vectors are shown together, the written voltage vectors are not always output. Here, the period during which the low arm side switching elements 41x, 41y, 41z are turned on (energization period) is further shortened. More specifically, energization is performed in a period of 60 ° around the center of the energization period. Thus, by setting the width of the square wave to 120 ° or less, it is possible to cope with the case where the switching operation has poor phase accuracy with respect to the power supply voltage.

  9 and 10 are both graphs showing a second modification of the present embodiment, and correspond to FIGS. 7 and 8, respectively. In this modification, pulse modulation is applied to a square wave. Since the duty can be varied by pulse modulation, it is possible to correspond to the pulse modulation after the bootstrap capacitor 426 is charged. Therefore, the circuit constants of the switching control circuits 42R, 42S, and 42T can be designed with a high degree of freedom.

Fifth embodiment.
The above operation can also be performed by providing a current limiting circuit on a plurality of input lines, for example, all input lines. FIG. 11 is a circuit diagram showing a modification on the multiphase power source 1 side of the reactor group 3 in the circuit of FIG. Current limiting circuits 2R, 2S, 2T are provided for all input lines 8R, 8S, 8T, respectively, and main switches 5R, 5S, 5T are provided in parallel for current limiting circuits 2R, 2S, 2T, respectively. Yes.

  All the main switches 5R, 5S, 5T are turned off, and the multiphase power supply 1 is connected to the active converter 4 via the current limiting circuits 2R, 2S, 2T, thereby charging the smoothing capacitor 44 while avoiding the inrush current. Can do. More specifically, at the initial charging stage of the smoothing capacitor 44, the low arm side diodes Dx, Dy, and Dz are reverse-biased by the DC power source 423, and the DC power source 423, the bootstrap diode 424, the bootstrap resistor 425, and the bootstrap capacitor. Flows through 426.

  When the amount of charge of the bootstrap capacitor 426 increases and a voltage approximately equal to the voltage of the DC power supply 423 is maintained, a forward bias is applied to both ends of the low arm side diodes Dx, Dy, and Dz.

  In this manner, the bootstrap capacitor 426 of each phase can be charged while avoiding a short circuit between the phase power supplies 1R, 1S, and 1T via the current limiting circuits 2R, 2S, and 2T, so that the current limiting circuit 2R is operated. During this period, the bootstrap capacitor 426 can be charged, and the power capacity required for the current limiting circuit can be reduced. Since the bootstrap capacitor 426 can be charged during the operation of the current limiting circuit 2R (while the smoothing capacitor 44 is being charged), the voltage held by the bootstrap capacitor 426 during charging of the smoothing capacitor 44 is reduced. Can be avoided. Therefore, the capacitance required for the bootstrap capacitor 426 can be reduced, and the charging time can be shortened.

1 is a circuit diagram illustrating a first embodiment according to the present invention. It is a circuit diagram explaining the flow of the electric current in 1st Embodiment concerning this invention. It is an equivalent circuit diagram explaining charge of a bootstrap capacitor. It is a graph explaining the period which can charge a bootstrap capacitor | condenser in 2nd Embodiment concerning this invention. It is the graph which showed the both-ends voltage of a bootstrap capacitor. It is a graph which illustrates the value of the switching signal in a 3rd embodiment concerning the present invention. It is a graph which illustrates the value of the switching signal in 4th Embodiment concerning this invention. It is a graph which illustrates the value of the switching signal in the 1st modification of the 4th embodiment concerning the present invention. It is a graph which illustrates the value of the switching signal in the 2nd modification of a 4th embodiment concerning the present invention. It is a graph which illustrates the value of the switching signal in the 2nd modification of a 4th embodiment concerning the present invention. It is a circuit diagram explaining a 5th embodiment concerning the present invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Multiphase power supply 1R, 1S, 1T Each phase power supply 4 Active converter 40R, 40S, 40T Switching circuit 41r, 41s, 41t High arm side switching element 41x, 41y, 41z Low arm side switching element 42R, 42S, 42T Switching control circuit 421 High arm Side driver element 422 Low arm side driver element 423 DC power supply 424 Bootstrap diode 426 Bootstrap capacitor 43 Switching signal generation circuit 44 Smoothing capacitor 8R, 8S, 8T Input line Dr, Ds, Dt High arm side diode Dx, Dy, Dz Low arm side Diode P Positive power output terminal H Negative power supply output terminal Sr, Ss, St, Sx, Sy, Sz Switching signal

Claims (11)

A positive power output terminal (P) and a negative power output terminal (H);
A plurality of input lines (8R, 8S, 1T) corresponding to each phase (1R, 1S, 1T) are provided between the positive power supply output terminal and the negative power supply output terminal to the multiphase power supply (1). 8T), a plurality of switching circuits (40R, 40S, 40T) provided corresponding to each phase,
A switching signal generation circuit (43) for outputting a switching signal (Sr, Ss, St, Sx, Sy, Sz) for controlling the operation of the switching circuit for each phase;
A smoothing capacitor (44) provided between the positive power output terminal and the negative power output terminal,
Each of the switching circuits
A high arm side switching element (41r; 41s; 41t) for controlling whether or not current flows from the positive power supply output terminal to the input line (8R; 8S; 8T) of the corresponding phase;
A high-arm side diode (Dr; Ds; Dt) connected in parallel to the high-arm side switching element and having an anode connected to the input line of the corresponding phase and a cathode connected to the positive-side power supply output terminal; ,
A low arm side switching element (41x; 41y; 41z) for controlling whether or not a current flows from the input line (8R; 8S; 8T) of the corresponding phase to the negative power source output terminal;
A low arm side diode (Dx; Dy; Dz) connected in parallel with the low arm side switching element and having a cathode connected to the input line of the corresponding phase and an anode connected to the negative power supply output terminal; ,
A switching control circuit (42R; 42S; 42T) for driving the high arm side switching element and the low arm side switching element based on the switching signal;
Each of the switching control circuits includes:
A low arm side driver element (422) for driving the low arm side switching element based on the first switching signal (Sx; Sy; Sz) of the corresponding phase;
A high arm side driver element (421) for driving the high arm side switching element based on the second switching signal (Sr; Ss; St) of the corresponding phase;
A bootstrap capacitor (426) having one end connected to the input line (8R; 8S; 8T) of the corresponding phase and the other end and supplying power to the high arm side driver element;
A bootstrap diode (424) having a cathode and an anode connected to the other end of the bootstrap capacitor;
A DC power supply (423) having a positive electrode connected to the anode of the bootstrap diode and a negative electrode connected to the negative power supply output terminal, and supplying power to the low arm driver element ; Only for the phase taking the maximum value and the phase taking the minimum value, at least one of the low arm side switching element and the high arm side switching element is turned off, and both the low arm side switching element and the high arm side switching element of the other phase There you charge the bootstrap capacitor is turned off (426), active converter (4).   In all phases (R, S, T), the high-arm side switching elements (41r; 41s; 41t) and the low-arm side switching elements (41x; 41y; 41z) are both turned off. The active converter (4) of claim 1, wherein 426) is charged. The multi-phase power source (1) is a three-phase power source (1R, 1S, 1T),
The bootstrap capacitor (426) of each phase is charged by switching the high arm side switching element (41r; 41s; 41t) and the low arm side switching element (41x; 41y; 41z) with an on period of 120 degrees or less. The active converter (4) according to claim 1, wherein:   The active converter (4) according to claim 3, wherein the bootstrap capacitor (426) of each phase is charged in a state in which the high arm side switching elements (41r; 41s; 41t) of all phases are turned off. The active converter (4) according to any one of claims 3 and 4, wherein the on period of the switching is pulse width modulated.   The current limiting circuit (2R, 2S, 2T) is provided in at least one of the plurality of input lines (8R, 8S, 8T), and further includes two or more main switches (5R, 5S, 5T). The active converter (4) according to any one of claims 1 to 4. The active converter (4)
A positive power output terminal (P) and a negative power output terminal (H);
A plurality of input lines (8R, 8S, 1T) corresponding to each phase (1R, 1S, 1T) are provided between the positive power supply output terminal and the negative power supply output terminal to the multiphase power supply (1). 8T), a plurality of switching circuits (40R, 40S, 40T) provided corresponding to each phase,
A switching signal generation circuit (43) for outputting a switching signal (Sr, Ss, St, Sx, Sy, Sz) for controlling the operation of the switching circuit for each phase;
A smoothing capacitor (44) provided between the positive power output terminal and the negative power output terminal,
Each of the switching circuits
A high arm side switching element (41r; 41s; 41t) for controlling whether or not current flows from the positive power supply output terminal to the input line (8R; 8S; 8T) of the corresponding phase;
A high-arm side diode (Dr; Ds; Dt) connected in parallel to the high-arm side switching element and having an anode connected to the input line of the corresponding phase and a cathode connected to the positive-side power supply output terminal; ,
A low arm side switching element (41x; 41y; 41z) for controlling whether or not a current flows from the input line (8R; 8S; 8T) of the corresponding phase to the negative power source output terminal;
A low arm side diode (Dx; Dy; Dz) connected in parallel with the low arm side switching element and having a cathode connected to the input line of the corresponding phase and an anode connected to the negative power supply output terminal; ,
A switching control circuit (42R; 42S; 42T) for driving the high arm side switching element and the low arm side switching element based on the switching signal;
Each of the switching control circuits includes:
A high arm side driver element (421) for driving the high arm side switching element based on the first switching signal (Sr; Ss; St) of the corresponding phase;
A bootstrap capacitor (426) having one end connected to the input line (8R; 8S; 8T) of the corresponding phase and the other end and supplying power to the high arm side driver element;
A bootstrap diode (424) having a cathode and an anode connected to the other end of the bootstrap capacitor;
A DC power supply (423) having a positive electrode connected to the anode of the bootstrap diode and a negative electrode connected to the negative power supply output end;
At least one of the plurality of input lines (8R, 8S, 8T) includes a parallel connection of a main switch (5R, 5S, 5T) and a current limiting circuit (2R, 2S, 2T),
(A) turning off the main switch, passing a current between the active converter and the multiphase power source via the current limiting circuit, and charging the smoothing capacitor;
(B) after the step (a), turning on the main switch,
In either of the steps (a) and (b), at least one of the low arm side switching element and the high arm side switching element is turned off only for the phase that takes the maximum value and the phase that takes the minimum value. An active converter control method in which both the low arm side switching element and the high arm side switching element are turned off. In the step (b), the high arm side switching elements (41r; 41s; 41t) and the low arm side switching elements (41x; 41y; 41z) are both turned off in all phases (R, S, T). Charging the bootstrap capacitor (426) of each phase;
The method of controlling an active converter according to claim 7. The multi-phase power source (1) is a three-phase power source (1R, 1S, 1T),
In the step (b), the high-arm side switching elements (41r; 41s; 41t) and the low-arm side switching elements (41x; 41y; 41z) are switched at an ON period of 120 degrees or less, so The method of controlling an active converter according to claim 7, wherein the bootstrap capacitor (426) is charged.   10. The active converter according to claim 9, wherein, in the step (b), the bootstrap capacitor (426) of each phase is charged with the high-arm side switching elements (41 r; 41 s; 41 t) of all phases being turned off. Control method. 11. The method of controlling an active converter according to claim 9, wherein the ON period of the switching is pulse width modulated.

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