JP4515202B2 - OFDM diversity receiver - Google Patents

Description

  The present invention relates to an OFDM receiver. Specifically, the present invention relates to a diversity reception technique for receiving an OFDM transmission signal using a plurality of antennas.

  In Japanese terrestrial digital television broadcasting, an OFDM (Orthogonal Frequency Division Multiplexing) system is adopted as a transmission system. The OFDM scheme is one of multicarrier transmission schemes in which a transmission signal is divided into a plurality of carriers and transmitted, and the spectrum of each subchannel can be densely arranged, which is strong against frequency selective fading of a multipath transmission path. There are advantages such as high frequency utilization efficiency.

  In addition, when a digital broadcast is mobilely received by an in-vehicle receiver that employs the OFDM method, diversity reception is performed in order to improve the quality of the received signal. There is a maximum ratio combining (MRC) method as an algorithm for combining the signals of a plurality of branches received with diversity.

  The MRC method is a method of determining a weighting coefficient for the subcarriers of each branch based on the transmission path power for each subcarrier of each branch and synthesizing the signals for each weighted subcarrier. The MRC method is a method of synthesizing a signal on the assumption that the amplitude of the transfer function of the branch is large = the C / N ratio (Carrier to Noise ratio) of the branch is good.

Specifically, a synthesis method by the conventional MRC method will be described. The k-th subcarrier signal of the i-th symbol of the p-th branch subjected to the FFT operation is represented as X _p (i, k). Then, for each signal X _p (i, k), respectively weighting factor W _p (i, k) is calculated, the signals X _p (i, k) to the weighting factor W _p (i, k) is Multiply and synthesize.

Here, the weighting coefficient W _p (i, k) is expressed by the following equation (1). Here, H _p (i, k) is the transmission path response of the k-th subcarrier of the i-th symbol of the p-th branch, and H _p ^* (i, k) represents its complex conjugate.

Also, a synthesized signal obtained by synthesizing the signals of each branch multiplied by the weighting coefficient W _p (i, k), that is, the k-th subcarrier synthesized signal Y (i, k) of the i-th symbol is expressed by the following equation (2). It is represented by Here, from S (i, k) × H _m = X _m (i, k), it can be seen that Formula 2 is established. S (i, k) is a desired signal included in the received signal X (i, k). That is, the composite signal Y (i, k) = S (i, k), and the desired signal S (i, k) is completely restored.

As expressed by Equation 1, the weighting coefficient W _p (i, k) depends on the amplitude of the transmission path response (transfer function) H. That is, a signal with a large amplitude of the transmission line response H is multiplied by a large weighting coefficient W, and the signal of that branch is emphasized. However, actually, even when the amplitude of the transmission line response H is large, the signal of the branch may not be strong or the C / N ratio of the branch may not be good.

  For example, although the C / N ratio of a signal of a certain branch is poor and the amplitude is small, there is a case where the amplitude increases due to the function of AGC (automatically gain control) when receiving, and the amplitude of the transmission line response H also increases. . In such a case, when synthesizing by the conventional MRC method, a branch with a good C / N ratio is pulled by a branch with a poor C / N ratio, and the C / N ratio after synthesis becomes poor.

  When an experiment for synthesizing the received signal by the MRC method with diversity reception of the 13-segment OFDM signal by the in-vehicle receiving apparatus is actually performed, if there is a large difference in the C / N ratio of the branch signal to be synthesized, There was a problem that the BER (bit error rate) was worse than the BER when single reception was performed on a branch having a good C / N ratio.

  In the case of mobile reception, the C / N ratio may vary greatly depending on the position of the antenna. In this case, there is a large difference in the C / N ratio of the MRC combining branch. Since the MRC synthesis algorithm is an algorithm developed on the assumption that there is no great difference in C / N of the branch to be synthesized, it cannot be handled when there is a great difference in C / N of the branch.

  In view of the above problems, an object of the present invention is to provide a receiving apparatus that can perform diversity reception with high adaptability to the reception environment and obtain stable reception quality.

In order to solve the above-mentioned problem, the invention according to claim 1 is an FFT operation for N (N is an integer) antennas for diversity reception of OFDM transmission signals and for each signal received by the N antennas. A circuit that outputs N symbol signals; a circuit that calculates a weighting value based on the C / N ratio for each symbol signal; and a circuit that outputs N first weighting values; and a transmission path estimation for each symbol signal A circuit that calculates a value and outputs N channel estimation values, a circuit that multiplies each symbol signal by each first weighting value to calculate N first weighting signals, and each channel estimation A circuit that multiplies the value by each first weighting value to calculate N weighted transmission channel estimation values, and a synthesis circuit that performs MRC combining using each weighting transmission channel estimation value and each first weighting signal It equipped with a door, The circuit for outputting the N first weight values includes a C / N arithmetic circuit for calculating a C / N ratio for each symbol signal for each symbol signal, and zero is provided for subcarriers constituting one symbol signal. A subcarrier for carrying the embedded dummy data and a subcarrier for carrying data other than the dummy data are included, and the C / N arithmetic circuit starts from a subcarrier carrying the dummy data. The acquired signal value is defined as noise power, the signal acquired from a subcarrier carrying data other than the dummy data is defined as carrier power, and the C / N ratio is determined based on the ratio of the noise power and the carrier power. It is characterized by calculating.

According to a second aspect of the present invention, in the OFDM diversity receiver according to the first aspect, the carrier power is acquired from a subcarrier carrying an SP signal among subcarriers carrying data other than the dummy data. It is characterized by calculating from the obtained signal.

According to a third aspect of the invention, the OFDM diversity receiver according to claim 1, wherein the carrier power, among the subcarriers data other than the dummy data is carried, the sub-carrier for transporting the SP signal or CP signal It is characterized in that it is calculated from the signal acquired from

According to a fourth aspect of the present invention, in the OFDM diversity receiver according to any one of the first to third aspects, the noise power is not removed by a filter among subcarriers carrying the dummy data. It is calculated from the signal acquired from the subcarrier of the band.

According to a fifth aspect of the present invention, in the OFDM diversity receiver according to any one of the first to fourth aspects, the circuit that outputs the first N weighting values includes a C / N ratio and the first number. A lookup table that converts the C / N ratio calculated using the C / N arithmetic circuit into the first weighting value by referring to a table that associates the weighting value of 1. It is characterized by that.

According to a sixth aspect of the present invention, in the OFDM diversity receiver according to any one of the first to fifth aspects, the circuit that outputs the first N weighting values includes the first weighting in each branch. The value is calculated by a calculation independent of a signal received by another branch.

According to a seventh aspect of the present invention, in the OFDM diversity receiver according to any one of the first to sixth aspects, the combining circuit transmits the conjugate complex value of each weighted transmission path estimation value to all N weighted transmissions. A circuit that calculates N second weighting values by dividing by the square sum of the path estimation values, and N second weighting values by multiplying each first weighting signal by each second weighting value. The circuit includes a circuit for calculating a weighted signal and a circuit for calculating a combined received signal by adding the N second weighted signals.

  In the present invention, since the received signal and the transmission path response are weighted by the C / N ratio and then MRC synthesis is performed, a large weight is assigned to a signal in a branch having a good C / N ratio, or the C / N ratio is assigned. It is possible to assign a small weight to a signal of a bad branch and output a synthesized signal. Thereby, the quality of the synthesized signal is improved.

{First embodiment}
The first embodiment of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram of an OFDM diversity receiver DR according to an embodiment of the present invention.

The diversity receiver DR of the present embodiment is a receiver with N branches. That comprises the N antenna 11 ₁ to 11 _N, by treatment with the receiving device DR ₁ ~DR _N to N branches a signal received from each antenna 11 ₁ to 11 _N respectively, and outputs the combined signal.

The configuration and processing content of each receiver device DR ₁ ~DR _N will be described. Since the configuration of each reception device DR ₁ ~DR _N is the same, in the following, a common description for the configuration and processing content of each branch.

Signals received from the antennas 11 _{1 to} 11 _N are processed by the front end processing units 12 _{1 to} 12 _N , respectively. In the front end processing units 12 _{1 to} 12 _N , the received signal is subjected to frequency conversion and filter processing and then AD conversion. Received digital signal output from the front-end processing unit 12 ₁ to 12 _N are input to the FFT operation unit 13 ₁ to 13 _N. In the FFT operation units 13 _{1 to} 13 _N , the time-domain OFDM symbol signal is converted into a frequency-domain OFDM symbol signal. Here, the output frequency domain signal is represented by X _p (i, k). Here, p is a branch number (an integer from 1 to N), i is a symbol number, and k is a subcarrier number. In the figure, for the signal X and the transmission path response H, only the branch number is added, and notations such as i and k are omitted. The signal X _p (i, k) after the FFT calculation is output to the C / N calculation units 14 _{1 to} 14 _N , the transmission path estimation units 15 ₁ to 15 _N , and the multiplication circuits 17 _{1 to} 17 _N.

In the present embodiment, the front-end processing units 12 _{1 to} 12 _N , the FFT calculation units 13 _{1 to} 13 _N , the C / N calculation units 14 _{1 to} 14 _N , and the transmission path estimation units 15 ₁ to 15 _N are Although configured by hardware circuits, some or all of these may be realized by software processing.

The transmission path estimators 15 ₁ to 15 _N calculate transmission path response estimation values based on the signals X _p (i, k) input from the FFT calculators 13 _{1 to} 13 _N. The transmission path estimators 15 ₁ to 15 _N calculate, for example, a transmission path response estimated value by dividing a received pilot signal by this complex amplitude using a pilot signal having a known complex amplitude. Furthermore, the transmission path response H _p (i, k) of each received data signal is obtained by interpolating the calculated transmission path response estimation value in the symbol direction and the carrier direction. The obtained transmission line response H _p (i, k) is output to the multiplication circuits 18 ₁ to 18 _N. At the same time, the pilot signal SP extracted from X _p (i, k) when performing transmission path estimation is output to the C / N computing units 14 _{1 to} 14 _N.

C / N calculation unit 14 ₁ to 14 _N, the signal from the FFT computation block _{13 1 ~13 N X p (i} , k) inputs the inputs SP signal from the channel estimation unit 15 ₁ to 15 _N. Then, the C / N arithmetic units 14 _{1 to} 14 _N calculate the C / N ratio using the signal X _p (i, k) and the SP signal. Here, in the first embodiment, it is possible to calculate the C / N ratio without particularly distinguishing the SP signal, and the C / N arithmetic units 14 _{1 to} 14 _N input the SP signal. Is not required. However, in the second embodiment, which will be described later, since the features of the SP signal are used, the C / N arithmetic units 14 _{1 to} 14 _N need to input the SP signal. A method for calculating the C / N ratio will be described with reference to FIG.

  FIG. 2 is a diagram illustrating a spectrum of a received OFDM signal. In the figure, the horizontal axis indicates the frequency f, and the vertical axis indicates the signal intensity. The figure shows the spectrum of an OFDM symbol signal in the Japanese mode 3 standard. As shown in the figure, there is a no-signal section in one symbol OFDM signal. In mode 3, there are 8192 subcarriers in one symbol, among which 5617 are data signals (in this case, data signals include pilot signals such as SP (Scattered pilot) signals in addition to actual data). And the control signal such as an AC (Auxiliary Channel) signal is also carried), and the remaining 2575 are carriers for carrying zero-padded dummy data when IFFT modulation is performed on the transmission device side. It is. FIG. 2 shows that in the signal of one symbol, the central 2575 subcarriers are carriers for carrying dummy data, and there are 5617 subcarriers for carrying data signals at both ends.

  The OFDM signal in which the data signal and the dummy data signal are buried in the OFDM transmitter is received by the receiver via the transmission path, and is again subjected to FFT operation and demodulated. Therefore, the demodulated signal of the subcarrier in which the dummy data signal is embedded is a pure noise signal. Therefore, in the present embodiment, the C / N ratio for each symbol is calculated by calculating the intensity of the noise signal.

  Formula 3 is an example of a calculation formula for calculating the C / N ratio for each symbol. Here, since a particular symbol signal is focused on, the branch number and symbol number are omitted.

  In Formula 3, 2575 subcarriers (the left end in FIG. 2 as the first and 2575 carriers from the left end) and the 2810th to 5384th dummy in which data signals from the first to 2575th are embedded, and The C / N ratio is calculated using 2575 subcarriers embedded with data signals. That is, the denominator is a value obtained by adding the squares of absolute values of pure noise signals acquired from 2575 subcarriers. The numerator is a value obtained by subtracting the sum of squares of absolute values of pure noise signals acquired from 2575 subcarriers from the sum of squares of absolute values of data signals inserted into 2575 subcarriers. The sum of squares of the data signal from which the signal is removed is shown. Note that π / 2 in the denominator is a correction coefficient for making the C / N in the frequency domain coincide with the C / N in the time domain.

  However, although the subcarriers from the first to the 2575th are selected here as the subcarriers in which the data signal is embedded, any subcarrier other than the subcarriers from the 2810th to the 5384th is selected. You may choose. In addition, although 2575 subcarriers are used here, the number of subcarriers used in the calculation may be less than 2575.

When the C / N ratio for each symbol is calculated by such calculation, the C / N arithmetic units 14 _{1 to} 14 _N output the calculated C / N ratios to the C / N tables 16 _{1 to} 16 _N. To do.

C / N table 16 ₁ ~ 16 _N is a lookup table, as shown in Table 1, the value of C / N ratio, holds a table associating a weight value, the C / N ratio When input, it has a function of outputting a corresponding weight value. In this embodiment, the value of the C / N ratio is divided into eight ranges, and the weight values w1 to w8 are associated with each range. Further, w1 to w8 are values that take decimal numbers from 0 to 1, and each value is represented by 8 bits. In this way, the C / N ratio-weight value conversion is a lookup table method, so that complicated calculations are omitted and the circuit scale is reduced.

When the C / N tables 16 _{1 to} 16 _N receive the value of the C / N ratio for each symbol, any one of the weight values w1 to w8 is multiplied by the multiplication circuits 17 _{1 to} 17 _N and the multiplication circuits 18 ₁ to 18 _N. Output to. The weight values calculated for each branch are represented by σ ₁ to σ _N.

Next, in multiplication circuits 17 _{1 to} 17 _N , signals X ₁ (i, k) to X _N (i, k) and weight values σ _{1 to} σ _N are multiplied. Then, the calculation results σ ₁ X _{1 to} σ _N X _N are output to the multiplication circuits 20 _{1 to} 20 _N , respectively.

Further, in the multiplication circuit 18 ₁ ~ 18 _N, the channel response H ₁ to H _N and the weight value σ ₁ ~σ _N is multiplied. The calculation results σ ₁ H _{1 to} σ _N H _N are output to the weight calculation units 19 _{1 to} 19 _N of all branches, respectively. That is, σ ₁ H ₁ output from the multiplication circuit 18 _{1 is} output to the weight calculation units 19 _{1 to} 19 _{N of} all branches, and σ ₂ H ₂ output from the multiplication circuit 18 ₂ so on ... are output to the weight calculation unit 19 ₁ ~ 19 _N of branches, is the output of the multiplier circuit 18 _p is output to all the weight calculation unit 19 ₁ ~ 19 _N.

Then, each of the weight calculation units 19 _{1 to} 19 _N receives the calculation results σ ₁ H _{1 to} σ _N H _N output from the multiplication circuits 18 ₁ to 18 _{N of} all the branches, and uses these N values as elements. The weight values W _{1 to} W _N are calculated. That is, the weight calculation unit 19 ₁ receives the calculation results σ ₁ H _{1 to} σ _N H _N output from the multiplication circuits 18 ₁ to 18 _{N of} all branches, calculates the weight value W ₁ , and the weight calculation unit. 19 ₂ receives the calculation results σ ₁ H _{1 to} σ _N H _N output from the multiplication circuits 18 ₁ to 18 _{N of} all the branches, calculates the weight value W _2, and so on. 19 _P inputs the operation results σ ₁ H _{1 to} σ _N H _N output from the multiplication circuits 18 ₁ to 18 _{N of} all branches, and calculates the weight value W _P.

Formula 4 is a diagram illustrating a calculation formula of the weight values W _{1 to} W _N obtained by the weight calculation units 19 _{1 to} 19 _N. In Equation 4, H _P ^* (i, k) is a complex conjugate of H _P (i, k).

When the weight values W _{1 to} W _N are output from the weight calculation units 19 _{1 to} 19 _N , the multiplication circuits 20 _{1 to} 20 _N output the weight values W _{1 to} W _N and the multiplication circuits 17 _{1 to} 17 _N. The values σ ₁ X _{1 to} σ _N X _N are multiplied, and the multiplication results W ₁ σ ₁ X _{1 to} W _N σ _N X _N are output.

Then, the adder circuit 31 adds the output values W ₁ σ ₁ X _{1 to} W _N σ _N X _N of the multiplier circuits 20 _{1 to} 20 _N and outputs a composite signal Y (i, k) expressed by Equation 5 below. To do.

  The output synthesized signal Y (i, k) is returned to an integer signal by the demapping process in the demapping unit 32 and then output to the FEC (forward error coding) unit. In the FEC section, Viterbi decoding and Reed-Solomon decoding are performed.

  As described above, according to the present embodiment, the weight value σ based on the C / N ratio is obtained for each symbol of the received OFDM signal, and both the received signal X and the transmission path response H are multiplied by this weight value σ. Thereafter, the weight value W is calculated. That is, in diversity combining, weighting is performed for each branch based on the C / N ratio, so a large weight value W is assigned to a signal in a branch having a good C / N ratio. As a result, the problem that the signal quality deteriorates due to the branch having a poor C / N ratio although the amplitude of the transmission line response H is large as in the prior art can be solved.

  Even if the value of the C / N ratio is small, the BER of the synthesized signal is always improved as compared with the case of single reception by assigning and combining a small weight value W.

  Furthermore, in the present invention, a subcarrier signal in which a dummy data signal is embedded is used, and the C / N ratio is calculated using this signal as a noise signal. Therefore, the calculation accuracy of the C / N ratio can be improved. It is. Further, the C / N ratio in each branch is calculated using the symbol signal in each branch, and is obtained by a calculation independent of the signals in other branches. Therefore, it is possible to reduce the data flow between the circuits of each branch.

In FIG. 1, in order to make the drawing easy to see, each branch is illustrated as having C / N tables 16 _{1 to} 16 _N , but in this embodiment, the C / N tables are all included. A common lookup table is prepared for each branch and shared. In this way, the circuit scale can be reduced.

{Second Embodiment}
Next, a second embodiment of the present invention will be described. The block configuration of the OFDM diversity receiver DR in the second embodiment is the same as that in the first embodiment shown in FIG. In the second embodiment, only the C / N calculation method in the C / N arithmetic units 14 _{1 to} 14 _N is different from the first embodiment.

  In the first embodiment, the C / N ratio is calculated using 2575 subcarriers embedded with dummy data and 2575 subcarriers embedded with a data signal. However, in some cases, all 2575 subcarriers in which these dummy data signals are embedded cannot be used. For example, if a tuner is subjected to a SAW (surface acoustic wave) filter, an ACI (adjacent channel interface) filter, or the like on the received signal, only a part of these 2575 subcarriers can be used. That is, these filters are used for the purpose of extracting the subcarriers of the data signal portion, and are filters that pass through the 0-2809th subcarrier region and the 5385-8192th subcarrier region in FIG. It is. Therefore, among the 2575 subcarriers in which the dummy data signal is embedded, only a carrier having a frequency close to that of the subcarrier in which the data signal is embedded can be used. In other words, since the filter is designed to pass a band in a slightly wider range than the frequency band that is originally required, there is a band that passes through the filter even around the frequency band that is originally passed by the filter. To do.

  For example, when an ACI filter is used, the filter width is 6 MHz. On the other hand, the subcarrier band in which the data signal of the OFDM signal is embedded is 5.572 MHz. Therefore, since subtracting 5.572 MHz from 6 MHz is 428 kHz, there are subcarriers in which dummy data signals usable in 214 kHz are embedded at both ends of the subcarriers in which the data signals are embedded. Here, in the case of mode 3, since the carrier interval is 0.992 kHz, there are about 214 usable subcarriers.

  Therefore, the C / N ratio is calculated using 214 × 2 subcarriers at both ends and 428 subcarriers in which the data signal is embedded. Furthermore, in this embodiment, since the number of subcarriers embedded with data signals necessary for calculating the C / N ratio is 428, SP (scattered pilot) signals or CP (continual pilot) signals are embedded. Use subcarriers. Since 468 SP signals are inserted per symbol and one CP signal is inserted per symbol signal, a carrier in which an SP signal (or SP signal and CP signal) and a dummy data signal are embedded is used. The C / N ratio is calculated.

  Formula 6 is a calculation formula for calculating the C / N ratio from the SP signal and the dummy data signal.

  In Equation 6, SP indicates a signal value of 428 SP signals selected from 468 SP signals. Alternatively, one CP signal may be selected from 428. Further, ε is a coefficient for correcting the C / N ratio calculated in the frequency domain and the C / N ratio calculated in the time domain to coincide with each other. In selecting an SP signal, it is preferable to consider CCI (co-channel interference). That is, it is desirable to extract the data signal from the 468 SP signals excluding the SP signal with CCI.

When the C / N ratio is calculated in this way, the weight values w1 to w8 are determined using the C / N tables 16 _{1 to} 16 _N in the same manner as in the first embodiment. The composite signal Y (i, k) is calculated by performing the calculation shown in equation (5).

  As described above, in the second embodiment, as in the first embodiment, diversity combining is performed after weighting each branch based on the C / N ratio. It is possible to improve.

  In particular, even when a SAW filter or an ACI filter is applied to the received signal, it is possible to obtain the C / N ratio by using a small number of dummy data signals that pass through the filter.

  Furthermore, since the dummy data signal is a noise signal, the SP signal (or the SP signal and the CP signal) is used as the data signal. That is, since all the SP signals have the same amplitude, there is no variation in the amplitude of the data signal, and the calculation of the C / N ratio can be performed stably. In addition, since the SP signal arrangement position is always uniform within one symbol and inserted every 12 carriers, it is possible to select a data signal in a balanced manner from one symbol, and the C / N ratio It is possible to perform the calculation stably.

  However, also in the second embodiment, selection may be made from all data signals without considering the SP signal or the CP signal. However, since the number of noise signals for calculating the C / N ratio is smaller than that in the first embodiment, it is desirable to use an SP signal or a CP signal. The data signal used when calculating the C / N ratio in the first embodiment may include an SP signal or a CP signal.

It is a block diagram of an OFDM diversity receiver. It is a figure which shows the spectrum of an OFDM symbol signal.

Explanation of symbols

DR OFDM diversity receivers 11 _{1 to} 11 _N antennas 12 _{1 to} 12 _N front end processing units 13 _{1 to} 13 _N FFT operation units 14 _{1 to} 14 _N C / N operation units 15 ₁ to 15 _N transmission path estimation units 16 ₁ to 16 _N C / N table 17 ₁ to 17 _N multiplying circuits 18 ₁ ~ 18 _N multiplying circuits 19 ₁ ~ 19 _N weight value calculation unit 20 ₁ to 20 _N multiplying circuit 31 the adding circuit 32 demapping unit

Claims (7)

N (N is an integer) antennas for diversity reception of OFDM transmission signals;
A circuit that performs an FFT operation on each signal received by the N antennas and outputs N symbol signals;
A circuit that calculates a weighting value based on the C / N ratio for each symbol signal and outputs N first weighting values;
A circuit that calculates a channel estimation value for each symbol signal and outputs N channel estimation values;
A circuit that multiplies each symbol signal by each first weighting value to calculate N first weighting signals;
A circuit for multiplying each transmission path estimation value by each first weighting value to calculate N weighting transmission path estimation values;
A combining circuit that performs MRC combining using each weighted transmission path estimation value and each first weighting signal;
Bei to give a,
The circuit that outputs the N first weight values includes a C / N arithmetic circuit that calculates a C / N ratio for each symbol signal for each symbol signal,
Subcarriers constituting one symbol signal include subcarriers that carry zero-padded dummy data and subcarriers that carry data other than the dummy data,
The C / N arithmetic circuit uses a signal value acquired from a subcarrier carrying the dummy data as noise power, a signal obtained from a subcarrier carrying data other than the dummy data as carrier power, An OFDM diversity receiving apparatus, wherein the C / N ratio is calculated based on a ratio between the noise power and the carrier power. In the OFDM diversity receiver according to claim 1,
The OFDM diversity receiver according to claim 1, wherein the carrier power is calculated from a signal acquired from a subcarrier carrying an SP signal among subcarriers carrying data other than the dummy data. In the OFDM diversity receiver according to claim 1 ,
The OFDM diversity receiver according to claim 1, wherein the carrier power is calculated from a signal acquired from a subcarrier carrying an SP signal or a CP signal among subcarriers carrying data other than the dummy data. The OFDM diversity receiver according to any one of claims 1 to 3 ,
The OFDM diversity receiver according to claim 1, wherein the noise power is calculated from a signal acquired from a subcarrier in a band not removed by a filter among subcarriers carrying the dummy data. The OFDM diversity receiver according to any one of claims 1 to 4,
The circuit for outputting the N first weighting values is:
By referring to a table in which the C / N ratio is associated with the first weight value in advance, the C / N ratio calculated using the C / N arithmetic circuit is converted into the first weight value. Lookup table to output,
An OFDM diversity receiver comprising: The OFDM diversity receiver according to any one of claims 1 to 5 ,
The circuit for outputting the N first weighting values calculates the first weighting value in each branch by an operation independent of signals received in other branches. . The OFDM diversity receiver according to any one of claims 1 to 6 ,
The synthesis circuit is:
A circuit that calculates N second weighted values by dividing the conjugate complex value of each weighted channel estimated value by the sum of squares of all N weighted channel estimated values;
A circuit for calculating N second weighted signals by multiplying each first weighted signal by each second weighted value;
A circuit for calculating a combined received signal by adding the N second weighted signals;
An OFDM diversity receiver comprising:

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