JP4364215B2 - Position detection device - Google Patents

Description

  The present invention relates to a position detection device that detects a moving position of an object by using a magnetic saturation characteristic of a magnetic material according to the proximity of a magnet, for example, a rotational position of a rotary or linear motor using a permanent magnet or an electromagnet. Alternatively, the present invention relates to a position detection device configured to detect a linear position.

2. Description of the Related Art Conventionally, a rotary differential transformer is well known as one that detects a rotational position of a magnetoresistive change member by using a magnetoresistive change member and converts the rotational position to a voltage level.
Similarly, a phase shift type rotational position detecting device is well known as a means for detecting the rotational position of a magnetoresistive variable member by using the magnetoresistive variable member and converting the rotational position into a phase signal. For example, those disclosed in JP-A-57-60212, JP-A-57-88317 and JP-B-62-58445 are known. Further, as phase shift type linear position detecting devices, those shown in Japanese Utility Model Laid-Open Nos. 57-13517, 58-136718, 59-175105, etc. are known. Yes.

  In the conventional position detection device using a change in magnetoresistance, two sets (first and second) of windings are wound on the stator side, so that a slip ring or a rotary transformer for extracting a detection signal from the rotor is provided. There is an advantage that it does not need to be provided. However, in order to obtain a change in magnetic resistance, the conventional position detection device as described above decenters the rotor, removes a part of the cylindrical shape to provide gear-like irregularities, or changes the shape of the rotor itself. A specially shaped magnetoresistive variable member is an essential component. Therefore, when detecting the rotational position of a rotary electric motor using such a position detector, the magnetoresistive change member of the position detector must be attached to the rotary shaft of the rotary electric motor via a coupling or the like. In other words, the axial length is increased by the axial length of the position detection device. In the case of a rotary motor, it is only necessary to attach a magnetoresistive change member to the rotary shaft. However, when this detects the linear position of the linear motor, the magnetoresistive change member is provided over the entire linear movement range. There must be. If the linear movement range is a short distance such as several centimeters, the magnetoresistive change member can be easily installed. However, if the linear movement range is several meters, it is very difficult to install the magnetoresistive change member over the movement range. It had the problem of being difficult.

The invention has been made in view of the above, the rotational position or linear position of the rotary or linear type motor with permanent magnets or electromagnets, the provision of such special magnetoresistance change member Therefore, an object of the present invention is to provide a position detecting device that can detect the position.

A position detection apparatus according to the present invention is a position detection apparatus in a linear or rotary electric motor having a movable part and a stationary part and having a magnet on the movable part or stationary part side. A detection unit provided to be relatively movable with respect to the magnet, including a winding means that is AC-excited and a magnetic core disposed in correspondence with the winding means, and the magnet for the magnetic core In response to the proximity of the magnet, magnetic saturation occurs at a location where the magnet core strongly receives the magnetizing force to lower the magnetic core magnetic permeability, and the magnetizing force associated with the relative movement of the magnet decreases. One that includes the detection unit that increases the magnetic permeability of the magnetic core by not causing magnetic saturation in response to a change, and does not provide the magnet among the movable unit or the stationary unit of the motor To the above The output unit is disposed, and a state where the relative positions of the detector and the magnet is changed in accordance with the displacement of the movable portion, the magnetizing force due to the magnet relative to the magnetic core in response thereto does not cause magnetic saturation Change in the state causing magnetic saturation, and the magnetic permeability of the magnetic core changes in accordance with the change, whereby the inductance of the winding means associated with the magnetic core changes. An output signal based on a change in inductance is generated from the winding means.

  Where magnetic saturation occurs in the magnetic core, the magnetic permeability of the magnetic core is reduced to the same level as air, and magnetic saturation occurs according to the change in the magnetizing force accompanying the relative movement of the magnet. The magnetic permeability of the magnetic core is increased by not being in the state. Thus, according to the relative displacement of the magnet with respect to the detection unit, the magnetization force of the magnet with respect to the magnetic core changes between a state in which magnetic saturation does not occur and a state in which magnetic saturation occurs. The permeability of the magnetic core gradually increases or decreases, and an AC output signal having an amplitude corresponding to the permeability, that is, the inductance, can be generated from the winding, thereby enabling position detection.

  In this case, as the magnet, a driving magnet group originally provided as a constituent element of a linear or rotary electric motor can be used, so that there is no need to provide a detection element and it is extremely simple. Furthermore, an electric motor with a position detection function can be provided. That is, linear motors, rotary motors, etc. are provided with electromagnets and permanent magnets in order to obtain driving force such as propulsive force and rotational force, and these magnets are arranged at a predetermined pitch. ing. Therefore, the present invention is extremely advantageous because an existing magnet group which is a component of such an electric motor can be used as the magnetoresistance change member.

  As an example, a magnet array composed of a plurality of magnets in which N poles and S poles are alternately arranged at a predetermined pitch is used. The magnetic flux is dense at the connection part of the magnet row, and the magnetic flux is sparse near the center of the magnet. Therefore, according to such a magnet array, the magnetic field distribution is shown so that the dense portion and the sparse portion of the magnetic flux, that is, the magnetizing force H are alternately repeated at a predetermined pitch. Since the magnetic saturation occurs at a location where the magnet core H, which is the strongest in the iron core, that is, the magnetic core, receives magnetic saturation, the permeability of the iron core, that is, the magnetic core is reduced to the same level as air. That is, since the magnetic flux passing through the iron core becomes dense due to the magnetic flux from the magnet, the iron core in that portion becomes magnetically saturated, as if it shows magnetic coupling (ie, permeability or inductance) in an air-core state where there is no iron core. Because it becomes. Therefore, in such a case, the induction output AC signal induced in the winding is the lowest. On the other hand, the magnetic flux generated by the magnet becomes relatively sparse in that portion as it moves away from the location where magnetic saturation occurs, so that magnetic saturation does not occur. The magnetic permeability of the iron core, that is, the magnetic core gradually increases (returns to the original iron), and the secondary electromotive force corresponding to that portion relatively increases. Thus, depending on the proximity of the magnet to the iron core, that is, the magnetic core, from a relatively high magnetic permeability corresponding to a portion where magnetic saturation does not occur to a low magnetic permeability corresponding to an air core corresponding to the portion where magnetic saturation occurs. The magnetic permeability, that is, the inductance gradually changes.

  As an example, the amplitude functions of the inductive output AC signals generated in the four secondary windings are sine function (sin), cosine function (cos), minus sine function (−sin), and minus cosine function (−cos), respectively. Correspondingly, iron core means, primary winding means and secondary winding means are constituted. The primary winding is excited with a predetermined AC signal, and amplitude modulation is performed in accordance with different amplitude function characteristics induced in the four winding portions in relation to the relative positional relationship between the iron core means and the magnet means. The relative position can be detected based on the induced output AC signal.

  When the relative position is detected based on the induction output AC signal from the secondary winding, the conventional one can be applied without providing a special position detection circuit.

Embodiments of the present invention will be described below with reference to the accompanying drawings.
FIG. 1 is a partial cross-sectional view showing a schematic configuration of a linear position detection apparatus according to an embodiment of a position detection apparatus according to the present invention. The linear position detection device 1 basically includes a cylindrical iron core 2 and a winding portion 3 wound around the core core 2 under a predetermined condition. The iron core 2 is silicon steel or the like having a large relative permeability and a small coercive force. The iron core 2 may be other than cylindrical silicon steel, or may be a rectangular parallelepiped iron core formed by laminating silicon steel plates, and may have any shape. The winding unit 3 includes a plurality of primary windings P1 to P5 excited by a predetermined AC signal and a plurality of secondary windings S1 to S4 wound in a predetermined direction X so as to have a predetermined positional relationship. It consists of.

  FIG. 2 is a diagram illustrating a connection state of the primary windings P1 to P5 and the secondary windings S1 to S4. As can be seen, the primary windings P1 to P5 may be any one that is commonly excited by a one-phase AC signal sin ωt (may be cos ωt), and the number of primary windings P1 to P5 is one or an appropriate number. There may be a plurality, and the arrangement may be appropriate. However, the primary windings P1 to P5 are appropriately separated and arranged so that the secondary windings S1 to S4 are sandwiched between the primary windings S1 to S4 as shown in FIG. Since the magnetic flux generated by the windings can be effectively applied to the individual secondary windings S1 to S4, it can be said to be a preferred embodiment. Also, the secondary winding S1 and the secondary winding S3, the secondary winding S2 and the secondary winding S4 are connected so as to operate differentially.

  The linear position detection device 1 cannot perform position detection only with the iron core 2 and the winding part 3. That is, when the mechanical system to be detected is a linear motor, the linear position detection device 1 is connected to the moving element side (not shown) of the linear motor and interlocks with the change in the linear position of the moving element. It is displaced linearly and reciprocally. In this embodiment, the moving element constituting the linear motor is configured to move on a rail 30 in which N poles and S poles of a plurality of magnets 31, 32,... As shown in FIG. Must have been. That is, when attention is paid to the magnet 31 and the magnet 32 constituting the rail 30, the magnetic flux is dense at the connection portion between them, and the magnetic flux is sparse near the center of each of the magnet 31 and the magnet 32. Therefore, dense and sparse magnetic flux portions are alternately arranged at a predetermined pitch P on the rail 30.

  If the pitch of the alternating arrangement of N poles and S poles of the magnets 31, 32,... Is P, the secondary windings S1 to S4 must be arranged as follows. That is, the secondary winding S3 is in a positional relationship (ie, S3 = S1 + (P / 2 + nP)) that is separated from the secondary winding S1 by an integral multiple of half the pitch P (P / 2). The secondary winding S2 has a positional relationship (ie, S2 = S1 + (P / 4 + nP)) that is separated from the secondary winding S1 by an integral multiple of a quarter of the pitch (P / 4). The wire S4 has a position relationship (ie, S3 = S1 + (P / 2 + nP)) that is separated from the secondary winding S2 by an integral multiple of half the pitch P (P / 2). Must be wound on.

  If the arrangement period pitch P of the magnets 31, 32,... On the rail 30 is 360 degrees, the secondary winding S3 is out of phase with the secondary winding S1 by 180 degrees in terms of electrical angle. The secondary winding S2 is out of phase with the secondary winding S1 by 90 degrees in electrical angle. The secondary winding S4 is out of phase with the secondary winding S2 by 180 degrees in electrical angle. means. When this relationship is represented by a trigonometric function, the secondary winding S1 is a sine function (sin), the secondary winding S3 is a minus sine function (-sin), the secondary winding S2 is a cosine function (cos), and the secondary winding. The line S4 becomes a minus cosine function (-cos).

Accordingly, the density of magnetic flux on the rail 30 alternately affects the iron core 2 and the winding portion 3 of the linear position detecting device 1 in accordance with the change in the linear position of the moving element, that is, the linear position detecting device 1 as the detection target. . That is, when the secondary winding is located in a portion where the magnetic flux is dense, the magnetic flux from the magnet passes densely in the iron core 2 in that portion, so the primary of the secondary winding located in the dense portion The magnetic coupling force to the winding is weakened. This causes magnetic saturation at a location where the magnetizing force H of the magnets 31, 32,... In the iron core 2, that is, the magnetic core is the strongest (the boundary between the N pole and the S pole). This is because the magnetic permeability of the core is reduced to the same extent as air. That is, since the magnetic flux passing through the iron core 2 becomes dense due to the magnetic flux from the magnet , the iron core 2 in that portion is in a magnetic saturation state, and it shows magnetic coupling (that is, magnetic permeability or inductance) in an air-core state where there is no iron core. Because it becomes like this. Therefore, when the secondary winding is located in a portion where the magnetic flux is dense, that is, the connection portion between the magnet 31 and the magnet 32, the secondary electromotive force is the lowest. On the other hand, as the distance from the point where magnetic saturation occurs, becomes a relatively sparse flux by the magnet at that part, since as not to cause magnetic saturation, the core 2 of the part, the magnetic flux, i.e. the magnetizing force from the magnet As H decreases, the magnetic permeability of the iron core 2, that is, the magnetic core gradually increases (returns to the original iron), and the secondary electromotive force corresponding to that portion relatively increases. Thus, depending on the proximity of the magnet to the iron core 2, that is, the magnetic core, a relatively low magnetic permeability corresponding to the air core corresponding to the portion where magnetic saturation has occurred, from a relatively high magnetic permeability corresponding to the portion where magnetic saturation does not occur. Until then, the magnetic permeability, that is, the inductance gradually changes.

  Therefore, the output AC signal induced in the secondary winding S <b> 1 changes according to the relative linear position relationship between the linear position detection device 1 and the rail 30. That is, the induction output AC signal that has been amplitude-modulated according to the relative linear position between the moving element to be detected and the rail 30 has different amplitude function characteristics depending on the displacement of the secondary windings S1 to S4. Are induced in the secondary windings S1 to S4. Each induction output AC signal induced in each secondary winding S1 to S4 has its electrical phase when the primary windings P1 to P5 are commonly excited by a one-phase AC signal sin ωt as shown in FIG. Are in-phase, and the amplitude function thereof periodically changes with the amount of displacement corresponding to the repetitive pitch P of the magnetic flux of the magnet of the rail 30 as one cycle.

  As described above, the four secondary windings S1 to S4 are arranged at predetermined intervals within the range of the repetitive pitch P of the rail 30, and the induction output AC signal generated in each secondary winding S1 to S4. The amplitude function has a desired characteristic. For example, when configured as a resolver type position detection device, the amplitude function of the induction output AC signal generated in the secondary windings S1 to S4 is a sine function (sin), a cosine function (cos), a minus sine function (−sin). ) And minus cosine function (-cos). In addition, since the arrangement of each winding can be changed slightly depending on various conditions, each winding arrangement is appropriately adjusted so as to obtain a desired function characteristic, or the secondary output level is adjusted by electrical amplification, A desired amplitude function characteristic may be finally obtained.

  For example, if the output of the secondary winding S1 corresponds to the sine function (sin), the output of the secondary winding S3 arranged so as to be shifted by P / 2 with respect to this corresponds to the minus sine function (−sin). The first output AC signal having a sine function amplitude function is obtained by differentially combining the outputs of both. Further, the output of the secondary winding S2 arranged with a deviation of P / 4 from the secondary winding S1 corresponding to the sine function output corresponds to the cosine function (cos) and arranged with a deviation of P / 2 with respect to this. The output of the secondary winding S4 thus made corresponds to a minus cosine function (-cos), and a second output AC signal having an amplitude function of the cosine function is obtained by differentially synthesizing both outputs. .

  FIG. 2 is a diagram showing a connection diagram of the winding section 3. A common excitation AC signal sin ωt is applied to the primary windings P1 to P5. In response to the excitation of the primary windings P1 to P5, an AC signal having an amplitude value corresponding to the relative position between the secondary windings S1 to S4 of the winding section 3 and the rail 30 is transferred to each secondary winding. It is guided to S1 to S4. Each induced voltage level indicates two-phase function characteristics sin θ and cos θ and its negative function characteristics −sin θ and −cos θ corresponding to the detection target linear position. That is, the induction output signals of the secondary windings S1 to S4 are amplitude-modulated with the two-phase function characteristics sin θ and cos θ and the negative function characteristics −sin θ and −cos θ corresponding to the detection target linear position. Each is output. Note that θ is proportional to x, for example, θ = 2π (X / P). For convenience of explanation, the coefficient according to other conditions such as the number of turns of the winding is omitted, the secondary winding S1 is a sine phase, the output signal is indicated as “sin θ * sin ωt”, and the secondary winding S2 is a cosine phase. The output signal is indicated by “cos θ * sin ωt”. Further, the output signal is indicated by “−sin θ * sin ωt” with the secondary winding S3 as a negative sine phase, and the output signal is indicated by “−cos θ * sin ωt” with the secondary winding S4 as a negative cosine phase. A first output AC signal (2 sin θ * sin ωt) having an amplitude function of a sine function is obtained by differentially combining the induction outputs of the sine phase and the minus sine phase. Further, a second output AC signal (2 cos θ * sin ωt) having an amplitude function of a cosine function is obtained by differentially combining the induction outputs of the cosine phase and the minus cosine phase. In order to simplify the expression, the coefficient “2” is omitted, and in the following, the first output AC signal is represented by “sin θ * sin ωt”, and the second output AC signal is represented by “cos θ * sin ωt”. Represent.

  Thus, the first output AC signal A = sin θ * sin ωt having the first function value sin θ corresponding to the detection target linear position x as an amplitude value and the second function value cos θ corresponding to the same detection target linear position x are obtained. A second output AC signal B = cos θ * sin ωt having an amplitude value is output. According to such a winding configuration, two output AC signals (sine output) having an in-phase AC and a two-phase amplitude function similar to those obtained in a conventionally known resolver which is a rotary position detecting device. And cosine output) can be obtained in the linear position detector. Therefore, the two-phase output AC signals (A = sin θ * sin ωt and B = cos θ * sin ωt) obtained in the linear position detection apparatus 1 of the present invention are handled in the same manner as a conventionally known resolver output processing method. Can do.

  In addition, as described above, the configuration in which the four secondary windings S1 to S4 are arranged at a predetermined interval within the range of the repetitive pitch P of the magnets of the rail 30 is that the size of the entire winding portion 3 is one piece of magnet. Can be accommodated in a relatively small size substantially corresponding to the above range, which helps to reduce the overall configuration of the linear position detecting device.

  As described above, according to the linear position detection device 1 according to the present invention, although it is a linear type position detection device, the two-phase output AC signals (A = sin θ * sin ωt and B = cos θ * are the same as those of the rotary resolver. sinωt) can be output from the secondary windings S1 to S4 of the winding section 3. Therefore, by applying an appropriate digital phase detection circuit, the phase value θ of the sine function sin θ and the cosine function cos θ can be detected by digital phase detection, and position detection data of the linear position x can be obtained based on this.

  For example, FIG. 3 shows an example in which a known RD (resolver-digital) converter is applied. Resolver type two-phase output AC signals A = sin θ * sin ωt and B = cos θ * sin ωt output from the secondary windings S1 to S4 of the winding unit 3 are input to the analog multipliers 30 and 31, respectively. The sequential phase generation circuit 32 generates digital data with a phase angle φ and outputs it to a sine / cosine generation circuit 33. The sine / cosine generation circuit 33 generates an analog signal of a sine value sinφ and a cosine value cosφ corresponding to the phase angle φ, and outputs it to the multipliers 30 and 31. The multiplier 30 multiplies the sine-phase output AC signal A = sin θ * sin ωt by the cosine value cos φ from the sine / cosine generation circuit 33 to generate “cos φ * sin θ * sin ωt”. The other multiplier 31 multiplies the output AC signal B = cosθ * sinωt of the cosine phase by the sine value sinφ from the sine / cosine generation circuit 33 to generate “sinφ * cosθ * sinωt”. The subtractor 34 obtains the difference between the output signals from both the multipliers 30 and 31 and sequentially outputs it to the phase generation circuit 32.

The sequential phase generation circuit 32 controls the phase generation operation as follows according to the output from the subtractor 34. That is, the sequential phase generating circuit 32 resets the generated phase angle φ to 0 at first, and then increases sequentially, and stops increasing when the output of the subtractor 34 becomes 0. The output of the subtractor 34 becomes 0 when “cos φ * sin θ * sin ωt” = “sin φ * cos θ * sin ωt” is satisfied, that is, when φ = θ is satisfied. At this time, the digital data of the phase angle φ sequentially output from the phase shift generation circuit 32 and the digital value of the phase angle θ of the amplitude function of the output AC signals A and B coincide with each other. Accordingly, a reset trigger is periodically applied at an arbitrary timing to sequentially reset the phase angle φ generated by the phase generation circuit 32 to 0, and then the increment of the phase angle φ is started, and the output of the subtractor 34 is 0. At that time, the increment process is stopped and digital data of the phase angle θ is obtained.
It is known that the sequential phase generating circuit 32 includes an up / down counter and a VCO, and the VCO is driven by the output of the subtractor 34 to control the up / down counting operation of the up / down counter. In that case, a periodic reset trigger is not necessary.

An error occurs in the electrical AC phase ωt in the secondary output AC signal due to changes in the impedances of the primary and secondary windings of the winding section 3 due to temperature changes or the like. In the phase detection circuit as shown in FIG. , Sinωt phase errors are automatically canceled out, which is desirable. On the other hand, in a method in which an electrical phase shift is generated in a one-phase output AC signal by exciting with a conventionally known two-phase AC signal (for example, sin ωt and cos ωt), such a temperature change is caused. The output phase error based on it cannot be removed.
By the way, since the phase detection circuit comprising the conventional RD converter as described above is a follow-up comparison method, there is a problem in terms of response performance due to a clock delay when the follow-up counting is performed.
Therefore, the present inventors have developed a novel phase detection circuit as described below, and it is desirable to use this.

FIG. 4 is a diagram showing an embodiment of a novel phase detection circuit applied to the linear position detection apparatus according to the present invention.
In FIG. 4, in the detection circuit unit 41, a counter 42 counts a predetermined high-speed clock pulse CK, and an excitation AC signal (sin ωt) is generated from the excitation signal generation circuit 43 based on the count value. To the primary windings P1 to P5. The modulo number of the counter 42 corresponds to one cycle of the excitation AC signal, and for convenience of explanation, the count value of 0 corresponds to the 0 phase of the reference sine signal sin ωt. For example, assuming that one cycle from the 0 phase to the maximum phase of the reference sine signal sin ωt is generated while the count value of the counter 42 makes one round from 0 to the maximum value, the same as the reference sine signal sin ωt. An AC signal sin ωt for excitation in phase is generated from the excitation signal generation circuit 43. Two-phase output AC signals A = sin θ * sin ωt and B = cos θ * sin ωt output from the secondary windings S 1 to S 4 of the winding unit 3 are input to the detection circuit unit 41.

  In the detection circuit unit 41, the phase shift circuit 44 receives the first AC output signal A = sin θ * sin ωt, shifts the electrical phase by a predetermined amount, advances, for example, 90 degrees, and the phase-shifted AC signal AS = sin θ * cos ωt is output. The detection circuit unit 41 includes an addition circuit 45 and a subtraction circuit 46. The addition circuit 45 outputs the phase-shifted AC signal AS = sin θ * cos ωt output from the phase shift circuit 44 and a winding. The second AC output signal B = cos θ * sin ωt output from the secondary windings S1 to S4 of the unit 3 is added, and the addition result B + AS = cos θ * sin ωt + sin θ * cos ωt = sin (ωt + θ) The electrical AC signal Y1 is output. The subtracting circuit 46 subtracts the phase-shifted AC signal AS = sin θ * cos ωt from the second AC output signal B = cos θ * sin ωt, and B-AS = cos θ * sin ωt−sin θ * cos ωt which is the subtraction result. A second electrical AC signal Y2 of = sin (ωt−θ) is output. In this way, the same detection target position as the first electrical AC signal Y1 = sin (ωt + θ) having the electrical phase angle (+ θ) shifted in the positive direction corresponding to the detection target position (x). A second electrical AC signal Y2 = sin (ωt−θ) having an electrical phase angle (−θ) shifted in the negative direction corresponding to (x) is obtained by electrical processing.

  The output signals Y1 and Y2 of the adder circuit 45 and the subtractor circuit 46 are input to the corresponding zero cross detection circuits 47 and 48, respectively, where the zero crosses of the respective output signals Y1 and Y2 are detected. As a method of detecting the zero cross, for example, a zero cross point where the amplitude values of the signals Y1 and Y2 change from negative to positive may be detected. The zero cross detection pulses detected by the zero cross detection circuits 47 and 48 are output to the latch circuits 49 and 50 as latch pulses LP1 and LP2. The latch circuits 49 and 50 latch the count value from the counter 42 at the timing of the respective latch pulses LP1 and LP2. As described above, the modulo number of the counter 42 corresponds to one cycle of the excitation AC signal, and the count value 0 corresponds to the 0 phase of the reference sine signal sinωt. The data D1 and D2 latched in the latch circuits 49 and 50 correspond to the phase shifts of the output signals Y1 and Y2 with respect to the reference sine signal sin ωt, respectively. The outputs of the latch circuits 49 and 50 are input to the error calculation circuit 51, where “(D1 + D2) / 2” is calculated and the phase variation error ± d is calculated. This calculation is actually performed by shifting the addition result of the binary data “D1 + D2” to the lower one bit.

Here, the influence of the length of the wiring cable between the winding part 3 and the detection circuit part 41 and the impedance change due to the temperature change or the like have occurred in each primary and secondary winding of the winding part 3. In consideration of the above, when the phase fluctuation error of the output signal is represented by “± d”, each signal in the detection circuit unit 41 is represented as follows.
A = sin θ * sin (ωt ± d)
AS = sin θ * cos (ωt ± d)
B = cos θ * sin (ωt ± d)
Y1 = sin (ωt ± d + θ)
Y2 = sin (ωt ± d−θ)
D1 = ± d + θ
D2 = ± d−θ

That is, each of the phase shift measurement data D1 and D2 performs a phase shift count using the reference sine signal sin ωt as a reference phase, and thus a value including the phase variation error “± d” is obtained as described above. . Therefore, by calculating “(D1 + D2) / 2” in the error calculation circuit 51,
(D1 + D2) / 2 = {(± d + θ) + (± d−θ)} / 2
= ± 2d / 2 = ± d
Thus, the phase variation error “± d” can be calculated.
The data of the phase fluctuation error “± d” obtained by the error calculation circuit 51 is given to the subtraction circuit 52, and is subtracted from one phase shift measurement data D1. That is, in the subtraction circuit 52, “D1− (± d)” is subtracted.
D1− (± d) = ± d + θ− (± d) = θ
Thus, digital data indicating the correct detected phase difference θ from which the phase fluctuation error “± d” has been removed is obtained. As described above, according to the phase detection circuit of FIG. 4, it can be understood that the phase fluctuation error “± d” is canceled and only the correct phase difference θ corresponding to the detection target position x is extracted.

This point will be further described with reference to FIG. FIG. 5 shows a sine signal sin ωt serving as a reference for phase measurement and waveforms near the zero phase of the first and second AC signals Y1 and Y2, and FIG. 5A shows a positive phase fluctuation error ( In the case of + d), FIG. 5B shows the case of minus (−d). In the case of FIG. 5A, the zero phase of the first signal Y1 advances by “θ + d” with respect to the zero phase of the reference sine signal sin ωt, and the corresponding phase difference detection data D1 becomes “θ + d”. The corresponding phase difference is shown. Further, the zero phase of the second signal Y2 is delayed by “−θ + d” with respect to the zero phase of the reference sine signal sin ωt, and the corresponding phase difference detection data D2 is a phase difference corresponding to “−θ + d”. Indicates. In this case, the error calculation circuit 51
(D1 + D2) / 2 = {(+ d + θ) + (+ d−θ)} / 2
= + 2d / 2 = + d
Thus, the phase fluctuation error “+ d” is calculated. Then, by the subtraction circuit 52,
D1 − (+ d) = + d + θ − (+ d) = θ
Is calculated, and the correct phase difference θ is extracted.

In the case of FIG. 5B, the zero phase of the first signal Y1 is advanced by “θ-d” with respect to the zero phase of the reference sine signal sin ωt, and the corresponding phase difference detection data D1 is “θ -D "represents the phase difference. Further, the 0 phase of the second signal Y2 is delayed by “−θ−d” with respect to the 0 phase of the reference sine signal sin ωt, and the corresponding phase difference detection data D2 becomes “−θ−d”. The corresponding phase difference is shown. In this case, the error calculation circuit 51
(D1 + D2) / 2 = {(− d + θ) + (− d−θ)} / 2
= -2d / 2 = -d
Thus, the phase fluctuation error “−d” is calculated. Then, by the subtraction circuit 52,
D1 − (− d) = − d + θ − (− d) = θ
Is calculated, and the correct phase difference θ is extracted.
The subtracting circuit 52 may subtract “D2− (± d)”. In principle, data (−θ) reflecting the correct phase difference θ is obtained in the same manner as described above.

  Further, as can be understood from FIG. 5, the electrical phase difference between the first signal Y1 and the second signal Y2 is 2θ, which is always accurate by canceling out the phase variation error “± d” between the two. This indicates a value twice as large as the phase difference θ. Therefore, even if the configuration of the circuit portion including the latch circuits 49 and 50, the error calculation circuit 51, the subtraction circuit 52, and the like in FIG. 4 is changed to a configuration that directly obtains the electrical phase difference 2θ of the signals Y1 and Y2. Good. For example, from the generation time point of the pulse LP1 corresponding to the 0 phase of the first signal Y1 output from the zero cross detection circuit 47, the pulse LP2 corresponding to the 0 phase of the second signal Y2 output from the zero cross detection circuit 48 is generated. Digital data corresponding to the electrical phase difference (2θ) that offsets the phase fluctuation error “± d” can be obtained by gating the period up to the time of occurrence by appropriate means and counting the gate period. Therefore, if this is shifted one bit lower, data corresponding to θ can be obtained.

  By the way, in this embodiment, the latch circuit 49 for latching + θ and the latch circuit 50 for latching −θ latch the output of the same counter 42, and the positive and negative of the latched data is obtained. No special reference is made to the code. However, an appropriate design process may be applied to the positive and negative signs of the data in accordance with the spirit of the present invention. For example, assuming that the modulo number of the counter 42 is 4096 (decimal number display), the digital counts 0 to 4095 may be appropriately processed according to the phase angle of 0 degrees to 360 degrees. In the simplest design example, the most significant bit of the count output of the counter 42 is a sign bit, the digital counts 0 to 2047 correspond to +0 degrees to +180 degrees, and the digital counts 2048 to 4095 are set to −180 degrees to −0 degrees. Correspondingly, arithmetic processing may be performed. Alternatively, as another example, by converting the input data or output data of the latch circuit 50 into a two's complement, the digital count 4095-0 can correspond to a negative angle data expression of -360 degrees to -0 degrees. May be.

Incidentally, there is no particular problem when the detection target position x is in a stationary state, but when the detection target position x changes with time, the corresponding phase angle θ also changes with time. In this case, the phase shift amount θ of each of the output signals Y1 and Y2 of the addition circuit 45 and the subtraction circuit 46 is not a constant value but shows a dynamic characteristic that changes with time according to the moving speed. t), each output signal Y1, Y2 is
Y1 = sin {ωt ± d + θ (t)}
Y2 = sin {ωt ± d−θ (t)}
It becomes. That is, with respect to the frequency of the reference signal sin ωt, the fast-phase output signal Y1 transitions in a frequency increasing direction according to + θ (t), and the slow-phase output signal Y2 according to −θ (t). The frequency transitions in the direction of decreasing frequency. Under such dynamic characteristics, the period of each of the signals Y1, Y2 successively changes in the opposite direction for each period of the reference signal sin ωt, so that the latch data D1, The measurement time reference for D2 is different, and an accurate phase variation error “± d” cannot be obtained by simply calculating both data D1 and D2 by the circuits 51 and 52.

The simplest method for avoiding such a problem is that in the configuration of FIG. 4, the output when the detection target position x is moving in time is ignored, and only the output in the stationary state is used. The function of the apparatus is limited so as to measure the detection target position x at rest. That is, the present invention may be implemented for such a limited purpose.
However, it is desirable to be able to accurately detect the phase difference θ corresponding to the detection target linear position x every moment even when the detection target position x is changing with time. Therefore, in order to solve the above problems, the phase difference θ corresponding to the detection target position x can be detected every moment even when the detection target linear position x is changing in time. The improved measures will be described with reference to FIG.

FIG. 6 shows an extracted example of a change in the error calculation circuit 51 and the subtraction circuit 52 in the detection circuit unit 41 of FIG. 4, and the configuration of other parts not shown is the same as FIG. Good. When the phase difference θ corresponding to the position x when the detection target linear position x changes with time is represented by + θ (t) and −θ (t), the output signals Y1 and Y2 are as described above. I can express. The phase shift measured value data D1 and D2 obtained by the latch circuits 49 and 50 corresponding to the
D1 = ± d + θ (t)
D2 = ± d−θ (t)
It becomes.
In this case, ± d + θ (t) repeatedly changes in time in the positive direction in the range of 0 ° to 360 ° in accordance with the time change of θ. In addition, ± d−θ (t) repeatedly changes in time in the minus direction in the range of 360 ° to 0 ° in accordance with the time change of θ. Therefore, there are cases where ± d + θ (t) ≠ ± d−θ (t), but there are also cases where the changes of both intersect, and in this case, ± d + θ (t) = ± d−θ (t) holds. . Thus, when ± d + θ (t) = ± d−θ (t) is satisfied, the electrical phases of the output signals Y1 and Y2 coincide with each other, and the latch corresponding to the respective zero-cross detection timings. The generation timings of the pulses LP1 and LP2 are the same.

In FIG. 6, the coincidence detection circuit 53 detects that the generation timings of the latch pulses LP1 and LP2 corresponding to the zero-cross detection timings of the output signals Y1 and Y2 coincide with each other, and in response to this detection, the coincidence detection pulse Generate an EQP. On the other hand, the time variation determination circuit 54 is a mode in which the detection target position x changes with time by an appropriate means (for example, by detecting the presence or absence of temporal change in the value of one phase difference measurement data D1). And the time variation mode signal TM is output in accordance with this determination.
A selector 55 is provided between the error calculation circuit 51 and the subtraction circuit 52. When the time variation mode signal TM is not generated, that is, TM = “0”, that is, the detection target linear position x changes with time. If not, the selector 55 selects the signal from the error calculation circuit 51 applied to the input terminal B and outputs it to the subtraction circuit 52. Thus, the circuit of FIG. 6 when the input terminal B of the selector 55 is selected operates equivalently to the circuit of FIG. That is, when the detection target linear position x is stationary, the output data of the error calculation circuit 51 is directly given to the subtraction circuit 52 via the input terminal B of the selector 55, and operates in the same manner as the circuit of FIG. To do.

  On the other hand, when the time variation mode signal TM is generated, that is, when TM = “1”, that is, when the detection target position x is temporally changing, the selector 55 is a latch circuit applied to the input terminal A. The signal from 56 is selected and output to the subtraction circuit 52. When the time variation mode signal TM is “1” and the coincidence detection pulse EQP is generated, the condition of the AND gate 57 is satisfied, and a pulse responding to the coincidence detection pulse EQP is output from the AND gate 57, and the latch circuit A latch instruction is given to 56. The latch circuit 56 latches the output count data of the counter 42 in response to the latch instruction. Here, when the coincidence detection pulse EQP is generated, the output of the counter 42 is simultaneously latched in the latch circuits 49 and 50, so that D1 = D2, and the data latched in the latch circuit 56 is D1 or D2 ( However, this corresponds to D1 = D2).

The coincidence detection pulse EQP is generated when the zero cross detection timings of the output signals Y1 and Y2 coincide, that is, when “± d + θ (t) = ± d−θ (t)” is established. Since the data latched in the latch circuit 56 in response to D1 corresponds to D1 or D2 (where D1 = D2),
(D1 + D2) / 2
Is equivalent to This means
(D1 + D2) / 2 = (± d + θ (t) + ± d−θ (t)) / 2
= 2 (± d) / 2 = ± d
This means that the data latched by the latch circuit 56 accurately indicates the phase fluctuation error “± d”.

Thus, when the detection target straight line position x fluctuates with time, data accurately indicating the phase fluctuation error “± d” is latched by the latch circuit 56 in accordance with the coincidence detection pulse EQP. The output data is given to the subtraction circuit 52 via the input A of the selector 55. Therefore, the subtraction circuit 52 can obtain data θ (or θ (t) in the case of temporal variation) that accurately responds only to the detection target position x from which the phase variation error “± d” has been removed.
In FIG. 6, the AND gate 57 may be omitted, and the coincidence detection pulse EQP may be directly applied to the latch control input of the latch circuit 56.
Further, the latch circuit 56 may latch not only the output count data of the counter 42 but also the output data “± d” of the error calculation circuit 51 as indicated by a broken line in FIG. In this case, the output timing of the output data of the error calculation circuit 51 corresponding to the generation timing of the coincidence detection pulse EQP is somewhat delayed according to the circuit operation delay of the latch circuits 49 and 50 and the error calculation circuit 51. The output of the error calculation circuit 51 may be latched by the latch circuit 56 after performing an appropriate time delay adjustment.
Further, when the detection circuit unit 41 is configured in consideration of only dynamic characteristics, the circuit 51 and selector 55 in FIG. 6 and one latch circuit 49 or 50 in FIG. 1 may be omitted.

FIG. 7 shows another embodiment of the phase difference detection calculation method that can cancel the phase fluctuation error “± d”.
The resolver type first and second AC output signals A and B output from the secondary windings S <b> 1 to S <b> 4 of the winding unit 3 are input to the phase shift circuit 44 of the detection circuit unit 60. As in the case of FIG. 4, the phase shift circuit 44 phase-shifts the first AC output signal A = sin θ * sin ωt by a predetermined amount by its electrical phase, and outputs the phase-shifted AC signal AS = sin θ * cos ωt. Output. The subtracting circuit 46 subtracts the phase-shifted AC signal AS = sin θ * cos ωt from the second AC output signal B = cos θ * sin ωt, and B-AS = cos θ * sin ωt−sin θ * cos ωt, which is the subtraction result. = Electric AC signal Y2 of sin (ωt−θ) is output. The zero cross detection circuit 48 detects the zero cross of the output signal Y2 from the subtraction circuit 46, and outputs a latch pulse LP2 corresponding to the zero cross to the latch circuit 50.

  The embodiment of FIG. 7 differs from the embodiment of FIG. 4 in that the phase shift amount θ is measured from the AC signal Y2 = sin (ωt−θ) including the electrical phase shift corresponding to the detection target position. The reference phase at the time is different. In the example of FIG. 4, the reference phase when measuring the phase shift amount θ is the zero phase of the reference sine signal sin ωt, which is not input to the position sensor 10, so This does not include the phase variation error “± d” based on the line impedance change and other various factors. Therefore, in the example of FIG. 4, two AC signals Y1 = sin (ωt + θ) and Y2 = sin (ωt−θ) are formed, and the phase difference error “± d” is calculated by obtaining the electrical phase difference. I try to offset it. On the other hand, in the embodiment of FIG. 7, the reference phase when measuring the phase shift amount θ is calculated based on the first and second AC output signals A and B output from the winding unit 3. Since the phase variation error “± d” is included in the reference phase itself, the phase variation error “± d” can be eliminated.

That is, in the detection circuit unit 60, the zero cross detection circuits 61 and 62 respectively receive the first and second AC output signals A and B from the winding unit 3 of the position detection device 1 and detect the respective zero crosses. . Note that the zero cross detection circuits 61 and 62 respond to both the zero cross (so-called 0 phase) in which the amplitude values of the input signals A and B change from negative to positive and the zero cross (so-called 180 degree phase) in which the amplitude changes from positive to negative. A zero cross detection pulse is output. This is because sin θ and cos θ that determine the positive / negative polarity of the amplitudes of the signals A and B are arbitrarily positive or negative depending on the value of θ. Therefore, when detecting a zero cross every 360 degrees based on the combination of both, This is because it is necessary to detect a zero cross every 180 degrees.
The OR circuit 63 outputs a logical sum signal of the zero-cross detection pulses from the zero-cross detection circuits 61 and 62 to the half-divided pulse circuit 64. The half-divided pulse circuit 64 receives the logical sum signal from the OR circuit 63 and outputs a zero-cross detection pulse to the counter 65 as a reference phase signal pulse PR every other one. As a result, the reference phase signal pulse PR output from the 1/2 frequency divider circuit 64 becomes a zero cross every 360 degrees, that is, a zero cross detection pulse corresponding to only the zero phase. The ½ divider circuit 64 includes a ½ divider circuit such as a T-type flip-flop and a pulse output AND gate. The reference phase signal pulse RP is output to the reset input terminal of the counter 65. The counter 65 continuously counts a predetermined clock pulse CK, but is reset according to the input of the reference phase signal pulse RP. The output of the counter 65 is input to the latch circuit 50 and is latched by the latch circuit 50 at the generation timing of the latch pulse LP2. The data D latched by the latch circuit 50 is output as measurement data of the phase difference θ corresponding to the detection target position x.

  The first and second AC output signals A and B output from the winding unit 3 are A = sin θ * sin ωt and B = cos θ * sin ωt, respectively, and the electrical phases are the same. Therefore, zero crossing should be detected at the same timing, but since the amplitude coefficient fluctuates by sine sin θ and cosine cos θ, either amplitude level may be 0 or close to 0. On the other hand, virtually no zero crossing can be detected. In this embodiment, therefore, zero cross detection processing is performed for each of the two AC output signals A = sin θ * sin ωt and B = cos θ * sin ωt, and either of the zero cross detection outputs is OR-synthesized, so that one of the amplitudes is amplitude. Even if the zero cross detection is impossible due to the low level, the zero cross detection output signal having the larger amplitude level on the other side can be used.

  In the case of the embodiment of FIG. 7, assuming that the phase fluctuation error due to the winding impedance change of the winding section 3 is “−d”, for example, the AC signal Y2 output from the subtracting circuit 46 is as shown in FIG. As shown in A), Y2 = sin (ωt−d−θ). In this case, the output signals A and B of the winding section 3 have amplitude values sin θ and cos θ corresponding to the angle θ, respectively, and A = sin θ * sin (ωt−d) as illustrated in FIG. ), B = cos θ * sin (ωt−d), and so on. Therefore, the reference phase signal pulse RP obtained at the timing as shown in FIG. 8C based on the zero cross detection is shifted from the zero phase of the original reference sine signal sin ωt by the phase variation error −d. Accordingly, if the phase shift amount of the output AC signal Y2 = sin (ωt−d−θ) of the subtracting circuit 46 is measured with reference to the reference phase signal pulse RP, an accurate value θ with the phase fluctuation error −d removed. Is obtained.

  In addition, if apparatus conditions, such as the wiring length of the coil | winding part 3, are decided, the impedance change will mainly depend on temperature. Then, the phase fluctuation error ± d corresponds to data indicating the temperature of the surrounding environment where the linear position detection device is provided. Therefore, in the circuit having the circuit 51 for calculating the phase fluctuation error ± d as in the embodiment of FIG. 4, the data of the phase fluctuation error ± d obtained there can be appropriately output as temperature detection data. Therefore, according to such a configuration of the present invention, not only the position of the detection target can be detected by one position detection device, but also data indicating the temperature of the surrounding environment can be obtained. It has an effect and can provide an unprecedented versatile type sensor. Of course, there is also an excellent effect that high-accuracy detection in response to the detection target position is possible without being affected by the impedance change on the sensor side due to temperature change or the length of the wiring cable. Further, since the embodiments of FIGS. 4 and 7 are methods for measuring the phase difference in the AC signal, detection with excellent high-speed response can be performed compared to the detection method as shown in FIG. , Has an excellent effect.

Hereinafter, various modifications of the position detection apparatus of FIG. 1 will be described.
FIG. 9 is a diagram showing a first modification of the position detection device of the present invention.
In the modification of FIG. 9, the arrangement of the four secondary windings S1 to S4 is the same as that of FIG. 1 in that the range of one pitch P is divided into four, and the difference is the primary winding. The point is that the wire is wound on both sides of the secondary windings S1 to S4. That is, the primary windings P1A and P1B are disposed on both sides of the secondary winding S1, the primary windings P2A and P2B are disposed on both sides of the secondary winding S2, and the primary windings are disposed on both sides of the secondary winding S3. Lines P3A and P3B are wound with primary windings P4A and P4B on both sides of the secondary winding S4, respectively. In addition, although the connection method of each primary winding and secondary winding may be the same as that in the case of FIG. 2, the connection method as shown in FIG. 10 is adopted in the embodiment of FIG.
That is, the primary windings P1A and P1B on both sides of the secondary winding S1 and the primary windings P3A and P3B on both sides of the secondary winding S3 are differentially wound with each other. The AC signal sinωt is commonly excited. The primary windings P2A and P2B on both sides of the secondary winding S2 and the primary windings P4A and P4B on both sides of the secondary winding S4 are differentially wound with each other. Are commonly excited by an AC signal cosωt. The secondary windings S1 to S4 are connected in series, and the combined output is extracted as an output signal Y = Ksin (ωt−θ). Instead of differentially winding the primary winding, as shown in FIG. 11, the secondary winding S1 and the secondary winding S3, the secondary winding S2 and the secondary winding S4 are respectively differentially wound. It may be wound.

FIG. 11 is a diagram showing an embodiment of a position converter 70 that detects a phase difference from the output signal Y = Ksin (ωt−θ) output from the position detector connected as shown in FIG. In FIG. 11, the position converter 70 detects a phase difference (phase shift amount) Dθ between the reference AC signal sinωt and the output signal Y, and a reference signal generator that generates the reference AC signals ia = Isinωt and ib = Icosωt. And a phase difference detection unit.
The reference signal generation unit includes a clock oscillator 71, a synchronization counter 72, ROMs 73a and 73b, D / A converters 74a and 74b, and amplifiers 75a and 75b. A phase difference detection unit includes an amplifier 76, a zero cross circuit 77, and a latch circuit 78. Composed.
The clock oscillator 71 generates a high-speed accurate clock signal, and other circuits operate based on this clock signal.
The synchronization counter 72 counts the clock signal output from the clock oscillator 71, and outputs the count value as an address signal to the ROMs 73a and 73b and the latch circuit 78 of the phase difference detector.
The ROMs 73a and 73b store amplitude data corresponding to the reference AC signal, and generate amplitude data of the reference AC signal according to the address signal (count value) from the synchronization counter 72. The ROM 73a stores sin ωt amplitude data, and the ROM 73b stores cos ωt amplitude data. Therefore, the ROMs 73a and 73b receive the same address signal from the synchronous counter 72, and output two types of reference AC signals sinωt and cosωt. It should be noted that two types of reference AC signals can be obtained in the same manner even when ROMs having the same amplitude data are read out with address signals having different phases.

The D / A converters 74a and 74b convert the digital amplitude data from the ROMs 73a and 73b into analog signals and output them to the amplifiers 75a and 75b. The amplifiers 75a and 75b amplify the analog signals from the D / A converters 74a and 74b and apply them to the primary windings P1A to P4B as reference AC signals Isinωt and Icosωt. When the frequency division number of the synchronous counter 72 is M, the M count corresponds to the maximum phase angle 2π radians (360 degrees) of the reference AC signal. That is, one count value of the synchronization counter 72 indicates a phase angle of 2π / M radians.
The amplifier 77 amplifies the combined output AC signal Y = Ksinωt of the outputs from the secondary windings S <b> 1 to S <b> 4 and outputs the amplified signal to the zero cross circuit 78.
The zero cross circuit 78 detects the zero cross point from the negative voltage to the positive voltage based on the output AC signal Y from the amplifier 77, and outputs the detection signal to the latch circuit 78.
The latch circuit 78 latches the count value of the synchronous counter started by the clock signal at the rising edge of the reference AC signal at the output time (zero cross point) of the detection signal of the zero cross circuit 77. Therefore, the value latched by the latch circuit 78 is exactly the phase difference (phase shift amount) Dx between the reference AC signal and the output signal Y. This phase difference Dx becomes position data within one pitch P of the rail 30 described above.

That is, the output signal Y = sin (ωt−θ) from the amplifier 76 is given to the zero cross circuit 77. The zero cross circuit 77 outputs a pulse L in synchronization with the timing when the electrical phase angle of the output signal Y is zero. The pulse L is used as a latch pulse of the latch circuit 78. Accordingly, the latch circuit 78 latches the count value of the synchronous counter 72 in response to the rise of the pulse L. A period in which the count value of the synchronization counter 72 makes a round is synchronized with one cycle of the sine wave signal Isinωt. Then, the count value corresponding to the phase difference θ between the reference AC signal Isinωt and the combined output signal Y = Isin (ωt−θ) is latched in the latch circuit 78. Therefore, the latched value is output as digital position data Dx. Note that the latch pulse L can also be used as a timing pulse as appropriate.
Note that the connection shown in FIG. 10 or FIG. 11 may be applied to the position detection apparatus of FIG.

FIG. 12 is a diagram showing a second modification of the position detection device of the present invention. 12 is the same as FIG. 1 in that the four secondary windings S1 to S4 are arranged at positions obtained by dividing the range of one pitch P into four, and the difference is that the primary winding P6 is different. The secondary winding S1 to S4 are integrally wound over the entire outer peripheral side. In this case, the connection shown in FIG. 2 is applicable. The primary winding may be interchanged with the secondary winding, and the secondary winding may be interchanged with the outside.
FIG. 13 is a diagram showing a third modification of the position detection device of the present invention. In the modification of FIG. 13, the primary windings P1 to P4 are wound along the outer periphery of the corresponding secondary windings S1 to S4. In this case, the connection shown in FIG. 2, FIG. 10, or FIG. 11 is applicable. The primary winding may be interchanged with the secondary winding, and the secondary winding may be interchanged with the outside.
FIG. 14 is a diagram showing a fourth modification of the position detection device of the present invention. In the modification of FIG. 14, the primary windings P1 to P4 and the secondary windings S1 to S4 are mixed at the same time without separating the primary winding and the secondary winding as in the above-described embodiment. It is wound around the iron core 2. Note that the primary windings P1 to P4 and the secondary windings S1 to S4 are arranged at positions obtained by dividing the range of the 1 pitch P into four, as described above. In this case, the connection shown in FIG. 2, FIG. 10, or FIG. 11 is applicable.

The above-described position detection device 1 can detect the linear position without any problem when the moving element is stationary. However, when the moving element is moving linearly, the secondary winding S1 is accompanied by the movement. When S4 crosses the magnetic flux of each magnet, an induced electromotive force corresponding to the moving speed is generated. Since this induced electromotive force becomes an error in position detection processing at the time of movement, it is necessary to remove the influence.
Hereinafter, a modified example of the position detection device in which the influence of the induced electromotive force is removed so that the position can be accurately detected even during movement will be described.
FIG. 15 is a diagram showing a fifth modification of the position detection device of the present invention. In FIG. 15, dummy iron cores 2D having the same shape are provided via connecting members 80 and 81 provided at both ends of the iron core 2 of the position detection device 1 of FIG. 1. Dummy windings D1 to D4 are wound around the dummy iron core 2D so as to have the same shape (same winding), the same arrangement, and the same arrangement as the secondary windings S1 to S4. The dummy windings D1 to D4 are differentially connected to the corresponding secondary windings S1 to S4.
By providing the dummy windings D1 to D4 as shown in FIG. 15, the influence of the induced electromotive force generated according to the moving speed can be eliminated.
Note that the position detection device shown in FIG. 15 may be the one shown in FIG.

FIG. 16 is a diagram showing a sixth modification of the position detection device of the present invention. The position detection device shown in FIG. 16 is a modification of the position detection device of FIG. The position detection device of FIG. 16 differs from that of FIG. 15 in that a primary winding wound on both sides of the secondary winding as shown in FIG. 9 is used as the position detection device. S1 and the secondary winding S3 are wound around the iron core 21, and the secondary winding S2 and the secondary winding S4 are wound around the iron core 22, respectively. Thus, by winding the secondary winding around different iron cores, magnetic interference between the two can be eliminated, so that position detection with less error can be performed.
FIG. 17 is a diagram showing a seventh modification of the position detection device of the present invention. The position detection device of FIG. 17 is a modification of that of FIG. 17 differs from that of FIG. 16 in that the secondary windings S1 to S4 are wound around separate iron cores 21 to 24, respectively, and the dummy windings D1 to D4 are also wound around the iron cores 21 to 24, respectively. It is a point that is wound separately. It goes without saying that only the iron core 23 for dummy winding as shown in FIG. 16 may be provided separately.
The arrangement patterns shown in FIGS. 15 to 17 are merely examples, and it goes without saying that various arrangements other than this may be performed.
Further, the position detection may be averaged by arranging a plurality of position detection devices as shown in FIGS. 15 to 17 in the X direction.
In FIGS. 15 to 17, the case where only the secondary winding is wound as the dummy winding has been described. However, the same winding (primary and secondary windings) as the position detection device winding is used for the dummy. And only the output of the secondary winding may be differentially wound. As a result, the influence of the magnetic flux on the dummy secondary winding caused by the induced electromotive force generated in the dummy primary winding can be eliminated. Note that the dummy iron core 2D may be omitted, and only the primary and secondary windings for the dummy or the secondary winding may be provided.

FIG. 18 is a view showing an eighth modification of the position detection device of the present invention, and FIG. 18 (A) is a side view of the position detection device as viewed from the direction perpendicular to the traveling direction X, FIG. 18B is a front view seen from the moving direction X side. The position detection device of FIG. 18 is a modification of that of FIG. That is, the position detection device of FIG. 18 is different from that of FIG. 17 in that each of the iron cores 21A to 24A has a U-shape (horse-shoe shape), and the magnetic shield plate 82 causes the magnetic flux from the magnet rail 30 to each primary winding. The point is that the lines S1 to S4 and the secondary windings P1 to P4 are not affected.
FIG. 19 is a view showing a ninth modification of the position detection device of the present invention, and FIG. 19A is a side view of the position detection device as viewed from the direction perpendicular to the traveling direction. 19 (B) is a front view seen from the moving direction side. The position detection device of FIG. 19 is a modification of that of FIG. 18, and the difference is that each of the iron cores 21 </ b> B to 24 </ b> B is annular and magnetically forms a closed circuit. Thereby, the magnetic flux generated in the primary windings P1 to P4 can be efficiently coupled to the secondary windings S1 to S4. In FIG. 18, the planar portions of the iron cores 21 </ b> A to 24 </ b> A are along the traveling direction X, whereas in FIG. 19, the planar portion of the annular core is along the vertical direction of the traveling direction X.

FIG. 20 is a diagram showing a tenth modification of the position detection device of the present invention, and FIG. 20 (A) is a side view of the position detection device as viewed from the direction perpendicular to the traveling direction. 20 (B) is a front view seen from the moving direction side. The position detection device of FIG. 20 is a modification of that of FIG. 19, and the difference is that an annular iron core 21 </ b> B to 24 </ b> B as shown in FIG. It is composed of iron cores 21 </ b> C to 24 </ b> C that are bent in a (U-shaped) shape and arranged so that the bending direction is along the vertical direction of the traveling direction X, and both end sides of each iron core 21 </ b> C to 24 </ b> C are in contact with the magnet rail 30. It is a point that has been.
FIG. 21 is a view showing an eleventh modification of the position detecting device of the present invention, and FIG. 21 (A) is a side view of the position detecting device as viewed from the direction perpendicular to the traveling direction. 21 (B) is a front view seen from the moving direction side. The position detection device of FIG. 21 is a modification of that of FIG. 19, and the difference is that the annular iron cores 21 </ b> B to 24 </ b> B as shown in FIG. The iron cores 21 </ b> D to 24 </ b> D bent into a (U shape) shape are configured such that both end sides of the iron cores 21 </ b> D to 24 </ b> D are in contact with the magnet rail 30.
FIG. 22 is a view showing a twelfth modification of the position detection device of the present invention, and is a side view of the position detection device as viewed from the direction perpendicular to the traveling direction. Since the front view is as shown in FIG. 20A, it is omitted here. The position detection device of FIG. 22 is configured by iron cores 21E to 24E obtained by bending annular iron cores 21B to 24B as shown in FIG. 19A into a U-shape along the movement direction X, Each iron core 21E-24E is arrange | positioned so that it may become a predetermined positional relationship. That is, the position detection device of FIG. 22 is arranged so that the traveling direction of the position detection device as shown in FIG. 20 is rotated 90 degrees and the positional relationship between the iron cores is the above-described predetermined relationship.
FIG. 23 is a view showing a thirteenth modification of the position detection device of the present invention, and is a side view as seen from the direction perpendicular to the traveling direction of the position detection device. Since the front view is as shown in FIG. 21A, it is omitted here. The position detection device of FIG. 23 is arranged so that the traveling direction of the position detection device of FIG. 21 is rotated by 90 degrees and the positional relationship between the iron cores 21F to 24F is the above-described predetermined positional relationship.

FIG. 24 is a view showing a fourteenth modification of the position detection device of the present invention, and FIG. 24 (A) is a side view of the position detection device viewed from the direction perpendicular to the traveling direction. (B) is the figure which looked at the position detection apparatus from the perpendicular | vertical upper surface direction with respect to the advancing direction, and FIG.24 (C) is the front view seen from the moving direction side. This position detection apparatus includes H-shaped iron cores 21G to 24G in which rectangular plate-shaped iron cores are coupled by a cylindrical iron core, and primary windings P1 to P4 and secondary windings S1 to S1 wound around the cylindrical iron core. It is comprised from S4 and the permanent magnet 35 provided in the one side of the rectangular plate-shaped iron core. In the case of this position detection device, the sum of the moving direction lengths of the two magnets 31 and 32 constituting the magnet rail 30 is one pitch. The lower surface of the permanent magnet 35 (the surface in contact with the iron cores 21G to 24G) is N pole, the upper surface of the magnet 31 (the surface in contact with the iron cores 21G to 24G) is N pole, and the upper surface of the magnet 32 (the surface in contact with the iron cores 21G to 24G). ) Is the S pole. Therefore, as shown in FIG. 24, when the area of the rectangular plate-shaped iron core on the lower side of the iron core 24 is in contact with the south pole magnet, the magnetic flux passing through the columnar iron core of the iron core 24 is saturated, so Thus, the degree of magnetic coupling between the primary winding P4 and the secondary winding S4 becomes small. On the other hand, when the area where the lower rectangular plate-shaped iron core is in contact with the N-pole magnet is the largest as in the iron core 21G, the magnetic flux passing through the columnar iron core of the iron core 24 is small, and the primary winding P1 The degree of magnetic coupling of the secondary winding S1 increases. Therefore, as a result, it is possible to detect the relationship of the relative positions on the magnet rail 30 based on the same principle as that of the position detection device described above.
24, if the permanent magnets 35 are omitted, the arrangement may be arranged so that the moving direction length of one magnet is 1 pitch.

Next, an example in which the position detection device of the present invention is applied to a rotary electric motor, that is, a rotary motor will be described.
25 and 26 are diagrams showing an embodiment in which the position detection device according to the present invention is applied to a rotary synchronous motor. FIG. 25 shows a cross-sectional structure including the rotary shaft of the rotary synchronous motor. FIG. 26 shows a cross-sectional structure of the AA plane in the rotary synchronous motor of FIG.
The rotary synchronous motor 90 is a three-phase AC drive type motor having four magnetic poles. The rotary synchronous motor includes a cylindrical stator frame 91 and a rotary shaft 94 that is rotatably provided on the stator frame 91 via a bearing 92 and a bearing 93. The stator frame 91 is provided with an armature core 95 and eight position detecting devices 96A to 96H, and the rotating shaft 94 is provided with four permanent magnets 97A to 97D.
The armature core 95 is formed of a cylindrical stratified iron core provided along the inner peripheral surface of the stator frame 1, and extends in the radial direction around the rotation axis as shown in FIG. There are only 24 slots. A three-phase armature winding is wound around each slot of the armature core 95. The stratified iron core of the armature core 95 is configured by stacking a plurality of thin silicon steel plates along the axial direction.

  The position detection devices 96A to 96H are formed of an annular core provided on both sides of the armature core 95 on the inner peripheral surface side of the stator frame 91, and a primary winding and a secondary winding wound around the annular core. Composed. In FIG. 26, only the position detection device corresponding to FIG. 1 is shown, but actually, the secondary winding S1 of the position detection device 96B is provided in the clockwise direction of the primary winding P5. The primary winding S4 of the position detection device 96D is provided in the counterclockwise direction of the primary winding P1, and the position detection device 96C is provided on the point symmetry side of the position detection device 96A. That is, position detecting devices 96A to 96D are provided in order along the annular core, and position detecting devices 96E to 96H are provided on one annular core, respectively, and each position detecting device has a primary winding and a secondary winding. Is wound.

As shown in FIG. 26, only one position detection device 96A may be provided. However, by providing a plurality of position detection devices, errors due to eccentricity of the rotation shaft can be averaged, and the position There is an effect that detection accuracy can be improved.
Further, in place of the position detecting device of FIG. 1, a position detecting device as shown in FIGS. 9 and 12 to 14 may be provided, or a dummy winding as shown in FIGS. 15 to 17 may be provided. A position detecting device as shown in FIGS. 18 to 24 may be provided. In addition, although the position detection apparatus demonstrated the case where it provided in the side surface side of the armature core 95, you may provide it in the side surface side of magnet 97A-97D.

  FIG. 27 is a diagram illustrating an example of application to an outer rotor type electric motor. In this outer rotor type electric motor, an armature winding serving as a stator is provided in the outer rotor, and the outer rotor provided with magnets 98A to 98D is rotated. Note that the shaft is omitted. In this case, a position detection device as shown in FIG. 26 is provided along the magnet on the outer peripheral surface side of the outer rotor. Note that the position detection device may be provided on the inner peripheral surface side of the outer rotor, or may be provided on the side surface side.

  In the above-described embodiment, the example using the permanent magnet has been described as an example. However, it is needless to say that the present invention can be similarly applied even when an electromagnetic magnet is used instead of the permanent magnet.

  According to the present invention, there is an effect that the rotational position or linear position of a rotary or linear motor using a permanent magnet or an electromagnet can be detected without providing a special magnetoresistance change member or the like.

1 is a partial cross-sectional view showing a schematic configuration of a linear position detection apparatus according to an embodiment of a position detection apparatus according to the present invention. The figure which shows the connection state of primary winding P1-P5 and secondary winding S1-S4 of FIG. The block diagram which shows an example of the measurement circuit of the phase detection type which converts the output from the position detection apparatus of FIG. 1 into position data. The block diagram which shows another example of the measurement circuit of a phase detection type which converts the output from the position detection apparatus of FIG. 1 into position data. Operation | movement explanatory drawing of the measurement circuit of FIG. The block diagram which shows the example of a change added to the measurement circuit of FIG. The block diagram which shows another example of the measurement circuit of the phase detection type which converts the output from the position detection apparatus of FIG. 1 into position data. FIG. 8 is an operation explanatory diagram of the measurement circuit of FIG. 7. The figure which shows the 1st modification of the position detection apparatus of this invention. The figure which shows the connection state of primary winding P1A-P4B and secondary winding S1-S4 of FIG. FIG. 11 is a block diagram showing an example of a phase detection type measurement circuit that converts the output from the position detection device of FIG. 9 wired as shown in FIG. 10 into position data. The figure which shows the 2nd modification of the position detection apparatus of this invention. The figure which shows the 3rd modification of the position detection apparatus of this invention. The figure which shows the 4th modification of the position detection apparatus of this invention. The figure which shows the 5th modification of the position detection apparatus of this invention. The figure which shows the 6th modification of the position detection apparatus of this invention. The figure which shows the 7th modification of the position detection apparatus of this invention. The figure which shows the 8th modification of the position detection apparatus of this invention. The figure which shows the 9th modification of the position detection apparatus of this invention. The figure which shows the 10th modification of the position detection apparatus of this invention. The figure which shows the 11th modification of the position detection apparatus of this invention. The figure which shows the 12th modification of the position detection apparatus of this invention. The figure which shows the 13th modification of the position detection apparatus of this invention. The figure which shows the 14th modification of the position detection apparatus of this invention. It is a figure which shows one Embodiment at the time of applying the position detection apparatus of this invention to a rotary synchronous motor, and is a figure which shows the cross-sectional structure containing the rotating shaft of the rotary synchronous motor. The figure which shows the cross-section of the AA surface in the rotary synchronous motor of FIG. The figure which shows one Embodiment at the time of applying the position detection apparatus of this invention to the motor of an outer rotor type | mold.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Position detection apparatus 2 Iron core 3 Winding part 30 Magnet rail 31, 32, 35 Magnet P1-P4, P1A-P4B, P6 Primary winding S1, S2, S3, S4 Secondary winding D1, D2, D3, D4 Dummy windings 21A-21G, 22A-22G, 23A-23G, 24A-24G

Claims (3)

A position detecting device in a linear or rotary electric motor having a movable part and a stationary part and having a magnet on the movable part or stationary part side,
A detection unit provided so as to be movable relative to the magnet, comprising: winding means for alternating current excitation; and a magnetic core disposed in association with the winding means, wherein the magnetic core In accordance with the proximity of the magnet to the magnetic core, magnetic saturation is caused at a location where the magnet core is strongly subjected to the magnetizing force to reduce the magnetic permeability of the magnetic core, and the magnetic core is moved along with the relative movement of the magnet. The detection unit in which the magnetic permeability of the magnetic core is increased by not causing magnetic saturation in response to a change in magnetizing force
Equipped with,
The detection unit is arranged on the movable unit or the stationary unit of the electric motor where the magnet is not provided, and the relative position of the magnet and the detection unit changes according to the displacement of the movable unit. Accordingly, the magnetizing force of the magnet on the magnetic core changes between a state where magnetic saturation does not occur and a state where magnetic saturation occurs, and the magnetic permeability of the magnetic core changes according to this change. A position detecting device characterized in that an inductance of the winding means associated with the magnetic core changes, and an output signal based on the change in the inductance is generated from the winding means. The position detection device according to claim 1 , wherein the magnet includes a plurality of magnets arranged at a predetermined pitch along a moving direction of a detection target. Said winding means includes a plurality of windings arranged in different positions along the moving direction of the detection target, that formed by arranging a plurality of said magnetic core in association with each of the plurality of windings The position detection device according to claim 1 or 2 , characterized in that

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