JP4342509B2 - Radio receiving apparatus and radio receiving method - Google Patents

Description

  The present invention relates to a wireless communication receiver, and more particularly to a spatial multiplexing transmission receiver that transmits and receives signals using a plurality of transmission antennas and reception antennas.

  As a technique for speeding up wireless communication, a technique has been proposed in which transmission signals are distributed to a plurality of radio units and signals are simultaneously transmitted from a plurality of transmission antennas using the same frequency (for example, see Non-Patent Document 1). ). In this method, since signals transmitted from each antenna are received through different propagation paths, the receiving terminal receives signals using a plurality of receiving antennas, and signals transmitted simultaneously using the received signals. By separating the signals, the signal can be decoded. As a result, the transmission speed can be increased according to the number of multiplexed signals without increasing the frequency bandwidth used for communication. Therefore, according to this method, the frequency utilization efficiency can be increased and the throughput can be improved.

  On the other hand, in an environment where signals with different propagation delay times between transmission and reception arrive in a multipath propagation path, waveform distortion due to intersymbol interference is a major factor that degrades communication quality. In such an environment, the Orthogonal Frequency Division Multiplexing (hereinafter referred to as OFDM) method is a method that can compensate for waveform distortion caused by intersymbol interference even when signals having different propagation delay times are received. Known as.

Since the signal becomes a complex signal in the OFDM transmission system, it is necessary to use a quadrature modulator in the transmitter and a quadrature demodulator in the receiver. At this time, if an error occurs in the amplitude of the in-phase component and the quadrature component in the quadrature modulator and the quadrature demodulator, or if a phase error occurs in the 90-degree transporter, among a plurality of subcarriers of the OFDM signal, Two subcarriers that are symmetrical with each other on the frequency axis with respect to the center frequency (of these two subcarriers, the subcarrier in the higher frequency band than the center frequency is the “upper-band subcarrier”, and the lower frequency than the center frequency. Band subcarriers may be referred to as "lower sideband subcarriers"), which may interfere with each other, greatly limiting transmission performance. In such an environment, a method has been proposed in which the amount of interference between upper and lower subcarriers is estimated, and signals are determined by maximum likelihood estimation or spatial filtering using both signals received by the upper and lower subcarriers (for example, Non-patent document 2).
A. Paulraj, R. Nabar, and D. Gore, Introduction to Space-Time Wireless Communications, Cambridge University Press, UK, 2003, pp. 6-10. Hiroyuki Kamada, Kei Sakaguchi, Junmichi Araki, "Effect of IQ Imbalance and its Compensation Method in MIMO-OFDM," IEICE Technical Report, A / P2004-238, pp. 7-12, March 2005.

  If the method of Non-Patent Document 2 is used, distortion caused by the quadrature modulator / demodulator can be compensated with high accuracy. However, when maximum likelihood estimation is used because signals of two subcarriers are processed at the same time, there is a problem that the amount of calculation increases by a square or more, which makes implementation difficult. On the other hand, an increase in calculation load can be minimized by using spatial filtering, but there is a problem that the performance is greatly degraded as compared with the case of using maximum likelihood estimation.

  As described above, the conventional radio receiving apparatus has a problem in that the transmission performance is largely limited by the accuracy of the amplitude and phase of the quadrature modulator / demodulator. Further, the conventional radio receiving apparatus has a problem that the amount of calculation increases when a method with high reception performance is used when compensating for waveform distortion caused by imperfection of the orthogonal modulator / demodulator.

  Therefore, in view of the above-described problems, the present invention provides two subcarriers (upper sideband / lower sideband) that are symmetrical with each other on the frequency axis with respect to the center frequency due to imperfection of the quadrature modulator / demodulator. The purpose is to compensate for interference occurring between subcarriers) and to improve reception performance. It is another object of the present invention to provide a radio receiving apparatus that realizes high-accuracy reception while minimizing an increase in calculation load caused by compensation.

The wireless receiver of the present invention is
(A) An OFDM (Orthogonal Frequency Division Multiplexing) signal including a plurality of subcarriers is received by an antenna;
(B) Combining the signals received on each of the two subcarriers for each of the two subcarriers at symmetrical positions on the frequency axis around the center frequency of the plurality of subcarriers of the received OFDM signal To obtain the combined signal,
(C) Extracting signal components transmitted from subcarriers in a frequency band higher than the center frequency out of the two subcarriers from the combined signals,
(D) Extracting a signal component transmitted from a subcarrier in a frequency band lower than the center frequency from the two subcarriers from the combined signal;
(E) Maximum likelihood estimation of the signal transmitted on the subcarrier from the signal component transmitted on the subcarrier in the high frequency band extracted from the combined signal;
(F) Maximum likelihood estimation of the signal transmitted on the subcarrier is performed from the signal component transmitted on the subcarrier in the low frequency band extracted from the combined signal.

  According to the present invention, reception performance can be improved with a small amount of computation.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

  Hereinafter, a radio reception apparatus in MIMO (Multi-Input Multi-Output) -OFDM (Orthogonal Frequency Division Multiplexing) transmission will be described.

(First embodiment)
FIG. 12 illustrates an example of a configuration of a transmission device that generates and transmits a signal received by the wireless reception device according to the first embodiment. The information signal to be transmitted is input to the serial-parallel converter 201 and converted into the same number of parallel signals as the number of multiplexed signals in spatial multiplexing. At this time, the serial-parallel conversion may be performed bit by bit or may be performed in units of bits corresponding to the modulation multilevel number applied by the modulators 202a to 202d. Also, conversion may be performed for each number of bits included in the OFDM symbol or in other units. The number of bits is a predetermined number, and conversion may be performed in any unit as long as the wireless reception device is known.

  As described above, the signal groups output from the serial-parallel conversion unit 201 (here, for example, four-sequence signal groups) are modulated by the sub-carriers (subcarriers) in the corresponding modulators 202a to 202d. Applied. Modulation schemes used in the modulators 202a to 202d include phase modulation such as BPSK, QPSK, and 8PSK, and quadrature amplitude modulation such as 16QAM, 64QAM, and 256QAM.

  Here, the operation of the modulators 202a to 202d when 64QAM is used as the modulation method will be described with reference to FIG. FIG. 16 shows a constellation of 64QAM, which shows the correspondence between input bit strings and generated symbols. In 64QAM, a 6-bit signal can be transmitted in one symbol, and different symbols are generated according to the input bit string. As an example, consider the case where 6 bits of “000100” are input. At this time, the signal point for the input bit string is the upper left point from FIG. 16, and the upper left point is output as a transmission symbol.

  The modulators 202a to 202d perform the above processing for each subcarrier. Therefore, when the number of OFDM subcarriers is N, N symbols are generated for 6 × N-bit input.

  Although the modulator has been described using 64QAM, the modulation method is not limited to 64QAM. Other modulation schemes described above may be used, or other modulation schemes may be used. Any modulation method may be used as long as the wireless reception device can demodulate with a predetermined modulation method.

  Also, each modulator may perform modulation using the same modulation method, or a different modulation method may be applied to each modulator.

  As described above, after modulation is performed for each subcarrier by the modulators 202a to 202d, the inverse Fourier transform units 203a to 203d convert the modulation signals output from the modulators 202a to 202d into signals in the time domain. Convert each one. Here, the inverse Fourier transform in the inverse Fourier transform units 203a to 203d may use IDFT (Inverse Digital Fourier Transform) or IFFT (Inverse Fast Fourier Transform). Any method may be used as long as the discrete inverse Fourier transform is applied to the transmission symbol generated for each subcarrier.

  The signals after the inverse Fourier transform by the inverse Fourier transform units 203a to 203d are input to the GI (guard interval) adding units 204a to 204d, respectively.

  The GI adding units 204a to 204d copy a part of the end of the waveform to the top as shown in FIG. This signal is called a guard interval or a cyclic prefix, and is a method generally used in OFDM transmission. As a result, even if a delay wave exists in a multiple propagation environment as shown in FIG. 10, it is possible to prevent intersymbol interference from the preceding and succeeding symbols by removing the redundant signal added by the guard interval.

  In addition to the information signals described above, in wireless communication, a known signal is generally transmitted for synchronization and propagation path estimation. As an example of a communication system that transmits such a known signal, a frame format in IEEE 802.11a will be described with reference to FIG. In FIG. 11, the SP (short preamble) at the head of the frame is a known signal for synchronization, and the subsequent LP (long preamble) is a known signal for channel estimation. When such a known signal is transmitted, the known signal generation unit 207 and the radio unit 205a are connected by the switch 208 that connects one of the GI addition units 204a to 204d and the known signal generation unit 207 to the radio units 205a to 205d. By connecting ~ 205d, the radio units 205a to 205d transmit the known signals generated by the known signal generation unit 207.

  The transmission of the known signal has been described above by taking the IEEE 802.11a frame format as an example. However, the signal that can be received by the wireless reception device according to the present invention is not limited to IEEE 802.11a.

  The above information signals and known signals are input to the radio units 205a to 205d, converted into analog signals, converted into radio frequency signals, and then transmitted via the transmission antennas 206a to 206d. Radio units 205a to 205d in the transmission apparatus are general radio units including a digital-analog converter, a quadrature modulator, a frequency converter, a filter, an amplifier, and the like, and do not constitute a main part of the present invention. Since there is no description, the detailed explanation is omitted.

  As for the transmission antennas 206a to 206d, any antenna may be used as long as a signal with a desired frequency can be transmitted.

  FIG. 1 shows a configuration example of a radio reception apparatus according to the first embodiment for receiving a signal transmitted by the transmission apparatus described above.

  1 includes a plurality (for example, four) of receiving antennas 1a to 1d, and a plurality of (for example, four) of radio units 2a to 2d connected to the receiving antennas 1a to 1d, respectively. , A plurality (here, for example, four) of GI (guard interval) removal units 3a to 3d connected to each of the plurality of radio units 2a to 2d and a plurality (here) connected to the plurality of GI removal units 3a to 3d, respectively. Then, for example, four) Fourier transform units 4a to 4d, a propagation path estimation unit 5 connected to the plurality of Fourier transform units 4a to 4d, and a reception signal combining unit 6 connected to the plurality of Fourier transform units 4a to 4d. And a weight calculation unit 7 connected to the propagation path estimation unit 5, a signal separation unit 8 connected to the reception signal combination unit 6 and the weight calculation unit 7, and a signal separation unit 8 and a weight calculation unit 7. Two Maximum likelihood estimation units 9 and 10, and two parallel likelihood conversion units 11 connected to the two maximum likelihood estimation units 9 and 10.

  Here, any of the plurality of receiving antennas 1a to 1d in FIG. 1 may be used as long as they can receive signals in a desired frequency band. Each antenna may be the same antenna or a different antenna.

  Next, the radio units 2a to 2d in FIG. 1 will be described. Each of the radio units 2a to 2d includes an amplifier, a filter, a frequency converter, an analog-digital converter, and the like, and receives radio frequency signals received by the receiving antennas 1a to 1d connected to the radio units 2a to 2d. Convert to digital signal. The radio units 2a to 2d may use any configuration as long as the received radio frequency signal can be converted into a complex digital signal. A radio frequency signal may be converted into an intermediate frequency signal and then converted into a baseband signal, or a radio frequency signal may be directly converted into a baseband signal without passing through the intermediate frequency. Further, analog quadrature demodulation may be performed, or digital / quadrature demodulation may be performed using an analog-digital converter output.

  In addition, amplifiers, filters, frequency converters, and analog-digital converters in the radio units 2a to 2d are general objects and are not the gist of the present invention, and thus detailed descriptions thereof are omitted.

  As described above, the signals converted into digital signals by the radio units 2a to 2d are input to the GI removal units 3a to 3d, respectively. As described in the explanation of the transmission apparatus, in OFDM transmission, a guard interval is generally added as shown in FIG. The GI removal units 3a to 3d in FIG. 1 remove the guard interval length signal added during transmission.

  The signals from which the guard interval has been removed are input to the Fourier transform units 4a to 4d, respectively. In OFDM transmission, a signal is transmitted using a plurality of subcarriers, and a signal transmitted on each subcarrier can be extracted by applying a discrete Fourier transform. Each Fourier transform unit in FIG. 1 applies a discrete Fourier transform for each signal in the Fourier transform application range shown in FIG. 10, which has the same length as the effective OFDM length shown in FIG.

  Here, the discrete Fourier transform may be calculated using DFT (Digital Fourier Transform) or FFT (Fast Fourier Transform). Any method may be used as long as discrete Fourier transform can be performed for each OFDM symbol.

Next, consider the signals output by each Fourier transform unit. Frequency selective fading occurs in the signal propagated through the multipath propagation path due to the influence of delayed waves having different propagation delay time differences. As a result, when performing OFDM transmission, the propagation path response differs for each subcarrier. Further, since the signals are transmitted by being spatially multiplexed from the plurality of transmission antennas, the received signal r _i ^(k) ( ^k) of the k-th subcarrier in the m-th OFDM symbol output from the i-th Fourier transform unit. m) can be expressed by the following formula (1).

Here, h _ij ^(k) represents a channel response (channel estimation value) in the k-th subcarrier between the j-th transmitting antenna and the i-th receiving antenna, and s _j ^(k) (m ) Represents the signal of the k-th subcarrier of the m-th OFDM symbol transmitted from the j-th transmission antenna, and n _i ^(k) (m) represents the signal at the output of the i-th Fourier transform unit. It represents the thermal noise added to the kth subcarrier of the mth OFDM symbol. Further, if a received vector having the element of the signal of the kth subcarrier in the mth OFDM symbol output from each Fourier transform unit is R ^(k) (m), it is expressed as a vector as the following equation: I can do things.

S ^(k) (m) is a transmission vector whose elements are signals of the k-th subcarrier of the m-th OFDM symbol transmitted from each transmission antenna, and N ^(k) (m) is each A noise vector whose element is noise in the kth subcarrier of the mth OFDM symbol output from the Fourier transform unit, and H ^(k) is a propagation path response matrix expressed by the following equation.

  On the other hand, when the quadrature demodulator is configured with an analog circuit in the radio unit, it is generally difficult to keep the gains of the I channel and Q channel the same. In addition, it is difficult to accurately generate a phase difference of 90 degrees in the quadrature demodulator.

  Similarly, it is difficult to accurately control the amplitude and phase of the quadrature modulator in the radio unit of the transmission apparatus.

  Due to the imperfection of the quadrature modulator / demodulator, it is known that the received signal vector of the kth subcarrier represented by Equation (2) is subject to interference from the -kth subcarrier. .

  Of the two subcarriers at the target position on the frequency axis around the center frequency of the plurality of subcarrier signals, the subcarrier on the frequency band side higher than the center frequency is kth from the center frequency. The subcarrier on the frequency band side lower than the center frequency is the -kth subcarrier from the center frequency.

As a result, the received vector can be expressed as follows:

Similarly, the reception vector of the -k-th subcarrier receives interference from the signal of the k-th subcarrier, and can be expressed as the following equation.

Here, H _α ^(k) and H _β ^(k) are the propagation path matrix of the signal transmitted using the kth subcarrier in the received signal of the kth subcarrier, and the −kth subcarrier. Respectively, H _α ^(−k) and H _β ^(−k) are transmitted using the kth subcarrier in the received signal of the −kth subcarrier. Represents the propagation path matrix of the transmitted signal and the propagation path matrix of the signal transmitted using the -k-th subcarrier, respectively, and is affected not only by the propagation path response but also by the imperfection of the quadrature modulator / demodulator. Yes.

(Description of received signal coupling unit)
As described above, the kth subcarrier is transmitted to both the received signals of the kth subcarrier and the -kth subcarrier at the target position on the frequency axis around the center frequency of the plurality of subcarriers. It can be seen that the received signal and the signal transmitted on the -k-th subcarrier are included. Therefore, by simultaneously estimating the signals transmitted on the two subcarriers using the signals received on the two subcarriers, the signals can be replaced with signals for determining the interference due to the imperfection of the quadrature modulator / demodulator.

For the above purpose, the reception signal combining unit 6 in FIG. 1 receives the reception signal R ^(k) (m) of the upper sideband subcarrier and the reception signal R ^(−k) (m) of the lower sideband subcarrier. ) Are combined as shown in the following equation (6) to generate an expanded received vector R _k (m).

In the example of FIG. 1, since R ^(k) (m) and R ^(−k) (m) are 4 × 1 complex vectors, R _k (m) is an 8 × 1 complex vector, It can be expressed as follows using equations (4) and (5).

  When the number of subcarriers for transmitting data is N, the reception signal combining unit 6 generates N / 2 combination vectors as shown in Expression (6) and outputs them.

In order to demodulate using the signal output from the reception signal combining unit 6, it is necessary to estimate what distortion the transmission signal has received due to radio wave propagation and the incompleteness of the quadrature modulator / demodulator. H _k must be estimated. In order to estimate this distortion, a known signal for propagation path matrix estimation is generally transmitted before data is transmitted. An example of such a transmission method for transmitting a known signal for channel response estimation is IEEE802.11a described above, and LP corresponds to the known signal for channel response estimation in the frame format shown in FIG. .

  In the case of a transmission method in which a plurality of signals are transmitted simultaneously using spatial multiplexing assumed in the present embodiment, it is necessary to estimate a propagation path response for each signal to be spatially multiplexed.

There are various methods for estimating the propagation path response using the transmitted known signal. Typical techniques include a technique of multiplying a received signal in the frequency domain by a generalized inverse matrix generated from a known signal, a technique of estimating an impulse response in the time domain, and converting it to the frequency domain. Note that the propagation path estimation unit 5 in this embodiment may use any method. Any method other than the above two methods may be used. Any method may be used as long as it can estimate the propagation path matrix H _k of the combined received signal shown in Expression (7).

By using the output R _k (m) of the reception signal combining unit 6 and the channel matrix H _k estimated by the channel estimation unit 5, the same technique as that used when receiving a conventional spatially multiplexed OFDM signal is used. Thus, S _k (m) can be demodulated. The maximum likelihood estimation method having the highest reception performance among the reception methods is a method for obtaining S _k (m) having the highest probability of receiving R _k (m) when a certain S _k (m) is transmitted. The estimation is performed by the following equation.

Here, when the modulation multi-level number of the j-th signal transmitted using spatial multiplexing is M _j , M ₁ M ₂ is used when estimation is performed for each subcarrier signal without combining the received signals. It was only necessary to search for a signal from among the M ₃ M ₄ signal points. However, since the signals of the two subcarriers must be estimated simultaneously by combining the received signals as in the present embodiment, (M ₁ M ₂ M ₃ M ₄ ) ^Two signal points need to be searched. Therefore, there is a problem that the number of signal searches becomes square by combining received signals, and the calculation load when searching for signals increases.

Therefore, in the radio reception apparatus according to the present embodiment, a spatial filter is applied to the signal R _k (m) combined by the reception signal combining unit 6, and the upper sideband signal extracting unit 8a in FIG. Only the signal S ^(k) (m) is extracted, and the lower sideband signal extraction unit 8b extracts only the lower sideband signal S ^(-k) (m).

After separating the upper sideband and lower sideband signals from the received signal in this way, the maximum likelihood estimation unit 9 performs maximum likelihood estimation only on the extracted upper sideband signal, and the maximum likelihood estimation unit 10 extracts the lower sideband. Maximum likelihood estimation is performed only on the waveband signal. As a result, the number of signal points searched by each maximum likelihood estimation unit is reduced to M ₁ M ₂ M ₃ M ₄ , and the number of signal searches is limited to the same number of searches as when the received signals are not combined. .

The signal separation unit 8 performs the above processing for each combined R _k (m), and the upper sideband signal extraction unit 8a and the lower sideband signal extraction unit 8b respectively have the following weight matrices W _u ^(k) and W _The result of multiplying _l ^(k) is output.

As described above, MMSE (Minimum ⁾ is an example of a technique for obtaining the weight W _u ^(k) for extracting only the upper sideband signal and the weight W _l ^(k) for extracting only the lower sideband signal. Mean Square Error) method. In the MMSE method, signals W _u ^{(k) H} R _k (m) (or W _l ^(k) R _k (m)) and S ^(k) (m) (or S ^(−k) (m )) Find the weight that minimizes the mean square of the error.

However, E [] represents a set average. Weight matrix of equation (10) can be calculated as follows by using the channel matrix H _k estimated.

Here, σ ² represents noise power, I is a unit matrix of dimension (2 × number of radio units) × (2 × number of radio units), and in the example of FIG. Become. | P | ² represents the average power of the spatially multiplexed wide-band carrier signal.

  The maximum likelihood estimation unit 9 in FIG. 1 estimates the signal transmitted by the upper sideband subcarrier extracted by the upper sideband signal extraction unit 8a described above by the maximum likelihood estimation method, and the maximum likelihood estimation unit 8b The signal transmitted on the lower sideband subcarrier extracted by the sideband signal extraction unit 8b is estimated by the maximum likelihood estimation method.

Here, consider the k-th subcarrier signal output by the upper sideband signal extraction unit 8a. From Equations (7) and (9), the signal can be written as:

Here, Expression (12) is a vector whose number of elements is equal to the number of spatially multiplexed signals, and the weight W _u ^{(k) H} is obtained so that j rows in the output become s _j ^(k) (m). It has been. However, s _j ^(k) (m) is not accurately obtained due to the propagation path response, the incompleteness of the quadrature modulator / demodulator, and the influence of thermal noise, and errors are included. Here, the error vector e _u ^(k) (m) is defined by the following equation (13), and a correlation matrix E _ee ^(k) of the error vector is obtained from the equation (14).

The above estimation is performed by the maximum likelihood estimation unit 9 for each of the 1st to N / 2th subcarriers. Similarly, for the lower sideband signal, the maximum likelihood estimator 10 estimates the signal transmitted on the 1st to -N / 2nd subcarriers using the following equation (17).

  Thus, the maximum likelihood estimator 9 and the maximum likelihood estimator 10 require an inverse matrix of the error vector correlation matrix. The matrix is calculated by the weight calculation unit 7 and input to the maximum likelihood estimation units 9 and 10.

In the maximum likelihood estimation unit 9 and the maximum likelihood estimation unit 10, any method may be used as a method for searching for a signal based on the equations (16) and (17). You may try all combinations of M ₁ M ₂ M ₃ M ₄ and like sphere decoding, K-Best, M-algorithm,
The search may be performed after converting E _ee ^{(k) -1} or E _ee ^{(-k) -1} to an upper triangular matrix and limiting candidates to be searched.

In addition, as shown in Expression (6), the signals received on the −kth subcarrier have been combined after obtaining the complex conjugate, but the complex of the signal received on the kth subcarrier has been obtained. You may obtain | require conjugation and may perform coupling | bonding. In that case, S ^(k) (m) in equations (7) to (16) is S ^{(k) *} (m), S ^{(−k) *} (m) is S ^(−k) (m), H _α ^(k) and H _β ^(k) are H _α ^{(k) *} and H _β ^{(k) *} , and H _α ^{(−k) *} and H _β ^{(−k) *} are H _α ^(−k) . It is only converted to H _β ^(−k) and there is no significant change.

In addition, each signal has been processed as a complex signal so far, but R ^(k) (m), R ^(−k) (m), S ^(k) (m), S ^(−k) (m) The vector may be expanded to twice the size so that the I channel signal and the Q channel signal are the respective elements, and processing may be performed by real number calculation. At this time, R _k (m) and S _k (m) are each a 16 × 1 real vector, and the combined propagation path matrix H _k is a 16 × 16 execution sequence, and Equations (8) to (18) The processing shown is also processed by real number calculation.

  As a result, the maximum likelihood estimation unit 9 outputs the estimation result of the signal transmitted on the upper sideband subcarrier, and the maximum likelihood estimation unit 10 outputs the estimation result of the signal transmitted on the lower sideband subcarrier. Is done.

  In the parallel-serial converter 11 of FIG. 1, the outputs of the two maximum likelihood estimation units 9 and 10 are parallel-serial converted to generate one signal series. This rearrangement order must be performed according to the order of serial-parallel conversion performed at the time of transmission, and the order of serial-parallel conversion at the time of transmission must be known on the receiving side.

  The order of serial-parallel conversion at the time of transmission in this embodiment may be any procedure. You may allocate to the signal spatially multiplexed 1 bit at a time, and you may allocate to the signal spatially multiplexed for every OFDM symbol. The allocation to each subcarrier may be performed in any order. Allocation may be performed in any order as long as it is a predetermined order and the receiving apparatus has a known order.

  As described above, according to the first embodiment, in the environment where the signals of the subcarriers in the upper sideband and the lower sideband interfere with each other due to imperfections of the quadrature modulator and the quadrature demodulator, The signals of each carrier wave are separated by spatial filtering (smart antenna) using both received signals, and the maximum likelihood estimation is performed for each subcarrier on the separated signal to perform reception with high accuracy. Is possible.

  In addition, if the maximum likelihood estimation is performed on the signals of the kth and −kth subcarriers at the same time, the computation load becomes enormous. However, the reception signal combining unit 6 receives signals on the upper and lower sideband subcarriers. Since the signals are combined and each subband signal of each sideband extracted by the signal separation unit 8 is estimated using the two maximum likelihood estimation units 9 and 10, the maximum likelihood estimation units 9 and 10 It is possible to prevent the calculation load from increasing.

(Second Embodiment)
A radio reception apparatus according to the second embodiment of the present invention will be described.

  The configuration of the radio receiving apparatus in this embodiment is the same as that of the radio receiving apparatus in the first embodiment of FIG. 1, and the received signal combining unit 6 has a frequency axis centered on the center frequency of a plurality of subcarriers. Combining the received signals of the subcarrier paths (kth and -kth subcarriers) (in the upper and lower sidebands) at symmetrical positions, the respective sidebands (kth and -k) are combined. It is also the same that the maximum likelihood estimation is performed after the signal transmitted on the (th subcarrier) is separated.

  The second embodiment is different from the first embodiment in that the weight calculation method in the weight calculation unit 7 calculates weights based on Zero Forcing (hereinafter referred to as ZF) rather than MMSE standards.

When obtaining the weight based on the ZF standard, W _u ^(k) and W _l ^(k) are obtained so as to satisfy the following expression.

When the above weight is used, the error signal of the signal output from the signal separation unit 8 represented by the equation (13) is expressed by the following equation (21), unlike the equation (14) in the first embodiment. The

An inverse matrix calculated by the weight calculation unit 7 of the above matrix, the inverse matrix of E _{ee ^(k)} is the maximum likelihood estimation unit 9, the inverse matrix of E _{ee ^(-k)} is outputted to the maximum likelihood estimator 10 .

  The maximum likelihood estimator 9 and the maximum likelihood estimator 10 perform signal estimation in the same manner as in the first embodiment using the above matrix, and the parallel-serial converter 11 operates in exactly the same manner as in the first embodiment. .

  As described above, according to the present invention, due to imperfection of the quadrature modulator and the quadrature demodulator, it is possible to receive with high accuracy even in an environment where the signals of the subcarriers of the upper sideband and the lower sideband interfere with each other. It becomes possible. Further, two maximum likelihood estimators are provided for each sideband subcarrier signal extracted by the signal separator 8 by combining signals received by the received signal combiner 6 with upper and lower sideband subcarriers. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Third embodiment)
An example of the configuration of a wireless reception apparatus according to the third embodiment is shown in FIG. The third embodiment is different from the first embodiment and the second embodiment in that a decoder 12 that inputs the output of the parallel-serial converter 11 is added. In FIG. 2, the same parts as those in FIG.

  First, a transmission apparatus that transmits a signal received by the wireless reception apparatus in FIG. 2 will be described with reference to FIG. FIG. 13 shows a configuration example of the transmission apparatus. In FIG. 13, the same parts as those in FIG.

  The information signal to be transmitted is input to the encoder 301, and the signal encoded by the encoder 301 is converted into parallel signals of the same number as the number of multiplexed signals when transmitting by spatial multiplexing in the serial-parallel converter 201. Here, various encoding schemes such as a convolutional code, a turbo code, and LDPC can be considered as the encoding scheme used in the encoder 301. However, the present embodiment does not limit the encoding scheme. Any method may be used as long as it is a predetermined encoding method and the wireless reception device is a known method.

  The signals converted in parallel by the serial / parallel conversion unit 201 are respectively input to the interleaving units 302a to 302d, and the signals are rearranged. In general, in OFDM transmission, characteristics are different for each subcarrier due to the influence of frequency selective fading. Therefore, if the encoded signal is continuously assigned to subcarriers with poor characteristics, the error correction capability deteriorates. In order to prevent such performance deterioration, signals are rearranged in the interleave units 302a to 302d.

  Here, the order of rearrangement of signals in the interleave units 302a to 302d may be any format, and may be any order as long as the wireless reception device is known in advance. Further, although the same number of interleave units as the number of spatial multiplexing is used, the rearrangement may be performed in the same order or may be performed in a different order for each interleave unit.

  The subsequent processing performed on the outputs of the interleave units 302a to 302d is the same as that of the transmission apparatus of FIG. 12 described in the description of the first embodiment, and detailed description thereof is omitted.

  Next, the operation of the radio reception apparatus according to the third embodiment for a signal transmitted from the transmission apparatus having the configuration shown in FIG. 13 will be described with reference to FIG.

  In FIG. 2, the decoder 12 performs hard-decision decoding in this embodiment, which performs error correction decoding.

  Receiving antennas 1a to 1d, radio units 2a to 2d, GI removing units 3a to 3d, Fourier transform units 4a to 4d, propagation path estimating unit 5, received signal combining unit 6, weight calculating unit 7, signal separating unit 8, maximum likelihood The estimation unit 9 and the maximum likelihood estimation unit 10 operate in the same manner as in the first embodiment and the second embodiment.

  On the other hand, in the transmission device of FIG. 13, since the signals are rearranged in the interleave units 302a to 302d, unlike the transmission device of FIG. 12, the parallel-serial conversion unit 11 of FIG. The parallel / serial conversion is performed in consideration of bit replacement in the serial / parallel conversion unit 201 and the interleave units 302a to 302d. Here, the order of bit rearrangement in the serial / parallel conversion unit 201 and the interleaving units 302a to 302d is determined in advance and is a known order in the radio reception apparatus, and therefore the parallel / serial conversion unit 11 in FIG. Rearranges the bits according to a predetermined order.

  The decoder performs decoding using the bit string output from the parallel-serial converter 11 and the Hamming distance of the replica signal generated by the decoder 12. Here, the Hamming distance represents the number of different bits in the two codes. For example, the Hamming distance of “000100” and “001000” is “2” because the third bit and the fourth bit are different. At this time, various decoding methods such as Viterbi decoding have been proposed for the decoder 12, but the decoder 12 in this embodiment does not limit the decoding method. Any method may be used as long as the information signal transmitted from the received code can be estimated.

  As described above, according to the present invention, due to imperfection of the quadrature modulator and the quadrature demodulator, it is possible to receive with high accuracy even in an environment where the signals of the subcarriers of the upper sideband and the lower sideband interfere with each other. It becomes possible. Furthermore, transmission characteristics can be further improved by applying encoding in the transmission apparatus.

  The received signal combining unit 6 combines signals received by upper and lower sideband subcarriers, and two maximum likelihood estimation units 9 for each sideband subcarrier signal extracted by the signal separation unit 8. Since the estimation is performed using 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Fourth embodiment)
A radio reception apparatus according to the fourth embodiment of the present invention will be described with reference to FIG.

  The configuration of the wireless reception device in the fourth embodiment is the same as that in the third embodiment. The fourth embodiment is different from the third embodiment in that the decoder 12 in FIG. 2 performs soft decision decoding instead of hard decision decoding.

  The radio units 2a to 2d, the GI removal units 3a to 3d, the Fourier transform units 4a to 4d, the propagation path estimation unit 5, the reception signal combination unit 6, the weight calculation unit 7, and the signal separation unit 8 are the same as in the third embodiment. It is.

  Hereinafter, a different part from 3rd Embodiment is demonstrated.

  When the decoder 12 performs soft decision decoding, since the decoder 12 uses the likelihood of the transmitted code bit, the maximum likelihood estimator 9 and the maximum likelihood estimator 10 determine the maximum likelihood symbol and the bits constituting the symbol. Instead of outputting, the likelihood of each bit is output. Here, the likelihood is the probability that “0” is transmitted and the probability that “1” is transmitted in each bit, or the ratio between the probability that “0” is transmitted and the probability that “1” is transmitted, or It is an approximation of them.

A specific example of outputting the probability that “0” and “1” are transmitted in each bit as an output of the maximum likelihood estimation unit 9 will be described below. From equation ^{(16), when S (k)} (m) is _transmitted, the probability density function to receive _{R k (m) p (R} k (m) | S (k) (m)) , the following It can be approximated by equation (22).

  However, L represents the spatial multiplexing number, and is “4” in the example of the transmission apparatus in FIG.

^{Here, S (k) (m)} s j (k) is the element of the ^(m) is the symbol from the number of bits corresponding to the modulation level is formed, and outputs the likelihood of each bit . The probability that the nth bit of s _j ^(k) (m) is “0” is the sum of all the probabilities in Equation (22) where the bit is “0”, and is expressed by the following equation.

The above calculation is performed for all the bits constituting S ^(k) (m), and the obtained probabilities are output.

Instead of the probability of each bit, the log likelihood ratio of each bit can be used as the output of the maximum likelihood estimation unit 9. Here, the log-likelihood ratio of the n-th bit of s _j ^(k) (m) can be expressed by the following equation using equations (23) and (24).

  The log likelihood ratio is a positive value when the probability that “1” is transmitted is higher than the probability that “0” is transmitted, and is negative when the probability that “0” is transmitted is higher. The larger the absolute value, the higher the probability that “1” or “0” was transmitted.

Further, as the output in the maximum likelihood estimation unit, the occurrence probability of each bit may be approximated only by the symbol that gives the maximum probability, instead of calculating the sum as in equations (23) and (24).

This output approximates the log-likelihood ratio of each bit and is not an exact log-likelihood ratio, but the calculation amount can be reduced because only the exponent part of Equation (22) is calculated.

As described above, the likelihood of each bit can be calculated by calculating Expression (22) for all combinations of S ^(k) (m). However, as described in the first embodiment, it becomes difficult to calculate all combinations of signals as the number of modulation levels of signals and the number of spatially multiplexed signals increase. A method for reducing the number of signal points to search such as -algorithm has been proposed. When these methods are used, it is impossible to calculate the equation (22) for all signal points. However, when these methods are applied to this embodiment, only the signal point for which the equation (22) is calculated is used. This can be applied by calculating equation (23), equation (24), (25) or equation (26) using.

  The method for obtaining the likelihood for each bit in the maximum likelihood estimation unit 9 has been described above, but the likelihood for each bit can also be obtained in the maximum likelihood estimation unit 10 by the same method. Further, although several methods have been described as methods for obtaining the likelihood for each bit, the method for obtaining the likelihood of each bit in the present embodiment is not limited to the above method. Any other method may be used as long as the likelihood for each bit is obtained.

  In this way, the reliability information of each bit output by the maximum likelihood estimation unit 9 and the maximum likelihood estimation unit 10 is used to convert the serial / parallel conversion unit 201 used in the transmission device by the parallel-serial conversion unit 11 of FIG. 2 in this embodiment. , Rearrangement is performed according to the order of bits in the interleave units 302a to 302d. Here, the order of bit rearrangement is known in advance as in the third embodiment.

  The decoder 12 performs soft decision decoding using the output of the parallel-serial converter 11 described above. Soft-decision decoding is more computationally intensive than hard-decision decoding, but performs decoding based on reliability information of each bit, and therefore has higher error correction capability than hard-decision decoding. In addition, various methods such as Viterbi decoding have been studied as the soft decision decoding method, but the decoder 12 in this embodiment does not limit the decoding method to a specific method, and any method may be used. . As long as the encoding is performed by the encoder 301 in the transmission apparatus as illustrated in FIG. 13, the decoding can be performed using the reliability information of each bit output from the parallel-serial conversion unit 11. Any method may be used.

  As described above, according to the fourth embodiment, due to imperfection of the quadrature modulator and the quadrature demodulator, high accuracy can be achieved even in an environment where the signals of the subcarriers in the upper sideband and the lower sideband interfere with each other. It is possible to receive. Furthermore, the transmission characteristics can be further improved by performing soft decision decoding on the signal that has been encoded in the transmission apparatus.

  Further, two maximum likelihood estimators are provided for each sideband subcarrier signal extracted by the signal separator 8 by combining signals received by the received signal combiner 6 with upper and lower sideband subcarriers. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Fifth embodiment)
FIG. 3 shows a configuration example of a wireless reception apparatus according to the fifth embodiment of the present invention.

  Also in the fifth embodiment, the received signals of two subcarriers (kth and -kth subcarriers) at symmetrical positions on the frequency axis around the center frequency of a plurality of subcarriers, After combining by the received signal combining unit 6, the signal separating unit 8 extracts signals transmitted on the kth and -kth subcarriers from the combined signal, respectively, and extracts the two maximum likelihood estimation units 9, 10 is the same as the third embodiment or the fourth embodiment in that the maximum likelihood bit or reliability information of each bit is obtained for each subcarrier signal.

  The fifth embodiment is different from the third embodiment and the fourth embodiment in that reception is performed on a transmission signal generated using a plurality of encoders.

  First, a transmission device corresponding to the wireless reception device of FIG. 3 will be described with reference to FIG. FIG. 14 shows a configuration example of the transmission apparatus. In FIG. 14, the same parts as those in FIG. 13 are denoted by the same reference numerals.

  The information signal to be transmitted is input to the parallel-serial conversion unit 201 and converted into the same number of parallel signals as the number of encoders 401a to 401d. Here, the serial / parallel conversion may be performed by converting each bit into a parallel signal as in the case of the serial / parallel conversion unit 201 in the transmission apparatus shown in FIG. 12 described in the first embodiment. It doesn't matter. As long as the number of parallel signals is the same as that of the encoders 401a to 401d, the number of bits is a predetermined number, and the wireless reception device has a known number of bits, the conversion may be applied in any number of bits.

  FIG. 14 shows an example in which the same number of encoders 401a to 401d as the number of spatial multiplexing are used. For example, as shown in FIG. 15, the number of encoders and the number of spatial multiplexing are not the same. It doesn't matter. As shown in FIG. 15, the number of encoders may be smaller than the spatial multiplexing number, or the number of encoders may be larger than the spatial multiplexing number. The number of encoders is a predetermined number, and any number may be used as long as the wireless reception device in the present embodiment is known.

  The signal distributor 402 distributes the signals encoded by a plurality of encoders (here, encoders 401a to 401d as shown in FIG. 14) to the same number of parallel signals as the spatial multiplexing number. Here, the signal distribution is in a predetermined order, and any order may be used as long as the wireless reception apparatus of the present embodiment knows the distribution order.

  The output signals of the signal distribution unit 402 are rearranged in the interleaving units 302a to 302d, further modulated for each subcarrier in the modulators 202a to 202d, and inverse Fourier transform units 203a to 203d. In step 4, an inverse Fourier transform is performed. GI adding sections 204a to 204d add guard intervals to the outputs from the inverse Fourier transform sections 203a to 203d, and transmit the signals from the antennas 206a to 206d via the radio sections 205a to 205d. Such a process is the same as in the first to fourth embodiments.

  Next, the operation of the radio reception apparatus when a signal is generated using a plurality of encoders in such a transmission apparatus will be described with reference to FIG. In FIG. 3, the same parts as those in FIG.

  Receiving antennas 1a to 1d, radio units 2a to 2d, GI removing units 3a to 3d, Fourier transform units 4a to 4d, propagation path estimating unit 5, received signal combining unit 6, weight calculating unit 7, and signal separating unit 8 Since it is the same as that of 3rd Embodiment and 4th Embodiment, description is abbreviate | omitted.

  In the maximum likelihood estimator 9 and the maximum likelihood estimator 10, when the decoding scheme used in the subsequent decoders 22 a to 22 d is hard decision decoding, the same operation as in the third embodiment is performed, and the decoders 22 a to 22 d When soft decision decoding is performed, the same operation as that of the fourth embodiment is performed, and thus the detailed description of the maximum likelihood estimation unit is also omitted.

  The outputs of the maximum likelihood estimator 9 and the maximum likelihood estimator 10 are input to the signal distributor 21, and each signal is distributed to the decoders 22a to 22d. Here, the number of decoders is the same as the number of encoders used in the transmission apparatus, and the radio reception apparatus in FIG. 3 corresponds to the transmission apparatus using four encoders as shown in FIG. Therefore, an example using four decoders is shown. However, the number of decoders in this embodiment is not limited to four. The number of decoders can be changed according to a predetermined number of encoders.

  In addition, the signal distribution order in the signal distribution unit 21 is determined according to the signal distribution order applied by the signal distribution unit 402 and the interleaving units 302a to 302d of the transmission apparatus, and the replacement order, and the encoders 401a to 401a. Distribution is performed so that the order of signals output by 401d matches the order of signals input to the decoders 22a to 22d. Here, as described in the description of the transmitting apparatus, the distribution order and replacement order of the signals applied by the signal distributing unit 402 and the interleave units 302a to 302d of the transmitting apparatus are predetermined orders. The wireless reception device in the embodiment is also known.

  The signals distributed as described above are input to the decoders 22a to 22d, respectively, and are decoded. Since the decoding in each decoder is as described in the third embodiment or the fourth embodiment, the details are omitted.

  The signals decoded by the decoders 22a to 22d are input to the parallel / serial conversion unit 11 and converted into serial signals. At this time, the bit unit to be converted into the serial signal is the bit unit used in the serial / parallel conversion unit 201 of the transmission apparatus. As described in the description of the transmitting apparatus, this number of bits is also known in the radio receiving apparatus according to the present embodiment because conversion is applied in units of bits determined in advance.

  As described above, according to the fifth embodiment, due to imperfection of the quadrature modulator and the quadrature demodulator, high accuracy can be achieved even in an environment where the signals of the subcarriers in the upper sideband and the lower sideband interfere with each other. It is possible to receive. Furthermore, since encoding is applied in the transmission device, transmission characteristics can be further improved. Since decoding is performed in parallel using a plurality of decoders, decoding can be performed at a higher speed when compared at the same clock speed than when a single decoder is used.

  Also, the received signal combining unit 6 combines signals received by upper and lower sideband subcarriers, and two maximum likelihood estimations for each sideband subcarrier signal extracted by the signal separation unit 8. Since the estimation is performed using the units 9 and 10, it is possible to prevent the calculation load of the maximum likelihood estimation units 9 and 10 from increasing.

(Sixth embodiment)
A radio receiving apparatus according to the sixth embodiment of the present invention will be described.

  FIG. 4 shows an example of the configuration of a wireless reception apparatus according to the sixth embodiment. In FIG. 4, the same parts as those in FIG.

  As shown in FIG. 1, the radio receiving apparatus shown in FIG. 4 includes a plurality of (for example, four) receiving antennas 1a to 1d and a plurality of (for example, four here) connected to the receiving antennas 1a to 1d. Radio units 2a to 2d, a plurality of (here, for example, four) GI (guard interval) removal units 3a to 3d connected to each of the plurality of radio units 2a to 2d, and a plurality of GI removal units 3a to 3d. A plurality (four in this example) of Fourier transform units 4a to 4d connected to each of 3d, a propagation path estimation unit 5 connected to the plurality of Fourier transform units 4a to 4d, and a plurality of Fourier transform units 4a to 4d A reception signal coupling unit 6 connected to the channel, a weight calculation unit 7 connected to the propagation path estimation unit 5, a signal separation unit 8 connected to the reception signal coupling unit 6 and the weight calculation unit 7, and a signal separation unit 8 and way Two maximum likelihood estimation unit 9, 10 is connected to the calculation unit 7 includes two parallel-to-serial conversion section 11 connected to the maximum likelihood estimation unit 9, 10.

  4 differs from FIG. 1 in that the maximum likelihood estimation unit 9 and the maximum likelihood estimation unit 10 are connected to a reference signal generation unit 32 and the propagation path estimation unit 5 is connected to the reference signal generation unit 32. Is a point.

As described in the first embodiment, the propagation path estimation unit 5 estimates the propagation path matrix H _k of Expression (7). At this time, if an error occurs in the propagation path estimation result, an error also occurs in the subsequent weight calculation unit 7, and the signal separation accuracy in the signal separation unit 6 and the estimation accuracy in the maximum likelihood estimation units 9 and 10 are deteriorated. .

In general, the accuracy of propagation path estimation can be increased by increasing the number of known signals for propagation path estimation. However, excessive transmission of signals other than information degrades transmission efficiency. Cannot be sent.
Therefore, in the sixth embodiment, a reference signal is generated using the maximum likelihood estimation result of the information signal, propagation path estimation is performed again using the reference signal as a known signal, and the obtained propagation path estimation result is used. After recalculating the weights and performing signal separation again, maximum likelihood estimation is also performed.

  At this time, until the maximum likelihood estimation is performed by the maximum likelihood estimation units 9 and 10, the description is omitted because it is the same as that of the first embodiment.

  From the results of the maximum likelihood estimation units 9 and 10, the reference signal generation unit 32 can generate symbols of signals that are spatially multiplexed on each subcarrier and transmitted. As a result, an unknown information signal can be treated as a known signal, and it can be considered that the known signals for channel response estimation are virtually increased.

The propagation path estimation unit 5 again estimates the propagation path matrix using the propagation path estimation known signal thus increased and the reception vector R _k (m) of the section in which the signal is transmitted.

  Using the new propagation path matrix estimated as described above, the weight calculation unit 7 calculates the weight as shown in the equations (11) and (20). Similarly, using the newly calculated propagation path matrix and the newly calculated weight, the inverse matrix of the error correlation matrix represented by the equation (14) or the equation (21) is calculated. It outputs to the maximum likelihood estimation part 9 and 10.

  Using the newly obtained weights, the signal separation unit 8 extracts the signals of the upper sideband and the lower sideband, respectively, and the result is the inverse of the correlation matrix of the error newly calculated by the weight calculation unit 7. The maximum likelihood estimators 9 and 10 estimate signals in each sideband using the matrix.

  The signal reception process is completed by converting the above result into a serial signal by the parallel-serial converter 11 as in the first embodiment.

  By using the propagation path matrix having a high estimation accuracy obtained in this way, the estimation accuracy of the maximum likelihood estimation units 9 and 10 is also high, so that the accuracy of the reference signal in the reference signal generation unit 32 is also high. For this reason, it is possible to further improve the estimation accuracy by repeating the same processing several times. Therefore, as described above, the parallel-serial conversion unit 11 may perform parallel-serial conversion after repeating the generation of the reference signal, propagation path estimation, weight calculation, signal separation, and maximum likelihood estimation several times.

  As described above, according to the sixth embodiment, in the environment where the subcarrier signals of the upper sideband and the lower sideband interfere with each other due to imperfections of the quadrature modulator and the quadrature demodulator. Can be received with high accuracy. Also, by performing channel estimation again using the maximum likelihood estimation result, the accuracy of channel estimation can be improved, and reception performance can be improved.

  In addition, two maximum likelihood estimators are provided for each subband signal in each sideband extracted by the signal separator 8 by combining the signals received by the subcarriers in the upper and lower sidebands in the received signal combiner 6. Therefore, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Seventh embodiment)
A radio reception apparatus according to the seventh embodiment will be described.

The configuration of the wireless reception apparatus according to the seventh embodiment is the same as that shown in FIG. The seventh embodiment differs from the sixth embodiment in that the known symbols for channel matrix estimation are insufficient and the channel matrix H _k shown in Equation (7) cannot be estimated. It is a point to perform the operation.

In the description of the first embodiment, it has been described that the propagation path matrix H _k represented by the equation (7) is estimated by the propagation path estimation unit 5. In MIMO-OFDM transmission, a known signal for estimating a propagation path matrix is designed to be able to estimate a propagation path matrix for each spatially multiplexed signal. There is a case where it is not designed to be able to estimate an interference component between subcarriers caused by the cause. Actually, it is impossible to estimate an interference component between subcarriers using the propagation path matrix estimation known symbol (LP) in the IEEE802.11a frame format shown in FIG.

Therefore, in the seventh embodiment, it is assumed that there is no interference between the subcarriers in the upper and lower sidebands at the time of the initial propagation path matrix estimation, and the above equation (7) is replaced with the following equation (27).

By assuming as described above, H _k ′ can be estimated if it is designed so that a propagation path of a signal in which known symbols for propagation path estimation are spatially multiplexed can be estimated. At this time, the interference component between the subcarriers is treated as a thermal noise signal, and the greater the degree of imperfection of the quadrature modulator / demodulator, the larger the interference component, so the accuracy of channel estimation also deteriorates.

  Using the propagation path matrix estimation result obtained as described above, the weight calculator 7 extracts the subcarrier signal of the upper sideband as in the first embodiment and the second embodiment. The weight for extracting the band signal is obtained, and the inverse matrix of the error correlation matrix of Equation (14) or Equation (21) is obtained.

  Using the weights obtained above, the signal of the upper and lower sidebands is separated by the signal separation unit 8 as described in the first embodiment, and the first likelihood estimation unit 9 and the maximum likelihood estimation unit 10 Maximum likelihood estimation is performed as described in the embodiment.

  Next, based on the results obtained by the maximum likelihood estimator 9 and the maximum likelihood estimator 10, a reference signal generator 32 generates a known signal as in the sixth embodiment. As described above, the number of known signals is increased by using the known signals created using the estimated information signals, so that the propagation path matrix of the interference component between the subcarriers can be estimated. Therefore, at the time of the second propagation path estimation, the propagation path estimation is performed on the assumption that the received signal is formed as in Expression (14) instead of Expression (27).

  The point that the weight is recalculated using the propagation path estimation result obtained as described above, and the signal separation and the maximum likelihood estimation are performed again is the same as in the sixth embodiment. Similarly to the sixth embodiment, the process from the generation of the reference signal using the maximum likelihood estimation result to the maximum likelihood estimation may be repeated a plurality of times.

  As described above, according to the seventh embodiment, due to imperfection of the quadrature modulator and the quadrature demodulator, reception is possible with high accuracy even in an environment where the signals of the subcarriers in the upper sideband and the lower sideband interfere with each other. It becomes possible to do. At this time, according to the present embodiment, even if a known signal for propagation path estimation is not designed so that the propagation path of the interference component between the subcarriers in the upper and lower sidebands can be estimated, reception with high accuracy is possible. It becomes possible to do.

  In addition, two maximum likelihood estimators are provided for each subband signal in each sideband extracted by the signal separator 8 by combining the signals received by the subcarriers in the upper and lower sidebands in the received signal combiner 6. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Eighth embodiment)
A radio reception apparatus according to the eighth embodiment will be described with reference to FIG. In FIG. 5, the same parts as those in FIG.

  5 is different from FIG. 4 in that the reference signal generation unit 32 generates a reference signal using a decoding result in the decoder 12.

  When the reference signal is generated using the determined information signal as in the sixth embodiment and the seventh embodiment, if an error occurs in the determination result, the actually transmitted signal is different from the reference signal. For this reason, if there are many judgment errors, the reliability of the reference signal is lowered, and the accuracy of propagation path estimation is not improved so much. The output of the decoder 12 is error-corrected, and generally has higher reliability than the determination result before decoding. Therefore, it is possible to generate a reference signal with higher reliability by generating a reference signal using a signal after error correction.

  Since the eighth embodiment is the same as the third embodiment or the fourth embodiment until the output of the decoder 12 shown in FIG. 5 is obtained, the description thereof is omitted.

  Next, a method for generating a reference signal using the output from the decoder 12 will be described. In FIG. 5, it is assumed that the transmission apparatus has the configuration shown in FIG. 13, and the decoder 12 estimates a signal input to the encoder 301 in FIG. Therefore, in order to generate a reference signal using the signal, encoding is performed, serial-parallel conversion and interleaving are performed according to the same rule as the method used in the transmission device, and the same method as that used in the transmission device is used. It is necessary to perform modulation.

  For example, in the transmission apparatus of FIG. 13, since the spatial multiplexing number is “4”, the reference signal generation unit 32 includes an encoder, a serial-parallel conversion unit, four interleave units, and four modulators. Yes. The encoder of the reference signal generation unit 32 encodes the signal output from the decoder 12, and the serial-parallel conversion unit converts the signal output from the encoder into a spatial multiplexing number (in the case of the transmission device in FIG. 13). , “4”) and the same number of sequences (here, 4 sequences) of signals, and each of the four interleave units is input with four sequences of signals output from the serial-parallel converter. Signals are rearranged, and each modulation unit connected to each interleaving unit modulates the signal output from the interleaving unit with the same modulation scheme as that of the modulators 202a to 202d of the transmission apparatus in FIG. To the propagation path estimation unit 5.

  The propagation path estimation unit 5 uses this signal as a reference signal and re-estimates the propagation path matrix.

  On the other hand, it is possible to cause the decoder 12 to output the encoded signal in anticipation of generating the reference signal using the determination result. In this case, it is not necessary to apply encoding in the reference signal generation unit 32, and the calculation load can be reduced.

  That is, the reference signal generation unit 32 may include a serial-parallel conversion unit, four interleave units, and four modulators excluding the encoder. The serial-to-parallel converter of the reference signal generator 32 converts the signal output from the decoder 12 into the same number of sequences (four sequences in this case) as the spatial multiplexing number (“4” in the case of the transmission device in FIG. 13). The signal is converted into a signal, and each of the four interleave units receives the four series of signals output from the serial-parallel converter, rearranges the signals, and modulates each interleave unit connected to each interleave unit. The unit modulates the signal output from the interleave unit with the same modulation scheme as that of the modulators 202a to 202d of the transmission apparatus in FIG. 13, and outputs the modulated signal to the propagation path estimation unit 5 in FIG.

  The point of performing weight calculation using the propagation path matrix recalculated as described above, separating the signals, and performing maximum likelihood estimation by the maximum likelihood estimation units 9 and 10 is the same as in the sixth embodiment or the seventh embodiment. This is the same as the embodiment, and a detailed description is omitted. However, since the present embodiment assumes that encoding is performed, the result output again by the maximum likelihood estimation units 9 and 10 is converted again into a serial signal by the parallel-serial conversion unit 11, and is decoded by the decoder 12. Error correction is performed.

  A reference signal is generated using the error correction result as described above, the channel matrix is re-estimated, and a series of procedures leading to re-decoding is performed in the same manner as in the sixth and seventh embodiments. You may repeat it.

  According to the eighth embodiment, due to imperfection of the quadrature modulator and the quadrature demodulator, reception is possible with high accuracy even in an environment where the signals of the subcarriers in the upper sideband and the lower sideband interfere with each other. It becomes possible to do. Further, according to the eighth embodiment, the estimation accuracy of the propagation path matrix can be improved by re-estimating the propagation path matrix, and the propagation path estimation of the interference component between the subcarriers in the upper and lower sidebands can be performed. Therefore, even when a known signal for propagation path estimation is not designed so that the propagation path matrix can be estimated, the propagation path matrix can be estimated, and reception with high accuracy is possible. In this case, the reliability of the reference signal can be increased by generating the reference signal using the error-corrected signal.

  In addition, two maximum likelihood estimators are provided for each subband signal in each sideband extracted by the signal separator 8 by combining the signals received by the subcarriers in the upper and lower sidebands in the received signal combiner 6. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation units 9, 10 from increasing.

(Ninth embodiment)
A wireless reception apparatus according to the ninth embodiment of the present invention will be described with reference to FIG. In FIG. 6, the same parts as those in FIGS. 3 and 5 are denoted by the same reference numerals.

  6 is different from FIG. 5 in that the reference signal generation unit 32 in FIG. 6 generates a reference signal using the decoding results in the plurality of decoders 22a to 22d.

  The ninth embodiment is the same as the fifth embodiment until parallel-serial conversion is performed on the outputs of a plurality of (for example, four) decoders 22a to 22d (see FIG. 3). Omitted. The process until the reference signal generation unit 32 generates the reference signal using the decoded signal is the same as that in the eighth embodiment. However, in the ninth embodiment, as shown in FIGS. 14 and 15, since signals transmitted using a plurality of encoders are received, the decoders 22 a to 22 d output the signals before encoding. In this case, the reference signal generation unit 32 includes the same number of encoders as the number of encoders used in the transmission device, a signal distribution unit, and the same number of interleave units and modulators as the signals to be spatially multiplexed.

  Here, the encoder, the interleave unit, and the modulator use the same system as that used in the transmission apparatus, as in the eighth embodiment.

  The encoder of the reference signal generation unit 32 encodes the signal output from each of the decoders 22a to 22d, and the signal distribution unit converts the signal output from the encoder into a spatial multiplexing number (for example, the transmission apparatus of FIG. 14). In this case, the signal is converted into a signal of the same number (4 in this case) as “4”), and each of the four interleaving units is input with the four sequences of signals output from the signal distributor. , Each signal is rearranged, and each modulation unit connected to each interleave unit modulates the signal output from the interleave unit with the same modulation scheme as the modulators 202a to 202d of the transmission device in FIG. It outputs to the propagation path estimation part 6 of FIG.

  When the decoders 22a to 22d output the encoded signals, the reference signal generation unit 32 includes a signal distribution unit, the same number of interleave units and modulators as the spatial multiplexing number, excluding the encoder. . Similar to the case where the decoder outputs a signal before encoding, each unit is operated in the same manner as the transmission apparatus.

  In this case, the signal distribution unit of the reference signal generation unit 32 converts the signal output from each of the decoders 22a to 22d into the same number of sequences as the spatial multiplexing number (for example, “4” in the case of the transmission apparatus in FIG. 14). The signals are converted into (in this case, four series) signals, and the four series of signals output from the signal distribution section are input to each of the four interleave sections, and the signals are rearranged, and each interleave section is arranged. 14 modulates the signal output from the interleave unit in the same modulation scheme as that of the modulators 202a to 202d of the transmission apparatus in FIG. 14, and outputs the modulated signal to the propagation path estimation unit 6 in FIG.

  As in the sixth to eighth embodiments, the propagation path estimation unit 5 recalculates the propagation path matrix using the reference signal generated as described above, and the weight calculation section 7 calculates the weight matrix and the error. The correlation matrix is recalculated, the signal separation unit 8 re-separates the signals in the upper and lower sidebands, the maximum likelihood estimation unit 9 and the maximum likelihood estimation unit 10 perform maximum likelihood estimation again, and performs re-decoding. Further, similarly to the sixth to eighth embodiments, the process from the generation of the reference signal to the decoding may be repeated a plurality of times.

  FIG. 6 illustrates a case where the outputs of the plurality of decoders 22a to 22d are input to the reference signal generation unit 32 and the reference signal is generated using the outputs of the plurality of decoders 22a to 22d. As shown, the output of the parallel-serial converter 11 may be input to the reference signal generator 32. In this case, the reference signal generation unit 32 includes a serial-to-parallel conversion unit, the same number of encoders as the number of encoders used in the transmission device, a signal distribution unit, the same number of interleaving units as the spatial multiplexing number, and a modulator. The serial-to-parallel converter is also operated according to the same rules as those used in the transmitting apparatus, as in other parts.

  As described above, according to the present invention, due to imperfection of the quadrature modulator and the quadrature demodulator, it is possible to receive with high accuracy even in an environment where the signals of the subcarriers of the upper sideband and the lower sideband interfere with each other. It becomes possible. Further, according to the present embodiment, it is possible to improve the estimation accuracy of the propagation path matrix by re-estimating the propagation path matrix and to perform propagation path estimation of the interference component between the subcarriers in the upper and lower sidebands. Even when a known signal for propagation path estimation is not designed, the propagation path matrix can be estimated and reception can be performed with high accuracy. In this case, the reliability of the reference signal can be increased by generating the reference signal using the signal after error correction, and decoding is performed in parallel using a plurality of decoders. Compared to the case where a device is used, decoding can be performed at the same clock speed.

  In addition, two maximum likelihood estimators are provided for each subband signal in each sideband extracted by the signal separator 8 by combining the signals received by the subcarriers in the upper and lower sidebands in the received signal combiner 6. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

(Tenth embodiment)
FIG. 8 shows a configuration example of a wireless reception apparatus according to the tenth embodiment of the present invention.

  As illustrated in FIG. 8, the wireless reception device according to the tenth embodiment is more than a plurality of (here, for example, eight) reception antennas 1 a to 1 h and the number of reception antennas (here, “8”). A small number (in this case, for example, four) of radio units 2a to 2d, a switch 41 for connecting the receiving antennas 1a to 1h and the radio units 2a to 2d, and a GI connected to each of the radio units 2a to 2d ( Guard interval) removal units 3a to 3d, Fourier transform units 4a to 4d connected to GI removal units 3a to 3d, propagation path estimation unit 5 connected to Fourier transform units 4a to 4d, and Fourier transform unit 4a Received signal combining unit 6 connected to ˜4d, weight calculating unit 7 connected to propagation path estimating unit 5, signal separating unit 8 connected to received signal combining unit 6 and weight calculating unit 7, and signal separation Part 8 and Two maximum likelihood estimation unit 9, 10 which are connected to the eight calculation unit 7, and two parallel-serial conversion unit 11 connected to the maximum likelihood estimation unit 9, and an antenna selector 42.

  In FIG. 8, the same parts as those in FIG.

  8 is different from FIG. 1 in that the radio receiving apparatus in FIG. 8 can switch the antenna for reception.

  In a multipath propagation environment, since a plurality of radio waves arrive at each receiving antenna with different phases, generally different fading occurs for each receiving antenna, and the signal power and phase received by each receiving antenna differ. Therefore, if the number of reception antennas is small, the reception characteristics of all the antennas are deteriorated, and the transmitted signal may not be decoded.

  It is known that such deterioration can be prevented by increasing the number of receiving antennas. However, if the number of antennas is increased and the number of radio units connected to the antennas is increased accordingly, the cost of the apparatus increases. Therefore, the number of antennas and radio units cannot be increased more than necessary.

  Therefore, a method is effective in which the number of radio units is kept to a minimum, more receiving antennas are installed, a receiving antenna with good reception characteristics is selected, and the radio unit is connected. The above method is a method generally used in small wireless devices such as mobile phones, and its effectiveness is widely known.

  The same technique can be applied to the wireless reception device according to the tenth embodiment. As a selection criterion, there is a method of selecting an antenna based on received power as in a generally used method. The antenna selection unit 42 in FIG. 8 selects the antennas in the descending order of reception power from all reception antennas based on the reception power measured by the radio units 2a to 2d, thereby calculating the total reception power obtained by the radio unit. Can be maximized.

  Further, when the number of reception antennas increases, it becomes difficult to measure and compare the power of all reception antennas. In this case, there is a method of selecting a necessary number of antennas from reception antennas exceeding the reference power. With this method, the total received power cannot be maximized, but at least power above the reference can be ensured, and a sub-optimal combination can be easily selected as compared with a method that compares the received power of all antennas.

  As described above, the method of selecting the reception antenna using the reception power has been described, but the antenna selection method in the present invention is not limited to the above method. Other antenna selection methods include methods that use channel capacity, correlation of fading, delay spread of channel response, etc. as criteria. Any selection method can be used as long as the antenna used for reception is selected from multiple antennas. You may use.

  The operations after the radio units 2a to 2d are the same as those in the first embodiment or the second embodiment, and thus the details are omitted.

  As the configuration of the tenth embodiment, the first and second embodiments have been described with reference to FIG. 8 as a system that extends to select an antenna to be used for reception from a plurality of reception antennas. However, the tenth embodiment can also be applied to the connection of the receiving antenna and the radio unit in the third to ninth embodiments in the same manner as described above, and the same effect can be obtained.

  As described above, according to the present invention, due to imperfection of the quadrature modulator and the quadrature demodulator, it is possible to receive with high accuracy even in an environment where the signals of the subcarriers of the upper sideband and the lower sideband interfere with each other. It becomes possible. Furthermore, reception performance can be further improved by decoding the signal that has been encoded by the transmission apparatus. At this time, according to the tenth embodiment, reception performance can be further improved by selecting an antenna having good reception characteristics from a plurality of reception antennas. Further, two maximum likelihood estimators are provided for each sideband subcarrier signal extracted by the signal separator 8 by combining signals received by the received signal combiner 6 with upper and lower sideband subcarriers. Since the estimation is performed using 9, 10, it is possible to prevent the calculation load of the maximum likelihood estimation unit from increasing.

  Other known signals for channel estimation so that the channel matrix estimation accuracy can be improved by re-estimating the channel matrix and the interference channel between the upper and lower sideband subcarriers can be estimated. Even if it is not designed, it is possible to receive with high accuracy. In this case, the reliability of the reference signal can be increased by generating the reference signal using the error-corrected signal. Furthermore, by performing decoding in parallel using a plurality of decoders, decoding can be performed at a higher speed than when a single decoder is used.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying the constituent elements without departing from the scope of the invention in the implementation stage. In addition, various inventions can be formed by appropriately combining a plurality of components disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, constituent elements over different embodiments may be appropriately combined.

The figure which showed the structural example of the radio | wireless receiver concerning 1st and 2nd embodiment. The figure which showed the structural example of the radio | wireless receiver concerning 3rd and 4th embodiment. The figure which showed the structural example of the radio | wireless receiving apparatus concerning 5th Embodiment. The figure which showed the structural example of the radio | wireless receiver concerning 6th and 7th embodiment. The figure which showed the structural example of the radio | wireless receiving apparatus concerning 8th Embodiment. The figure which showed the structural example of the radio | wireless receiving apparatus concerning 9th Embodiment. The figure which showed the structural example of the radio | wireless receiving apparatus concerning 9th Embodiment. The figure which showed the structural example of the radio | wireless receiving apparatus concerning 10th Embodiment. The figure for demonstrating the guard interval in OFDM transmission. The figure for demonstrating that it can avoid intersymbol interference by a guard interval in OFDM transmission. The figure which showed the frame format in IEEE 802.11a. The figure which showed the structural example of the transmitter corresponding to the radio | wireless receiver shown in FIG.1, FIG4 and FIG.8. The figure which showed the structural example of the transmitter corresponding to the radio | wireless receiver shown in FIG.2 and FIG.5. The figure which showed the structural example of the transmitter corresponding to the radio | wireless receiver shown in FIG.3, FIG6 and FIG.7. The figure which showed the other structural example of the transmitter corresponding to the radio | wireless receiver shown in FIG.3, FIG6 and FIG.7. The figure which shows the response | compatibility of the transmission bit and transmission symbol in 64QAM (constellation).

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1a-1d ... Reception antenna, 2a-2d ... Radio | wireless part, 3a-3d ... GI (guard interval) removal part, 4a-4d ... Fourier-transform part, 5 ... Propagation path estimation part, 6 ... Received signal coupling | bond part, 7 ... Weight calculation unit, 8 ... signal separation unit, 9, 10 ... maximum likelihood estimation unit, 11 ... parallel to serial conversion unit, 12 ... decoder, 21 ... signal distribution unit, 22a-22d ... decoder, 32 ... reference signal generation unit , 42 ... Antenna selection unit

Claims (24)

An antenna for receiving an OFDM (Orthogonal Frequency Division Multiplexing) signal including a plurality of subcarriers;
For each of two subcarriers that are symmetrical on the frequency axis around the center frequency of the plurality of subcarriers of the received OFDM signal, the signals received on each of the two subcarriers are combined and combined. A signal combining means for obtaining a signal;
First extraction means for extracting, from each of the combined signals, a signal component transmitted by a subcarrier in a frequency band higher than the center frequency among the two subcarriers;
Second extraction means for extracting, from each of the combined signals, a signal component transmitted by a subcarrier in a frequency band lower than the center frequency among the two subcarriers;
First maximum likelihood estimation means for performing maximum likelihood estimation of a signal transmitted on the subcarrier from a signal component transmitted on the high frequency band subcarrier extracted from the combined signal;
Second maximum likelihood estimation means for performing maximum likelihood estimation of a signal transmitted on the subcarrier from a signal component transmitted on the subcarrier in the low frequency band extracted from the combined signal;
A radio receiving apparatus comprising: A plurality of receiving antennas for receiving an OFDM (Orthogonal Frequency Division Multiplexing) signal including a plurality of subcarriers;
For each of two subcarriers that are symmetrical on the frequency axis around the center frequency of the plurality of subcarriers of the received OFDM signal, the signals received on each of the two subcarriers are combined and combined. A signal combining means for obtaining a signal;
For each subcarrier, the signal received by the subcarrier in the signal received by the subcarrier and the other subcarrier that is symmetrical to the subcarrier on the frequency axis centered on the center frequency are transmitted. First calculating means for obtaining a propagation path matrix of the received signal;
Using the propagation path matrix obtained for each subcarrier, a first signal component for extracting a signal component transmitted by a subcarrier in a higher frequency band than the center frequency from the two subcarriers from each combined signal. Second calculating means for obtaining a matrix and a second matrix for extracting a signal component transmitted by a subcarrier in a frequency band lower than the center frequency;
First extraction means for multiplying the combined signal by the first matrix and extracting a signal component transmitted from the high frequency band subcarrier from the combined signal;
Second extraction means for multiplying the combined signal by the second matrix and extracting a signal component transmitted from the combined signal on the sub-carrier in the low frequency band;
A first inverse matrix that is an inverse matrix of an error correlation matrix between a signal component transmitted from the subcarrier in the high frequency band extracted from the combined signal and a signal transmitted from the subcarrier; The inverse matrix of the correlation matrix of the error between the signal component transmitted from the sub-carrier in the low frequency band extracted from the combined signal and the signal transmitted from the sub-carrier is calculated. Third calculating means for calculating an inverse matrix of 2 using the second matrix;
A first maximum likelihood for estimating a signal transmitted on the subcarrier from a signal component transmitted on the subcarrier in the high frequency band extracted from the combined signal using the first inverse matrix; An estimation means;
A second maximum likelihood for estimating a signal transmitted on the subcarrier from a signal component transmitted on the subcarrier in the low frequency band extracted from the combined signal, using the second inverse matrix; An estimation means;
A radio receiving apparatus comprising:   3. The radio reception according to claim 2, further comprising decoding means for performing error correction decoding on the signal transmitted by each subcarrier obtained by the first and second maximum likelihood estimation means. apparatus.   3. The apparatus according to claim 2, further comprising a plurality of decoding means for performing error correction decoding on the signals transmitted by the subcarriers obtained by the first and second maximum likelihood estimation means. Wireless receiver.   The radio receiving apparatus according to claim 3, wherein the decoding means performs hard decision decoding.   The radio receiving apparatus according to claim 3, wherein the decoding means performs soft decision decoding.   The radio receiving apparatus according to claim 4, wherein the plurality of decoding units perform hard decision decoding.   5. The radio reception apparatus according to claim 4, wherein the plurality of decoding units perform soft decision decoding. Reference signal generation means for generating a reference signal used for propagation path estimation based on the signals transmitted on the respective subcarriers obtained by the first and second maximum likelihood estimation means,
The first calculation means recalculates the propagation path matrix for each carrier using the reference signal,
The second calculation means uses a transmission path matrix recalculated for each subcarrier, and uses a first matrix for extracting a signal component transmitted on the high frequency band subcarrier from each combined signal; , Again calculating a second matrix for extracting signal components transmitted on the subcarriers in the low frequency band,
The first extraction unit multiplies the combined signal by the recalculated first matrix to extract again a signal component transmitted on the high frequency band subcarrier from the combined signal,
The second extraction means multiplies each combined signal by the recalculated second matrix to extract again the signal component transmitted on the low frequency band subcarrier from the combined signal,
The third calculation means recalculates the first inverse matrix using the recalculated first matrix and uses the second matrix calculated again to calculate the second inverse matrix. And recalculate
The first maximum likelihood estimator uses the recalculated first inverse matrix to reestimate the signal transmitted on the high frequency band subcarrier again,
3. The radio reception apparatus according to claim 2, wherein the second maximum likelihood estimation unit reestimates a signal transmitted on the low frequency band subcarrier again using the recalculated second inverse matrix. 4. . Based on the signal obtained as a result of decoding by the decoding means, further comprising a reference signal generating means for generating a reference signal used for propagation path estimation,
The first calculation means recalculates the propagation path matrix for each carrier using the reference signal,
The second calculation means uses a propagation path matrix recalculated for each subcarrier, and uses a first matrix for extracting a signal component transmitted by the high frequency band subcarrier from each combined signal; , Again calculating a second matrix for extracting signal components transmitted on the subcarriers in the low frequency band,
The first extraction unit multiplies the combined signal by the recalculated first matrix to extract again a signal component transmitted on the high frequency band subcarrier from the combined signal,
The second extraction means multiplies the combined signal by the recalculated second matrix to extract again the signal component transmitted on the low frequency band subcarrier from the combined signal,
The third calculation means recalculates the first inverse matrix using the recalculated first matrix and uses the second matrix calculated again to calculate the second inverse matrix. And recalculate
The first maximum likelihood estimator uses the recalculated first inverse matrix to reestimate the signal transmitted on the high frequency band subcarrier again,
The second maximum likelihood estimation means uses the re-calculated second inverse matrix to reestimate a signal transmitted on the low frequency band subcarrier again,
The radio reception apparatus according to claim 3, wherein the decoding means performs error correction decoding again on a signal transmitted on each subcarrier obtained as a result of maximum likelihood estimation again. An OFDM (Orthogonal Frequency Division Multiplexing) signal including the plurality of subcarriers is spatially multiplexed and transmitted.
The reference signal generation means includes
Encoding means for encoding a signal obtained as a result of decoding by the decoding means;
A number of modulation means equal to the spatial multiplexing number;
Means for converting and rearranging the signals encoded by the encoding means into signals having a number of sequences equal to the spatial multiplexing number, in order to input the signals to the modulation means;
The wireless receiver according to claim 10, comprising: An OFDM (Orthogonal Frequency Division Multiplexing) signal including the plurality of subcarriers is spatially multiplexed and transmitted.
The reference signal generation means includes
A number of modulation means equal to the spatial multiplexing number;
Means for converting and rearranging the signal obtained as a result of decoding by the decoding means to a signal having a number of sequences equal to the spatial multiplexing number;
The wireless receiver according to claim 10, comprising: Based on signals obtained as a result of decoding by the plurality of decoding means, further comprising reference signal generating means for generating a reference signal used for propagation path estimation,
The first calculation means recalculates the propagation path matrix for each carrier using the reference signal,
The second calculation means uses a transmission path matrix recalculated for each subcarrier, and uses a first matrix for extracting a signal component transmitted on the high frequency band subcarrier from each combined signal; , Again calculating a second matrix for extracting signal components transmitted on the subcarriers in the low frequency band,
The first extraction unit multiplies the combined signal by the recalculated first matrix to extract again a signal component transmitted on the high frequency band subcarrier from the combined signal,
The second extraction means multiplies the combined signal by the recalculated second matrix to extract again the signal component transmitted on the low frequency band subcarrier from the combined signal,
The third calculation means recalculates the first inverse matrix using the recalculated first matrix and uses the second matrix calculated again to calculate the second inverse matrix. And recalculate
The first maximum likelihood estimator uses the recalculated first inverse matrix to reestimate the signal transmitted on the high frequency band subcarrier again,
The second maximum likelihood estimation means uses the re-calculated second inverse matrix to reestimate a signal transmitted on the low frequency band subcarrier again,
The radio receiving apparatus according to claim 4, wherein the plurality of decoding units perform error correction decoding again on a signal transmitted on each subcarrier obtained as a result of maximum likelihood estimation again. The reference signal generation means includes
Encoding means for encoding a signal obtained as a result of decoding by the plurality of decoding means;
A number of modulation means equal to the spatial multiplexing number;
Means for converting and rearranging the signals encoded by the encoding means into signals having a number of sequences equal to the spatial multiplexing number, in order to input the signals to the modulation means;
The wireless receiver according to claim 13, comprising: An OFDM (Orthogonal Frequency Division Multiplexing) signal including the plurality of subcarriers is spatially multiplexed and transmitted.
The reference signal generation means includes
A number of modulation means equal to the spatial multiplexing number;
Means for converting and rearranging the signal obtained as a result of decoding by the plurality of decoding means into a signal having a number of sequences equal to the spatial multiplexing number;
The wireless receiver according to claim 13, comprising:   The signal transmitted on each subcarrier is obtained as a result of maximum likelihood estimation again by the first and second maximum likelihood estimation means, wherein the signal transmitted on each subcarrier is obtained. Wireless receiver.   In the case where propagation path estimation is performed using a propagation path estimation known signal included in a signal transmitted by each subcarrier, the first calculation means, for each subcarrier, in the signal received by the subcarrier When the propagation path matrix of the signal transmitted by the subcarrier is obtained and the propagation path is estimated using the reference signal generated by the reference signal generation means, each subcarrier is received by the subcarrier. In the signal, a propagation path matrix of a signal transmitted on the subcarrier and a signal transmitted on another subcarrier at a position symmetrical to the subcarrier on the frequency axis around the center frequency is obtained. The wireless reception device according to any one of claims 9, 10, and 13. An OFDM (Orthogonal Frequency Division Multiplexing) signal including the plurality of subcarriers is spatially multiplexed and transmitted.
The receiving antenna group having a number greater than the number of spatial multiplexing, further comprising selection means for selecting a plurality of receiving antennas for receiving the OFDM signal from the receiving antenna group. 2. The wireless receiving device according to 2.   The radio reception according to claim 1, further comprising decoding means for performing error correction decoding on the signals transmitted by the subcarriers obtained by the first and second maximum likelihood estimation means. apparatus. Receiving an OFDM (Orthogonal Frequency Division Multiplexing) signal including a plurality of spatially multiplexed subcarriers;
For each of two subcarriers that are symmetrical on the frequency axis around the center frequency of the plurality of subcarriers of the received OFDM signal, the signals received on each of the two subcarriers are combined and combined. A second step for determining a signal;
A third step of extracting, from each of the combined signals, a signal component transmitted by a subcarrier in a frequency band higher than the center frequency, out of the two subcarriers;
A fourth step of extracting, from the combined signal, a signal component transmitted by a subcarrier in a frequency band lower than the center frequency out of the two subcarriers;
A fifth step of performing maximum likelihood estimation of the signal transmitted on the subcarrier from the signal component transmitted on the subcarrier in the high frequency band extracted from the combined signal;
A seventh step of performing maximum likelihood estimation of a signal transmitted on the subcarrier from a signal component transmitted on the subcarrier in the low frequency band extracted from the combined signal;
A wireless reception method including: Receiving an OFDM (Orthogonal Frequency Division Multiplexing) signal including a plurality of spatially multiplexed subcarriers;
For each of two subcarriers that are symmetrical on the frequency axis around the center frequency of the plurality of subcarriers of the received OFDM signal, the signals received on each of the two subcarriers are combined and combined. A second step for determining a signal;
For each subcarrier, the signal received by the subcarrier in the signal received by the subcarrier and the other subcarrier that is symmetrical to the subcarrier on the frequency axis centered on the center frequency are transmitted. A third step for obtaining a propagation path matrix of the received signal;
Using the propagation path matrix obtained for each subcarrier, a first signal component for extracting a signal component transmitted by a subcarrier in a higher frequency band than the center frequency from the two subcarriers from each combined signal. A fourth step of obtaining a matrix and a second matrix for extracting a signal component transmitted by a subcarrier in a frequency band lower than the center frequency;
A fifth step of multiplying the combined signal by the first matrix and extracting a signal component transmitted on the high frequency band subcarrier from the combined signal;
A sixth step of multiplying the combined signal by the second matrix and extracting a signal component transmitted on the low frequency band subcarrier from the combined signal;
A first inverse matrix that is an inverse matrix of an error correlation matrix between a signal component transmitted from the subcarrier in the high frequency band extracted from the combined signal and a signal transmitted from the subcarrier; The inverse matrix of the correlation matrix of the error between the signal component transmitted from the sub-carrier in the low frequency band extracted from the combined signal and the signal transmitted from the sub-carrier is calculated. A seventh step of calculating an inverse matrix of 2 using the second matrix;
An eighth step of using the first inverse matrix to estimate the maximum likelihood of the signal transmitted on the subcarrier from the signal component transmitted on the high frequency band subcarrier extracted from the combined signal; ,
A ninth step of performing maximum likelihood estimation of a signal transmitted on the subcarrier from a signal component transmitted on the subcarrier in the low frequency band extracted from the combined signal, using the second inverse matrix; ,
A wireless reception method including: A tenth step of generating a reference signal used for propagation path estimation based on a signal received by each subcarrier obtained as a result of maximum likelihood estimation;
An eleventh step of recalculating the propagation path matrix for each carrier using the reference signal;
Using the propagation path matrix recalculated for each subcarrier, the first matrix for extracting the signal component transmitted on the high frequency band subcarrier from each combined signal, and the low frequency band subcarrier A twelfth step of recalculating the second matrix for extracting signal components transmitted on a carrier wave;
A thirteenth step of multiplying the combined signal by the recalculated first matrix and again extracting a signal component transmitted on the high frequency band subcarrier from the combined signal;
A fourteenth step of multiplying the combined signal by the recalculated second matrix and re-extracting the signal component transmitted on the low frequency band subcarrier from the combined signal;
A fifteenth step of recalculating the first inverse matrix using the recalculated first matrix and recalculating the second inverse matrix using the recalculated second matrix; When,
A sixteenth step of re-maximizing the likelihood of the signal transmitted on the high frequency band subcarrier using the recalculated first inverse matrix;
A seventeenth step of using a recalculated second inverse matrix to estimate a maximum likelihood of a signal transmitted on the subcarrier in the low frequency band again;
The radio reception method according to claim 21, further comprising: repeating the tenth to seventeenth steps a plurality of times.   The radio reception method according to claim 21, further comprising a tenth step of performing error correction decoding on the signals transmitted on the subcarriers obtained in the eighth and ninth steps. An eleventh step of generating a reference signal used for propagation path estimation based on a signal obtained as a result of decoding;
A twelfth step of recalculating the channel matrix for each carrier using the reference signal;
Using the propagation path matrix recalculated for each subcarrier, the first matrix for extracting the signal component transmitted on the high frequency band subcarrier from each combined signal, and the subband of the low frequency band A thirteenth step of recalculating the second matrix for extracting signal components transmitted on a carrier wave;
A fourteenth step of multiplying the combined signal by the recalculated first matrix and re-extracting a signal component transmitted on the high frequency band subcarrier from the combined signal;
A fifteenth step of multiplying the combined signal by the recalculated second matrix and re-extracting the signal component transmitted on the low frequency band subcarrier from the combined signal;
A sixteenth step of recalculating the first inverse matrix using the recalculated first matrix and recalculating the second inverse matrix using the recalculated second matrix; When,
A seventeenth step of reestimating a maximum likelihood of a signal transmitted on the subcarrier in the high frequency band using the recalculated first inverse matrix;
An eighteenth step of re-maximizing the likelihood of the signal transmitted on the low frequency subcarrier using the recalculated second inverse matrix;
A nineteenth step of performing error correction decoding on the signal transmitted on each subcarrier obtained again by maximum likelihood estimation;
24. The wireless reception method according to claim 23, further comprising: repeating the eleventh to nineteenth steps a plurality of times.

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