JP4229210B2 - Light emitting element driving apparatus and light emitting element driving system - Google Patents

Description

  The present invention relates to a light emitting element driving apparatus and a light emitting element driving system for driving a light emitting element that emits light by passing a direct current, and particularly to driving a light emitting element having a large internal resistance (series resistance) typified by a surface emitting laser element. The present invention relates to a light emitting element driving device and a light emitting element driving system suitable for use.

  In the field of laser xerography using laser light as a light source, there is an increasing demand for higher resolution and higher speed. There is a limit to the speed (hereinafter referred to as modulation speed) at which on / off control of the driving of the laser element is performed according to input image data. In the case where the number of laser beams is one, if the resolution in the sub-scanning direction is increased as well as the resolution in the main scanning direction, the modulation speed must be sacrificed. Therefore, the only way to increase the resolution in the sub-scanning direction without increasing the modulation speed is to increase the number of laser beams. For example, when the number of laser light beams is four, the resolution in the main scanning and sub-scanning directions can be doubled assuming that the modulation speed is the same as in the case of one.

  By the way, the semiconductor laser has an edge emitting laser element (hereinafter simply referred to as an edge emitting laser) having a structure in which laser light is extracted in a direction parallel to the active layer, and a structure in which the laser light is extracted in a direction perpendicular to the active layer. The surface emitting laser element (hereinafter simply referred to as a surface emitting laser) is roughly classified. Conventionally, in laser xerography, an edge emitting laser is generally used as a laser light source.

  However, from the viewpoint of increasing the number of laser light beams, edge-emitting lasers are technically difficult, and structurally, surface-emitting lasers increase the number of laser light beams more than edge-emitting lasers. It is advantageous. For these reasons, in recent years, in the field of laser xerography, in order to meet the demand for higher resolution and higher speed, an apparatus using a surface emitting laser capable of emitting a large number of laser light beams as a laser light source. Development is underway.

  Here, a laser driving device conventionally used for laser xerography will be described. Conventionally, laser driving devices are roughly classified into three types: voltage driving type, current output voltage driving type, and current driving type. Hereinafter, each type of laser driving device will be described.

  First, as a voltage drive type laser drive device, one having a configuration in which a voltage applied to a laser element is directly controlled on the drive circuit side is known (for example, see Patent Document 1). Since this laser driving device can adjust the amount of light by directly controlling the power supply voltage of the logic gate, it can be constructed at a very low cost.

  Next, as a current output voltage driving type laser driving device, a current source and a laser element are connected in series, and a driving voltage is generated in the vicinity of the laser element using a termination resistor connected in parallel to the laser element. The thing of a structure is known (for example, refer patent document 2). In the case of this laser drive device, the output is a current flowing from a current source, and it is easier to generate an arbitrary current than to generate many voltages with low output impedance.

  Finally, as a current-driven laser driving device, a configuration is known in which a current generated by a constant current circuit is ON / OFF controlled by a current switch and supplied to a laser element (see, for example, Patent Document 3). ). Conventionally, this current-driven laser driving device has been generally used for driving an edge-emitting laser. The reason is as follows.

  As shown in FIG. 26, in the edge-emitting laser, the driving current increases exponentially with respect to the applied voltage. Therefore, when controlling by the voltage, the differential resistance (ΔV / ΔI) varies depending on the bias point, Since a non-linear element enters the negative feedback loop for control, control becomes difficult. On the other hand, when driven by current, the negative feedback loop is composed of linear elements because the amount of light is proportional to the current above the laser oscillation threshold current, which facilitates control. Further, even when a large number of laser elements have to be driven, a current source can be provided for each laser element relatively easily with current driving.

JP-A-11-68198 JP 59-18964 JP 57-13790 A

  Here, an electrical difference in driving between a surface emitting laser that can be used for laser xerography and a conventional edge emitting laser will be described. The difference is that, as shown in FIG. 26, the current increases exponentially up to about 100 mA with respect to the voltage applied to the laser element in the conventional edge emitting laser, whereas the surface emitting laser has several hundred μA. The voltage-current characteristic has a linear relationship with a small current.

  The reason is as follows. That is, when a surface emitting laser is used for laser xerography, it is necessary to emit light in a single mode in order to prevent the laser light from diffusing. For this purpose, the light emitting region must be narrowed down. By constricting the light emitting region, the junction area is reduced, which is caused by the increase in the resistance value of the internal resistance in the equivalent circuit of the surface emitting laser shown in FIG. For this reason, the voltage-current characteristic enters the linear region only by passing a small current.

  On the other hand, in the case of the edge emitting laser, when the current is increased, the voltage-current characteristic finally becomes a straight line due to the internal resistance. However, the current value entering the linear region differs by an order of magnitude or more compared to the surface emitting laser. That is, in the surface emitting laser, the resistance value of the internal resistance is several tens of ohms in the equivalent circuit shown in FIG. 28, whereas in the surface emitting laser, the resistance value of the internal resistance is several hundreds ohms, which is one digit or more larger. It has become.

  Further, in the case of a surface emitting laser, in order to meet the demand for higher resolution and higher speed in laser xerography, if a large number of light emitting units emitting a large number of laser beams are provided, a large number of light emitting units must be driven. Therefore, the drive device tends to be large. For this reason, as shown in FIG. 29, the wiring distance of the routing wiring becomes long. As is clear from FIG. 29, a large number of routing wires are arranged in parallel, so that parasitic capacitance increases and crosstalk due to line-to-line capacitance and common impedance is likely to occur.

  From the viewpoint of the modulation speed, in the case of the edge emitting laser, the resistance value of the internal resistance is small (see FIG. 28), and as shown in FIG. 30, the routing wiring is short and the parasitic capacitance is small. As a result, since the time constant τ determined by the resistance value R of the internal resistance and the capacitance value C of the parasitic capacitance is small, the rising and falling edges of the drive current waveform become steep as shown in FIG. On the other hand, in the case of the surface emitting laser, as described above, the resistance value of the internal resistance is large (see FIG. 27), the wiring length is long, and the capacitance including the parasitic capacitance with the adjacent wiring is large. Becomes larger. Therefore, as shown in FIG. 31, the rising and falling edges of the drive current waveform are very slow.

  In the current-driven laser driving apparatus according to the above-described conventional example, the edge-emitting laser starts up in about 1 nsec. In contrast, in the case of a surface emitting laser, the time constant is several tens of times that of an edge emitting laser, and the modulation speed is about several tens of MHz. This means that although the number of emitted laser beams is large, the overall modulation speed does not increase. Therefore, unless this modulation rate is significantly improved, there is no merit of using a surface emitting laser as the laser light source in laser xerography.

  From the above viewpoint, in order to drive a surface emitting laser capable of emitting a large number of laser light beams, a voltage drive type drive device is more advantageous than a current drive type drive device. That is, assuming that the current drive type drive device has an ideal current source and the voltage drive type drive device has an ideal voltage source, the stray capacitance at the drive end of the light emitting element having the internal resistance Ri as the drive target. In parallel with C, it is considered to have infinite and zero resistance values Ro, respectively. Therefore, the resistance R of the time constant CR that determines the rising and falling speed can be regarded as a combined resistance of Ro and Ri, with the former being the internal resistance of the light emitting element and the latter being the resistance value on the driving device side. Become. In addition, in the case of a current output voltage drive type driving device, there is a method in which the current output is passed through a resistor connected in parallel with the laser element and the laser element is driven by the voltage drop, but in order to increase the modulation speed, the resistance of the parallel resistor The value must be lowered, and the current consumption increases greatly.

  Here again, the voltage-driven laser driving device described in Patent Document 1 will be considered. As shown in FIG. 33, the voltage-driven laser driving apparatus according to this conventional example uses a CMOS logic gate 101 and switches between two potentials of a ground level and a power supply voltage, and connects to a laser element 103 via a resistor 102. The laser element 103 receives the back light output from the laser element 103 by the photodiode 104 and directly controls the power supply voltage of the logic gate 101 via the feedback circuit 105 based on the amount of the received light. The light quantity control is automatically performed so as to emit light with a desired light quantity. Further, the feedback circuit 105 is provided with a voltage source 106 for light quantity control.

  However, in the voltage-driven laser driving device according to the conventional example having the above-described configuration, the controllability is ensured by providing the resistor 102 between the logic gate 101 and the laser element 103 and substantially driving the current. As a result, when the surface-emitting laser is driven by the voltage-driven laser driving apparatus according to this conventional example, the resistance 102 interposed between the surface-emitting laser and the surface-emitting laser causes the modulation speed to be suppressed. This will hinder speeding up.

  In addition, when the surface emitting laser to be driven is applied to laser xerography, automatic light quantity control must be performed for each of a large number of light emitting units. Therefore, a voltage is applied to each logic gate 101 for driving. It is necessary to provide the source 106 separately. Furthermore, since the power supply of the logic gate is usually common, it is not possible to individually control the laser (light emitting unit) with one IC including a plurality of gates.

  At this time, as the performance required for the voltage source 106, the output impedance must be low. For this purpose, it is necessary to take measures such as providing a decoupling capacitor for power supply output in the IC chip and generally increasing the bias current in order to lower the output impedance of the power supply circuit. However, when measures such as providing a decoupling capacitor or increasing a bias current are taken, there are significant restrictions in designing an IC chip from the viewpoint of mounting and power consumption.

  The present invention has been made in view of the above-mentioned problems, and the object of the present invention is to increase the power consumption and to make a light emitting element such as a surface emitting laser by voltage-current driving without increasing the power consumption. It is an object of the present invention to provide a light emitting element driving apparatus and a light emitting element driving system that can be driven to increase the modulation speed.

The light emitting element driving device according to claim 1 is a voltage driving means for voltage driving the light emitting element, a current driving means for current driving the light emitting element, a voltage driving by the voltage driving means based on pulse data , and a current by the current driving means. It has a configuration comprising switching means for switching between driving. In the light emitting element driving device having such a configuration, the voltage driving by the voltage driving unit and the current driving by the current driving unit are switched based on the input data, thereby effectively combining the advantages of the voltage driving and the current driving. it can. Therefore, it is possible to realize drive control closer to ideal. The light emitting element to be driven is preferably a light emitting element having a relatively large internal resistance, such as a surface emitting laser or an EL (electro luminescence) element.

In addition, the voltage driving means includes a bias voltage applying means for applying a bias voltage to the light emitting element, and the switching means switches to voltage driving by the voltage driving means during the rising period of the pulse data, and thereafter to current driving by the current driving means. It is configured to switch to voltage driving by the bias voltage applying means during the switching and then the falling period of the pulse data . Thus, for example, the voltage data is driven by the voltage driving means in the rising period of the pulse data, and the current driving means is switched to the current driving by the current driving means after the rising, so that the light emitting element can be driven instantaneously in response to the rising edge of the pulse data, It is possible to prevent the light amount fluctuation of the light emitting element that occurs when voltage driving is continued after the rise of the pulse data. Here, as at least one period, voltage driving is performed during the rising (positive logic) or falling (negative logic) period of the pulse data corresponding to the timing of turning on (turning on) the light emitting element. preferable. In addition to this, voltage drive may be performed during a fall (positive logic) or rise (negative logic) period of pulse data corresponding to the timing of turning on (lighting) to turning off (extinguishing) of the light emitting element.

The light emitting element driving device according to claim 2 is the light emitting element driving device according to claim 1, wherein the switching means can simultaneously select voltage driving by the voltage driving means and current driving by the current driving means. When switching to voltage driving, the current driving by the current driving means is selected at the same time, and the current by the current driving is also supplied to the light emitting element. In the light emitting element driving apparatus having such a configuration, when the output current of the voltage driving means changes due to a change in load, the terminal voltage of the light emitting element changes at the moment of switching by the switching means. Therefore, by simultaneously selecting the current driving by the current driving means during the voltage driving by the voltage driving means and supplying the current by the current driving as a compensation current to the light emitting element, fluctuations in the terminal voltage of the light emitting element can be suppressed.

According to a third aspect of the present invention, in the light emitting element driving device according to the first aspect, the voltage driving period by the voltage driving means is equal to or less than the minimum pulse width of the pulse data. In the light emitting element driving device having such a configuration, when the voltage driving period satisfies the above-described conditions, the light emitting element can be driven in response to the rise (or fall) of pulse data having any pulse width. It can be done reliably.

The light emitting element driving device according to claim 4 is the light emitting element driving device according to claim 1, further comprising correction means for correcting a temperature variation of the light emitting element based on a terminal voltage of the light emitting element. In the light emitting element driving device having such a configuration, when a constant current flows through the light emitting element, the terminal voltage varies depending on the temperature of the light emitting element. Therefore, by detecting the terminal voltage, the temperature of the light emitting element is detected. Can be monitored more quickly and more accurately, so temperature compensation can be ensured.

The light-emitting element drive system according to claim 5 is provided with a plurality of light-emitting element drive devices according to claim 1 corresponding to the plurality of light-emitting elements, detection means for detecting light amounts of the plurality of light-emitting elements, and the detection means. And an error amplifying means for comparing the voltage corresponding to the detection result and the reference voltage and amplifying the error, and each of the light emitting element driving devices drives the light emitting element based on the output of the error amplifying means; It has become. In the light emitting element driving system having such a configuration, the light amount of the light emitting element is detected and fed back to each of the light emitting element driving devices, whereby each light amount of the plurality of light emitting elements can be controlled to be always constant.

  According to the light emitting element driving device of the first aspect, the voltage driving by the voltage driving means and the current driving by the current driving means are switched based on the input data, so that the advantage by the voltage driving and the advantage by the current driving are effective. Therefore, it is possible to realize drive control closer to ideal.

In addition, bias voltage applying means for applying a bias voltage to the light emitting element is provided, and when the light emitting element is turned off, the bias voltage is applied to the light emitting element in advance by the bias voltage applying means, so that the voltage is applied when the light emitting element is driven. The amplitude of the drive voltage applied from the source to the light emitting element can be reduced. In addition, switching to voltage driving by the voltage driving means during the rise period of the pulse data , switching to current driving by the current driving means , and then switching to voltage driving by the bias voltage applying means during the falling period of the pulse data The light emitting element can be driven instantaneously in response to the rising edge of the data, and the light quantity fluctuation of the light emitting element can be prevented as in the case where the voltage driving is continued after the rising edge of the pulse data.

According to the light emitting element driving apparatus of claim 2, when the output current of the voltage driving means changes due to a change in load, the terminal voltage of the light emitting element changes at the moment of switching by the switching means. Sometimes the current driving by the current driving means is selected at the same time, and the current driven by the current driving is supplied to the light emitting element as a compensation current, so that the output current of the voltage driving means can change the terminal voltage of the light emitting element due to the load fluctuation. Can be suppressed.

According to the light emitting element driving device of claim 3, by setting the voltage driving period by the voltage driving means to be equal to or less than the minimum pulse width of the pulse data, the rising edge (or the pulse data of any pulse width) (or The light emitting element can be reliably driven in response to the falling).

According to the light emitting element driving device of claim 4 , when a constant current flows through the light emitting element, the terminal voltage varies depending on the temperature of the light emitting element. Since the temperature of the light emitting element can be monitored more quickly and more accurately, temperature compensation can be reliably performed.

According to the light emitting element driving system of claim 5 , the light amount of the light emitting element is detected and fed back to each of the light emitting element driving devices so that the light amounts of the plurality of light emitting elements are always constant. Therefore, a light emitting element driving system capable of more accurate driving control can be realized.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[Basic concept]
FIG. 1 is a block diagram showing a basic concept of a light emitting element driving apparatus according to the present invention. Here, for example, a case where a surface emitting laser 11 having a large number (n) of light emitting portions LD1 to LDn each emitting laser light is used as a light emitting element to be driven is shown as an example. In FIG. 1, n light emitting units LD1 to LDn of the surface emitting laser 11 are connected to output terminals c of the switches SW1 to SWn, for example, with each cathode grounded and each anode serving as a driving end.

  Each of the input terminals a of the switches SW1 to SWn is commonly supplied from the first voltage source 12 with a bias voltage Vbias that puts the light emitting portions LD1 to LDn in a forward bias state and is lower than the laser oscillation threshold voltage. Each of the other input terminals b of the switches SW1 to SWn has the light emitting units LD1 to LDn in a forward bias state and drive voltages (control voltages) V1 to Vn that are equal to or higher than the laser oscillation threshold voltage are supplied to a second voltage source (variable voltage). Source) 13 individually.

  The second voltage source 13 has operational amplifiers OP1 to OPn having the driving voltages V1 to Vn as non-inverting (+) inputs, respectively, and is configured to feed back the output potential as its inverting (−) input. As a basic idea, the operational amplifiers OP1 to OPn may be omitted, and the drive voltages V1 to Vn may be directly applied to the other input terminals b of the switches SW1 to SWn.

  The switches SW1 to SWn switch the bias voltage Vbias and the driving voltages V1 to Vn as appropriate and apply them to the light emitting units LD1 to LDn of the surface emitting laser 11, respectively. Specifically, the switches SW1 to SWn are switched to the input terminal a before applying the driving voltages V1 to Vn to the light emitting portions LD1 to LDn, and the bias voltage is lower than the laser oscillation threshold voltage. Vbias is applied in advance to each of the light emitting portions LD1 to LDn, and then the drive voltage V1 to Vn equal to or higher than the laser oscillation threshold voltage is applied to each of the light emitting portions LD1 to LDn. As the switches SW1 to SWn, for example, CMOS transfer gates are used.

  In this way, the drive voltages V1 to Vn selected by the switches SW1 to SWn are directly applied to the drive ends (anodes) of the light emitting units LD1 to LDn of the surface emitting laser 11 from the output terminals c of the switches SW1 to SWn. By driving each of the light emitting units LD1 to LDn, there is no resistance component that causes the modulation speed to be suppressed between each output terminal of the switches SW1 to SWn and each driving end of the light emitting units LD1 to LDn. Modulation speed can be realized.

  Further, the bias voltage Vbias and the driving voltages V1 to Vn are appropriately switched by the switches SW1 to SWn and applied to the respective light emitting portions LD1 to LDn of the surface emitting laser 11. Specifically, when the laser is turned off, the forward bias is applied to the laser. By applying a voltage lower than the oscillation threshold voltage in advance, the amplitude of the applied voltage at the time of modulation can be kept small, so that the mode can be quickly shifted to the modulation mode.

  Moreover, the drive voltages V1 to Vn can be individually controlled for each of the light emitting units LD1 to LDn to uniformize the laser light amount, and if the voltage sources 12 and 13 are ideal, the modulation speed to the switching speed of the switches SW1 to SWn. Can be raised. If a sub-micron MOS transfer gate is used as the switches SW1 to SWn, the switching time (speed) can be set to 1 nsec or less. As a result, it is possible to realize a surface emitting laser driving device capable of achieving both individual light quantity control and high speed modulation for each of the light emitting portions LD1 to LDn of the surface emitting laser 11.

  Here, it is assumed that the voltage sources 12 and 13 are ideal. However, the ideal voltage source here is a voltage source whose output impedance is several tens of Ω or less regardless of the frequency.

  By the way, although the light emitting element driving device according to the present invention is of a voltage driving type, driving a circuit element directly with a voltage is a well-known technique (see, for example, JP-A-57-76884). Here, assuming a voltage drive as shown in this well-known technique, a switch SW is added thereto, and as shown in FIG. 2, a bias voltage Vbias is applied to one input terminal a of the switch SW, and the other is applied. Consider a case in which a surface emitting laser (light emitting unit) LD is driven by applying a driving voltage Vdrive through an emitter follower transistor Q to an input terminal b.

  Here, the emitter follower is a bipolar transistor, but this is the same even when a source follower using a FET (field effect transistor) or a constant voltage element such as an avalanche diode or a Zener diode is used. Such an element is equivalently represented by a voltage source Vp and an internal resistance Rp as shown in FIG. Further, it is usual to improve the drive pulse rising characteristics by inserting a decoupling capacitor C so that the voltage does not fluctuate at the moment when the switch SWd is switched.

  In the equivalent circuit shown in FIG. 3, when a current equivalent to that of the surface emitting laser LD is supplied to the transistor Q of the emitter follower, the switch SWd is switched from Vb side (bias voltage side) → Vp side (drive voltage side) → Vb. When switched to the side, as shown in the waveform diagram of FIG. 4, the B point potential at the moment of switching to the Vb side decreases with a time constant Rp × C, and finally the switch SWd due to the current to the surface emitting laser LD The value becomes lower by the voltage drop at the internal resistance Rsw1 and the internal resistance Rp.

  Thus, the laser drive voltage (potential at point B) fluctuates after the surface emitting laser LD is turned on when the switch SWd is connected to the Vb side and the voltage source Vp is in an unloaded state. This is because the voltage drop due to the internal resistance Rp of Vp is eliminated and the decoupling capacitor C is charged by the open terminal voltage of the voltage source Vp. That is, if there is an internal resistance Rp of the voltage source Vp, overshoot occurs at the moment when the surface emitting laser LD is turned on.

  Further, in the case of an emitter follower (or source follower), if high speed driving is to be realized, a current larger than the current flowing to the surface emitting laser LD must be supplied to the transistor Q of the emitter follower. Power consumption will increase. In particular, when driving a surface emitting laser having a large number of light emitting portions, a large current flows through each of the emitter follower transistors Q prepared for the number of light emitting portions, and the power consumption of the entire driving device is also reduced. Since it becomes extremely large, it is difficult to make the drive device into an IC.

  On the other hand, in the technique based on this basic concept showing the circuit configuration for one channel in FIG. 5, the potential at point A is fed back as the voltage source Vp in the equivalent circuit shown in FIG. The operational amplifier OP is made so small that it can be ignored. Further, since the effect of reducing the output impedance due to negative feedback decreases with frequency, a decoupling capacitor C is connected for the compensation.

  In this way, when a voltage source that can ignore the output impedance (= internal resistance) over the entire frequency band is connected, the potential at point B is controlled to be constant regardless of the presence or absence of the load current. As shown in FIG. 7, fluctuations in the drive voltage due to the ON / OFF of the switch SWd can be prevented.

  Further, since the decoupling capacitor C is connected to the voltage source output terminal whose voltage is stabilized by feedback, the output potential does not change depending on the presence or absence of a load, so that overshooting at the time of laser lighting can be prevented. By setting the capacitance value of the decoupling capacitor C to be larger than the capacitance value of the parasitic capacitance at the driving end of the surface emitting laser LD viewed from the switch SWd, it is possible to have an effect of suppressing voltage fluctuation.

  The rise time constant τ of the drive pulse (voltage pulse) when the laser is turned on is the resistance from the voltage source output terminal to the drive end of the surface emitting laser LD, that is, the internal resistance Rsw1 of the switch SWd in the equivalent circuit shown in FIG. And the internal resistance Rld of the surface emitting laser LD is determined by the product of the resistance value of the combined parallel resistance and the capacitance value of the parasitic capacitance Cs caused by the switch IC and the wiring. Therefore, at least the resistance value from the voltage source output terminal to the driving end of the surface emitting laser LD is set to be smaller than the differential resistance value (several hundreds Ω) of the surface emitting laser LD, so that the rise time constant of the driving pulse when the laser is turned on. τ can be reduced (shortened).

  Here, the capacitance value of the parasitic capacitance Cs is about several tens of pF by adding the capacitance value of the parasitic capacitance in the printed circuit board wiring to the input / output capacitance of the IC. Although it is necessary, if a recent CMOS transistor is used as a switch, several tens of Ω can be easily realized as an on-resistance, so that there is no obstacle to the modulation speed.

  Further, in the technology based on this basic concept, since the current flowing through the operational amplifier OP constituting the second voltage source 13 is set to be smaller than the current flowing through the surface emitting laser LD, the light emitting unit is provided. Even when high-speed driving of a large number of surface-emitting lasers is realized, the power consumption of each operational amplifier OP can be kept low, and the overall power consumption of the driving device can be reduced.

  Here, the current flowing through the operational amplifier OP is an average current flowing through the transistors constituting the final stage of the operational amplifier OP. Further, in the second voltage source 13, by adopting a circuit configuration in which negative feedback is applied to the operational amplifier OP, an average current flowing through a transistor constituting the final stage of the operational amplifier OP is set smaller than a current flowing through the surface emitting laser LD. Like to do.

[First Embodiment]
FIG. 8 is a circuit diagram showing a configuration example of a driving system using the light emitting element driving apparatus according to the first embodiment of the present invention, for example, a surface emitting laser driving apparatus. In the present embodiment, for example, a surface emitting laser 21 having 36 light emitting portions LD1 to LD36 is used as a light emitting element to be driven.

  8, the surface emitting laser driving system according to the present embodiment includes a first voltage source 22 that generates a bias voltage Vbias, a second voltage source (variable voltage source) 23 that generates a driving voltage (control voltage), and The first voltage source 22 and the second voltage source 23 are the first voltage source 12 and the second voltage source shown in FIG. 13 respectively.

  The bias voltage Vbias generated by the first voltage source 22 determines the voltage applied to the light emitting unit LD when the light emitting unit (laser) LD is turned off, and the light emitting unit LD does not emit light in order to increase the modulation speed. The voltage is set as high as possible. Normally, the voltage is set slightly lower than the lowest laser oscillation threshold voltage among the plurality of light emitting units.

  The second voltage source 23 includes an amplifier 231, a variable resistor VR, a buffer 232, a switch SW 1, and a drive control circuit 233 for 36 channels (ch) provided corresponding to the light emitting units LD 1 to LD 36 of the surface emitting laser 21. −1 to 233-36, switches SWfb1 to SWfb36, and capacitors Cfb1 to Cfb36.

  The amplifier 231 uses the reference voltage Vref set corresponding to the target laser power as a non-inverted (+) input, and uses the detection signal supplied from the light amount detection circuit 24 via the switch SW1 as an inverted (−) input. Yes. The switches SWfb1 to SWfb36 and the capacitors Cfb1 to Cfb36 are provided in correspondence with the light emitting portions LD1 to LD36 of the surface emitting laser 21, respectively. The amplifier 231 is connected in series between the inverting input terminal and the output terminal.

  The variable resistor VR is connected between the output terminal of the amplifier 231 and the bias line L to which the bias voltage Vbias is applied. A capacitor C1 is connected between the bias line L and the ground. The output voltage of the amplifier 231 is supplied as a control voltage Vcont to the drive control circuits 233-1 to 233-36 through the variable resistor VR and the buffer 232. The drive control circuits 233-1 to 233-36 are commonly supplied with the bias voltage Vbias via the bias line L.

  In this example, the bias voltage Vbias is commonly applied to all the drive control circuits 233-1 to 233-36 for 36 channels. However, the bias voltage Vbias is applied between the 36 light emitting units LD1 to LD36 of the surface emitting laser 21. When there is a large variation in the laser oscillation threshold voltage, a plurality of bias voltages Vbias having different voltage values are prepared, and the light emitting units LD1 to LD36 are individually grouped or those having similar laser oscillation threshold voltages are grouped for each group. It is also possible to provide a bias voltage Vbias having a voltage value close to optimum. According to this, it is possible to cope with variations in the laser oscillation threshold voltage without complicating the circuit configuration as compared with the case where the bias voltage Vbias is provided for 36 channels.

  The drive control circuits 233-1 to 233-36 for 36 channels all have the same circuit configuration. Therefore, the specific circuit configuration will be described with reference to FIG. 9 showing an enlarged circuit configuration of the drive control circuit 233 for one channel. The drive control circuits 233-1 to 233-36 include the switches SW 1 to SWn (n = 36) in FIG.

  As is apparent from FIG. 9, the drive control circuit 233 includes an amplifier 235, a resistor R1, three capacitors Csh, Cp, Cld, eight switches SWsh, SWp, SWn, SWshp, SWs, SWd, SWc, SWe and The configuration has three current sources I1, I2, and I3.

  A control voltage Vcont supplied from the second voltage source 23 is applied to one end of the resistor R1. The switch SWsh has an input terminal connected to the other end of the resistor R1 and an output terminal connected to the inverting input (−) terminal of the amplifier 235. The capacitor Csh is connected between the inverting input terminal of the amplifier 235 and the ground. The non-inverting (+) input terminal of the amplifier 235 is connected to the node N1. Here, the switch SWsh is connected to the inverting input terminal side of the amplifier 235. However, in order to avoid switching noise generated in the switch SWsh and leakage due to the switch SWsh and the resistor R1, It is also possible to change the form.

  Current source I1 and switch SWp are connected in series between power supply Vcc and node N1. The current source I 1 has its current Ip controlled by the output voltage of the amplifier 235. The switch SWn and the current source I2 are connected in series between the node N1 and the ground. The switches SWp and SWn are normally closed switches.

  The current source I3 and the switch SWs are connected in series between the power supply Vcc and the node N2 to which the anode of the light emitting part LD of the surface emitting laser 21 is connected. The switch SWshp has one terminal connected to the output of the amplifier 235 and the other terminal connected to the current source 3. The capacitor Cp is connected between the power supply Vcc and the other terminal of the switch SWshp. The current sources I1 to I3 are configured by a current mirror circuit, for example.

  The switch SWc and the capacitor (decoupling capacitor) Cld are connected in series between the node N2 and the ground. The switch SWd has one input terminal b connected to the node N1, its output terminal c connected to the node N2, and the other input terminal a supplied with the bias voltage Vbias from the first voltage source 21. The switch SWe is connected between the node N1 and a connection point between the capacitor Cld and the switch SWc.

  In FIG. 8 again, the light quantity detection circuit 24 uses, for example, a photodiode PD as a photodetector for detecting the laser light emitted from the light emitting portions LD1 to LD36 of the surface emitting laser 21. The cathode of this photodiode PD is connected to the power supply Vcc. One end of a resistor R is connected to the anode of the photodiode PD. The other end of the resistor R is grounded. A non-inverting input terminal of the amplifier 241 is connected to a connection point between the anode of the photodiode PD and one end of the resistor R. The reference voltage Vref described above is applied to the inverting input terminal via the switch SW2. The amplifier 241 has a configuration in which its inverting input terminal and output terminal are connected.

  In the light amount detection circuit 24 having such a configuration, the photodiode PD detects the laser light emitted from the light emitting portions LD1 to LD36 of the surface emitting laser 21, and outputs a detection signal corresponding to the light amount. The detection signal of the light amount detection circuit 24, that is, the output signal of the amplifier 241 is supplied to the inverting input terminal of the amplifier 231 in the second voltage source 23 described above via the switch SW1. That is, by returning the detection signal of the light amount detection circuit 24 to the drive control circuits 233-1 to 233-36 via the second voltage source 23, each laser power of the light emitting units LD1 to LD36 of the surface emitting laser 21 is obtained. A feedback system is configured to perform automatic light quantity control (hereinafter referred to as APC: auto power control) for controlling the power so that the power is regulated by the reference voltage Vref.

  Next, the circuit operation of the surface emitting laser driving circuit according to this embodiment having the above-described configuration will be described with reference to FIGS. The timing chart is shown in FIG. In the timing chart of FIG. 10, the APC is sequentially executed for the 36 light emitting units LD1 to LD36 of the surface emitting laser 21 in one APC mode, and this is repeated four times, and then the mode is shifted to the modulation mode. Is taken as an example.

  In the timing chart of FIG. 10, the switch SWfb and the switch SW1 of the second voltage source 23, the switches SWsh, SWp, SWn, SWshp, SWs, SWd, SWc, and SWe of the drive control circuit 233, and the light amount detection circuit 24. Each switching pulse for controlling ON / OFF of the switch SW2 is positive logic, and the same reference numeral as that of each switch is given for easy identification.

  In the drive control circuit 233, the switch SWd connected to the terminal a side is turned off and the state connected to the terminal b side is turned on. The switch SWc is simultaneously turned on so that the anode potential quickly becomes the specified potential when the light emitting part LD of the surface emitting laser 21 is turned on. Further, when the light emitting unit LD is turned off, the switch SWe is turned on to charge the decoupling capacitor Cld with the anode voltage when the light emitting unit LD is turned on.

  First, after turning on the power (Power ON), at time T0-, the switch SWfb1, the switch SW1, the switch SWsh, the switch SWs, the switch SWshp, and the switch SWd are turned on, and the switch SWp, the switch SWn, the switch SWc, and the switch SWe are turned off. . At this time, the current Is of the current source I3 flows to the light emitting unit LD1 via the switch SWs. As a result, the light emitting unit LD1 is turned on.

  When the light emitting unit LD1 is turned on, the laser light is received by the photodiode PD of the light amount detection circuit 24, and a current corresponding to the light amount flows through the photodiode PD. The current flowing through the photodiode PD is converted into a voltage by the resistor R, amplified by the amplifier 241 and output as a detection voltage corresponding to the laser power of the light emitting unit LD1.

  This detection voltage is supplied to the second reference voltage source 23 and becomes an inverting input of an amplifier (error amplifier) 231 via the switch SW1. The amplifier 231 amplifies and outputs the difference (error voltage) between the detected voltage and the reference voltage Vref. The output voltage of the amplifier 231 is obtained as a voltage obtained by dividing the difference voltage from the bias voltage Vbias by the variable resistor VR, and is supplied to the ch1 drive control circuit 233-1 via the buffer 232.

  In the drive control circuit 233-1, that is, the drive control circuit 233 shown in FIG. 9, the control voltage Vcont input via the buffer 232 is supplied to the amplifier (op-amp) 235 via the resistor R1 and the switch SWsh. The amplifier 235 controls the laser power of the light emitting unit LD1 by controlling the current Ip of the current source I1 according to the input voltage. By this feedback control, the detection voltage of the light amount detection circuit 24 eventually converges with the reference voltage Vref. The series of controls described above is APC (automatic light quantity control).

  Thereafter, when the switch SWfb1, the switch SWshp, and the switch SWsh are turned OFF, the respective control voltages at that time are held in the capacitor Cfb1, the capacitor Cp, and the capacitor Csh that are connected in series. At this time, the voltages held in the capacitor Cfb1, the capacitor Cp, and the capacitor Csh are the output voltage of the amplifier 231 at ch1, the control voltage for setting the drive current for the light emitting unit LD1, and the terminal voltage of the light emitting unit LD1 at that time.

  By repeating the above operation continuously for the number of light emitting portions LD of the surface emitting laser 21 (36 in this example), all the control voltages of the drive control circuits 233-1 to 233-36 for 36 channels are obtained. These are held in 36 capacitors Cfb1 to Cfb36 connected between the inverting input terminal and the output terminal of the amplifier 231. When the 36-channel APC is completed, the switch SW1 is turned off and the switch SWfb1 is turned on to prepare the control voltage at ch1 as the output voltage of the amplifier 231 for the next APC.

  Until the next APC, the switch SW2 of the light amount detection circuit 24 is turned on so that the output voltage of the amplifier 241 becomes the reference voltage Vref during the modulation period. Thereby, the time required for the detection output of the photodiode PD to be in a steady state at the start of the next APC can be shortened. As a result, when APC is started next, each node is subjected to negative feedback control from the final voltage at the time of the previous light amount control, so it is not always necessary to converge to the final voltage by one control. This is particularly important in laser xerography using a polygon mirror in an optical scanning system that scans laser light in the main scanning direction, and intermittent exposure control prevents unnecessary exposure of the photoconductor. Can be prevented.

  In the second voltage source 23, the variable resistor VR having one end connected to the output terminal of the amplifier 231 is provided for adjusting the gain of the negative feedback, and can achieve both stability and accuracy of the negative feedback loop. Thus, the resistance ratio, that is, the voltage division ratio of the differential voltage between the output voltage of the amplifier 231 and the bias voltage Vbias is set.

  The other end of the variable resistor VR is connected to the bias line L of the bias voltage Vbias, which is a measure for the terminal voltage of the light emitting unit LD being controlled to be equal to or higher than the laser oscillation threshold voltage. In this way, even if the gain of the negative feedback loop is reduced, the voltage supplied to the buffer 232 is lower than the laser oscillation threshold voltage, thereby preventing control from becoming impossible. Here, the bias voltage Vbias is applied to the other end of the variable resistor VR. However, a separate power supply may be provided and a predetermined power supply voltage may be applied from the power supply. It can also be controlled.

  In the drive control circuit 233 shown in FIG. 9, the capacitor SW (decoupling capacitor) Cld is turned on, and the switch SWc is turned on at the moment when the current flows to the light emitting portion LD, so that the terminal voltage of the light emitting portion LD is quickly increased. The driving voltage is set to be as follows. However, since the capacitance value of the capacitor Cld is limited, the terminal voltage decreases and the amount of laser light also decreases with the capacitor Cld alone. In order to compensate for this, the negative feedback amplifier 235 is connected via the switch SWd. Therefore, the capacitance value of the capacitor Cld is determined from the response speed of the amplifier 235.

In general, a CMOS operational amplifier requires about 1 μsec to respond, and when a CMOS operational amplifier is used as the amplifier 235, the degree of drop in the terminal voltage of the capacitor Cld within 1 μsec is set within an allowable variation. Specifically, if the driving current of the surface emitting laser 21 is 1 mA, the voltage fluctuation is 1 / C × 1 mA × 1 μsec = 1 / C × 10 ⁻⁹ .

  Assuming that the allowable light amount variation is 2%, the internal resistance of the surface emitting laser 21 is 500Ω, and the voltage variation with respect to the light amount allowable variation is 10 mV, 0.1 μF is required as the capacitance C of the capacitor Cld. However, this value is too large when the entire driving apparatus is to be accommodated in a one-chip IC. Even if such a capacitor can be realized, for example, by connecting to the outside of the IC chip, the output potential of the amplifier 235, that is, the internal control potential changes due to load fluctuations, so that light is emitted at the moment when the switch SWd is turned on. The terminal voltage of the part LD varies. Then, the voltage source output becomes unstable although it is 2% until the negative feedback converges.

  As a countermeasure, the switch SWs is turned on in synchronization with the switch SWd, and the laser drive current, that is, the current Is of the current source I3 is supplied as a compensation current to the drive end (anode) of the light emitting unit LD. In this way, since the output current fluctuation from the amplifier 235, that is, the output current fluctuation of the voltage source is suppressed to be small regardless of the state of the switch SWd, the transient voltage of the voltage source due to the load fluctuation when the switch SWd is turned on. Variations can be prevented. Further, in this way, the time for which the capacitor Cld maintains the terminal voltage of the light emitting unit LD is the time until the current source I3 starts to flow current to the driving end of the light emitting unit LD.

  When the current source I3 is configured by the MOS transistor M10 of FIG. 11, the response is much faster than the response of the operational amplifier, and the burden on the capacitor Cld is reduced accordingly. As a result, the capacitance value of the capacitor Cld can be reduced. Further, since the laser drive current is supplied from the current source I3, the voltage fluctuation due to the ON resistance of the switch SWd or the switch SWc can be reduced to a negligible level. Here, the necessary capacitance of the capacitor Cld is about 100 to 1000 times the parasitic capacitance because it is sufficient to charge the parasitic capacitance connected to the laser terminal.

  In the timing chart of FIG. 10, for example, when time T1 is taken as an example, two times T1- and T1 + whose phases are slightly shifted back and forth with respect to time T1 are shown. T1 + represents a timing for an operation that the S / H (sample hold) circuit or the like that is not desired to overlap with the operation at the time T1, which is a standard operation timing.

  FIG. 11 is a circuit diagram showing a specific circuit configuration of the drive control circuit 233 shown in FIG. 9. The amplifier 235 and the switches SWsh, SWp, SWn, SWshp, SWs, SWd, SWc, and SWe are replaced with MOS transistors. This shows the case where it is realized. Here, the switch SWsh, the resistor R1, and the capacitor Csh in the drive control circuit 233 shown in FIG. 9 are omitted. In FIG. 11, the same parts as those in FIG. 9 are denoted by the same reference numerals.

  In FIG. 11, an amplifier 235 includes NchMOS transistors M1 and M2 whose sources are connected to each other to form a differential pair, an NchMOS transistor M3 connected between the source common connection point and the ground, and a differential pair transistor M1. , M2 and PchMOS transistors M4 and M5 connected between the drains of the M2 and the power source Vcc.

  In the amplifier 235 having the above configuration, the control voltage Vcont via the resistor R1 and the switch SWsh is applied to the gate of the MOS transistor M1, and the potential of the node N1 is applied to the gate of the MOS transistor M2. The MOS transistor M3 forms a constant current circuit with its gate biased by a constant voltage Vg. The MOS transistors M4 and M5 are active loads, and each gate is connected in common and the transistor M5 forms a current mirror circuit by connecting the gate and drain in common.

  PchMOS transistors M6 and M7 are connected in series between the power supply Vcc and the node N1. The MOS transistor M6 constitutes a current source I1, and the drain output of the MOS transistor M1 in the amplifier 235 is given to the gate thereof. The MOS transistor M7 constitutes a switch SWp, and a switching pulse SWpnX obtained by inverting the switching pulse SWpn shown in FIG. 10 is applied to the gate of the MOS transistor M7.

  NchMOS transistors M8 and M9 are connected in series between the node N1 and the ground. The MOS transistor M8 constitutes a switch SWn, and a switching pulse SWpn shown in FIG. 10 is applied to the gate thereof. The MOS transistor M9 forms a current source I2, and a constant bias voltage Vg is applied to its gate. These transistors M6 to M9 are also operational amplifiers, and constitute a two-stage operational amplifier together with the previous stage amplifier 235.

  Further, PchMOS transistors M10 and M11 are connected in series between the power supply Vcc and the node N1. The MOS transistor M10 constitutes a current source I3. The MOS transistor M11 constitutes a switch SWs, and a switching pulse SWsX obtained by inverting the switching pulse SWs shown in FIG. A capacitor Cp is connected between the gate of the MOS transistor M10 and the power supply Vcc.

  A PMOS transistor M12 and an NMOS transistor M13 are connected in parallel between the gates of the MOS transistors M6 and M10. These MOS transistors M12 and M13 constitute a transfer gate. A switching pulse SWshpX obtained by inverting the switching pulse SWshp shown in FIG. 10 is applied to the gate of the MOS transistor M12, and the switching pulse shown in FIG. 10 is applied to the gate of the MOS transistor M13. SWshp is applied.

  Here, the control voltage applied to the gate of the MOS transistor M6 is also applied to the gate of the MOS transistor M10 via the CMOS transfer gate (M12, M13). During APC, the MOS transistor M10 causes the surface emitting laser 21 to The light emitting unit LD is driven. At this time, the control voltage is held in the capacitor Cp. The MOS transistors M10 and M11 are preferably composed of dual gate MOS transistors. As a result, the source-drain parasitic capacitance at the connection between the transistors M10 and M11 can be minimized, so that the rising speed (response speed) of the current source I3 including the transistor M10 can be increased.

  A Pch MOS transistor M14 is connected between the node N1 and the node N2, and a Pch transistor M15 is connected between the Vbias power supply and the node N2. The MOS transistors M14 and M15 constitute a switch SWd. A switching pulse SWdX obtained by inverting the switching pulse SWd shown in FIG. 10 is applied to the gate of the MOS transistor M14, and a switching pulse SWd is applied to the gate of the MOS transistor M15. .

  Further, PchMOS transistors M16 and M17 are connected in series between the node N1 and the node N2. The MOS transistor M16 constitutes a switch SWe, and a switching pulse SWeX obtained by inverting the switching pulse SWe shown in FIG. The MOS transistor M17 constitutes a switch SWc, and a switching pulse SWcX obtained by inverting the switching pulse SWc shown in FIG. 10 is applied to its gate.

  A capacitor Cld is connected between the drain of the MOS transistor M16 (source of the MOS transistor M17) and the ground. Here, since the current source I3 is overwhelmingly faster than the voltage source by the operational amplifier, the contribution of suppressing the voltage fluctuation by the capacitor Cld is reduced, so that the capacitance value of the capacitor Cld can be reduced. The Vbias power supply corresponds to the first voltage source 22 shown in FIG. 8 for generating the bias voltage Vbias.

  A resistor R0 and a capacitor Cc are connected in series between the drain of the MOS transistor M2 of the amplifier 235 and the gate (node N1) of the MOS transistor M2. The resistor R0 depends on the configuration of the operational amplifier, that is, the amplifier 235, but is provided to prevent oscillation due to a capacitive load. The capacitor Cc is a phase compensation capacitor.

  What is necessary here is that the potential of the node N1 is negatively fed back to the input side of the operational amplifier, and the impedance of the node N1 is so small that it can be ignored. The existing technology for devising the configuration of the operational amplifier and connecting the capacitive load without oscillation is included in this.

  In the surface emitting laser driving apparatus according to the present embodiment, the anode of the light emitting part LD of the surface emitting laser 21 is used as a driving end, and driving is performed by flowing a driving current from the anode. For this reason, in the drive control circuit 233 having the above-described configuration, it is necessary to adopt a configuration (so-called source) that sweeps out current with respect to the anode of the light emitting section LD as a current source. Therefore, the current source of the amplifier 235 is an NchMOS transistor. In M3, the active load is composed of PchMOS transistors M4 and M5, respectively.

  However, the circuit configuration is not limited to this. The MOS transistor constituting the amplifier 235 is of a reverse conductivity type, and the current direction is reversed using a current mirror circuit, whereby a current is supplied to the anode of the light emitting unit LD. It is also possible to configure a current source of the sweep type. FIG. 12 shows a circuit configuration of a drive control circuit 233 ′ according to the modification.

  In the amplifier 235 of the drive control circuit 233 according to the previous configuration example, the differential amplifier is composed of the Nch MOS transistors M1 and M2, the constant current circuit is composed of the Nch MOS transistor M3, and the current mirror circuit is composed of the Pch MOS transistors M4 and M5. On the other hand, in the amplifier 235 ′ of the drive control circuit 233 ′ according to this modification, the differential amplifier is composed of PchMOS transistors M21 and M22, the constant current circuit is composed of PchMOS transistor M23, and the current mirror circuit is composed of NchMOS transistors M24 and M25. Have

  The gates of NchMOS transistors M26 are further connected to the gates of the MOS transistors M24 and M25. The MOS transistor M26 forms a current mirror circuit together with the MOS transistors M24 and M25. This current mirror circuit takes out the direction of the current flowing through the amplifier 235 'by inverting it. This current is converted into a voltage by a diode-connected PchMOS transistor M27 in which a gate and a drain are connected, and is supplied as a control voltage to the gate of the PchMOS transistor M10.

  As described above, in the surface emitting laser driving device according to the first embodiment, attention is paid to the fact that the surface emitting laser emitting in the single mode has a large internal resistance (one or more orders of magnitude larger than the edge emitting laser), Since the driving end (in this example, the anode) of the surface emitting laser is substantially directly driven by a voltage, there is no resistance component between the driving end to suppress the modulation speed. The modulation speed can be increased.

  For example, in the field of laser xerography, a surface emitting laser that emits a number of laser light beams is used as the laser light source, and the surface emitting laser is differentiated by using the driving device according to this embodiment for driving the surface emitting laser. Driving with a voltage source having an output impedance smaller than the resistance value (several hundreds Ω) can greatly contribute to higher resolution and higher speed.

  As an example, FIG. 13 shows the dependency of the printing speed on the voltage source output impedance when assuming 2400 dpi, 256 gradations and 36 beams. As can be seen from the figure, as the voltage source output impedance becomes smaller than several hundred Ω, the printing speed can be further increased, and an extremely high printing speed can be obtained at about 10 Ω.

  By the way, when voltage driving by applying voltage to the surface emitting laser is continued, as shown in the waveform diagram of FIG. 14, the current flowing through the surface emitting laser increases as time passes after the voltage is applied, Since the amount of light of the surface emitting laser depends on the flowing current, the amount of light increases with time. That is, after the voltage is applied, as the temperature of the laser element rises, the drive current is increased by continuing to apply the same voltage to the place where the terminal voltage that gives the same light amount decreases, and as a result, the laser element The amount of light also increases. For example, in the field of laser xerography, this variation in the amount of light causes a reduction in the image quality of the formed image. In view of this point, the voltage driving and the current driving are used together in a light emitting element driving apparatus according to a second embodiment of the present invention described below.

[Second Embodiment]
FIG. 15 is a circuit diagram showing a configuration example of a driving system using a light emitting element driving apparatus according to the second embodiment of the present invention, for example, a surface emitting laser driving apparatus. Also in this embodiment, for example, a surface emitting laser 31 having 36 light emitting portions LD1 to LD36 is used as a light emitting element to be driven.

  In FIG. 15, the surface emitting laser driving system according to the present embodiment includes drive control circuits 32-1 to 32-36 for 36 channels provided corresponding to the light emitting units LD1 to LD36 of the surface emitting laser 31, A light amount detection circuit 33 for detecting the light amount of the surface emitting laser 31, an error amplifying circuit 34 constituting a feedback system for feeding back the detection result of the light amount detection circuit 33 to the drive control circuits 32-1 to 32-36, and drive control; The control circuit 35 controls the circuits 32-1 to 32-36.

  The drive control circuits 32-1 to 32-36 for 36 channels all have the same circuit configuration. That is, the drive control circuit 32 (32-1 to 32-36) has voltage drive means at the front stage and current drive means at the rear stage, as shown in principle in FIG. In the figure, as a typical example of voltage driving means, there is a negative feedback loop from the output terminal to the inverting input terminal, and the input side holding the control voltage at the time of light amount control (APC) on the input side of the non-inverting input terminal A buffer amplifier having a capacitor as holding means and further having a capacitor as output side holding means for holding the output voltage on the output side is shown.

  As the current driving means, a constant current source to which a control voltage is applied corresponding to the output of the voltage driving means is shown. In this constant current source, the supply current is controlled according to the control voltage. For example, when the constant current source is constituted by an FET, this is realized by controlling the gate voltage of the FET.

  In FIG. 15, in principle, the voltage drive by the voltage drive means is performed based on the input data by the switch means provided between the respective outputs of the voltage drive means and the current drive means and the laser element LD as the light emitting element. The current driving by the current driving means is switched.

  Next, a specific circuit configuration of the drive control circuit 32 for one channel will be described using the circuit diagram of FIG. 16 which is substantially the same as FIG.

  As is apparent from FIG. 16, the drive control circuit 32 includes a voltage driving circuit 41 that drives the light emitting unit LD of the surface emitting laser 31 with a voltage, a current driving circuit 42 that drives the light emitting unit LD of the surface emitting laser 31 with current, A switch circuit 43 that switches between voltage driving by the voltage driving circuit 41 and current driving by the current driving circuit 42 is provided. Here, the switch circuit 43, together with a control circuit 35 (see FIG. 15) for controlling ON / OFF of the switch circuit 43, performs voltage driving by the voltage driving circuit 41 and current driving by the current driving circuit 42 based on input data. Switching means for switching between the two is configured.

  The voltage drive circuit 41 selectively takes in a voltage corresponding to the voltage at the time of light quantity control (APC), that is, an output voltage of the error amplifier circuit 34 (see FIG. 15), and a voltage taken in by the switch 411 412, an operational amplifier 413 using the holding voltage of the capacitor 412 as a non-inverting (+) input, a buffer 414 for buffering the output voltage of the operational amplifier 413, and an output voltage of the buffer 414 selectively. The switch 415 for outputting, a capacitor 416 for holding the output voltage of the switch 415, and a bias voltage source 417 for generating the bias voltage Vbias are provided.

  The voltage drive circuit 41 is configured such that the output voltage of the voltage drive circuit 41, that is, the voltage held in the capacitor 416 is negatively fed back to the operational amplifier 413 as its inverted (−) input via the switch circuit 43. Yes. Note that the output voltage of the voltage drive circuit 41 is set to a voltage value equal to or higher than the laser oscillation threshold voltage with the light emitting portion LD of the surface emitting laser 31 in the forward bias state. The bias voltage Vbias is set to a voltage value that places the light emitting part LD of the surface emitting laser 31 in a forward bias state and is lower than the laser oscillation threshold voltage.

  The current drive circuit 42 includes an inverter 421 that inverts the output voltage of the operational amplifier 413, a switch 422 that selectively receives the output voltage of the inverter 421, a capacitor 423 that holds the voltage captured by the switch 422, and the capacitor 423. And a constant current source 424 that outputs a current corresponding to the voltage held in the circuit.

  The switch circuit 43 includes a switch 431 connected between the output terminal of the voltage driving circuit 41, that is, the output terminal of the capacitor 416, and the node N2, and the output terminal of the current driving circuit 423, that is, the output terminal of the constant current source 424. The switch 432 connected between the node N2, the switch 433 connected between the node N2 and the driving end of the light emitting unit LD of the surface emitting laser 31, the driving end of the light emitting unit LD, and the bias voltage source 417 The switch 434 is connected to the output terminal. Here, the voltage held in the capacitor 416 of the voltage driving circuit 41 is negatively fed back to the operational amplifier 413 via the switch 431.

  In the drive control circuit 32 configured as described above, the switches 411 and 415 of the voltage drive circuit 41, the switch 422 of the current drive circuit 42, and the switches 431 to 434 of the switch circuit are input as pulse data by the control circuit 35 shown in FIG. ON / OFF control is performed based on the data.

  The drive control circuit 32 basically has the same configuration as the drive control circuit 233 (see FIG. 9) in the first embodiment. 16 and FIG. 9, the voltage driving circuit 41 in FIG. 16 corresponds to the amplifier 235 and its peripheral circuit in FIG. 9, and the current driving circuit 42 in FIG. 16 is the current source I3 in FIG. Corresponding to the circuit, the switch circuit 43 of FIG. 16 corresponds to the switches SWc, SWd, and SWs of FIG.

  In FIG. 16, a constant current source 424 symbolizes a PMOS transistor, and the control voltage corresponds to the gate voltage of the PMOS transistor. For this reason, when the control voltage is increased, the current decreases, and when it is decreased, the current increases. As described above, since the change in the current with respect to the control voltage is opposite to that of the current source using the NMOS transistor, the control voltage is inverted by an inverter (inverting amplifier) 421 in FIG.

  In FIG. 9, since the current source 11 is controlled without being inverted by the output of the amplifier 235, when the voltage of the node N1 is fed back to the amplifier 235, it is input to the non-inverting input side. In FIG. 16, since the inverter 421 is inserted, the potential of the node N2 is fed back to the inverting input side of the error amplifier 413. However, in FIG. 9 and FIG. 16, negative feedback is applied. The two circuits are identical.

  In FIG. 15 again, the error amplifying circuit 34 uses the switch 341 that is turned on during light quantity control (APC) and the reference voltage Vref set corresponding to the target laser power as a non-inverted (+) input. An error amplifier 342 having a detection signal supplied from the light amount detection circuit 33 via the switch 341 as an inverting (−) input, and a negative feedback loop for negatively feeding the output of the error amplifier 342 to the error amplifier 342 as its inverting input 343.

  The negative feedback loop 343 includes capacitors Cfb1 to Cfb36 that hold voltages corresponding to the output voltage of the error amplifier 342 during APC of the surface emitting laser 31, and switches SWfb1 to SWfb1 that are connected in series to the capacitors Cfb1 to Cfb36, respectively. SWfb 36 is provided corresponding to the number of drive control circuits 32-1 to 32-36.

  The light quantity detection circuit 33 uses, for example, a photodiode PD as a photodetector for detecting the laser light emitted from the light emitting portions LD1 to LD36 of the surface emitting laser 31. As this light quantity detection circuit 33, the thing of the completely same circuit structure as the light quantity detection circuit 24 in 1st Embodiment can be used.

  Next, regarding the circuit operation of the drive control circuit 32 in the surface emitting laser drive system according to the second embodiment having the above-described configuration, the operation description of FIGS. 18 to 21 corresponding to each operation mode based on the timing waveform diagram of FIG. This will be described with reference to the drawings. 17, (A) is input data (pulse data), (B) is a terminal voltage of the surface emitting laser 31, (C) is a temperature of the surface emitting laser 31, and (D) is a surface emitting laser 31. Each waveform of the amount of light is shown.

  Control of the circuit operation of the drive control circuit 32 is performed based on input data under the control of the control circuit 35. In the timing waveform diagram of FIG. 17, in the rising period of the pulse data (A) as the input data, the control circuit 35 turns on the switches 431, 432, 433 and switches 411, 415, 422 as shown in FIG. , 434 are turned off. As a result, the voltage held in the capacitor 416 is applied to the light emitting part LD of the surface emitting laser 31 via the switch 431 and the switch 433.

  As a result, voltage driving is performed by the voltage driving circuit 41 during the rising period of the pulse data (A). When driving a surface emitting laser capable of emitting a large number of laser beams, as described in the prior art, voltage driving is more advantageous than current driving, and in response to the rise of pulse data (A). The surface emitting laser 31 can be driven to emit light instantaneously. The voltage driving period by the voltage driving circuit 41 is set to be equal to or less than the minimum pulse width of the pulse data. As a result, laser driving can be reliably performed in response to the rise (or fall) of pulse data of any pulse width.

  In the rising period of the pulse data (A), switching to voltage driving by the voltage driving circuit 41 is performed, and at the same time, the current driving by the current driving circuit 42 is selected by turning on the switch 432. As a result, a current is output from the constant current source 424 in accordance with the voltage held in the capacitor 423, and this current is supplied to the light emitting unit LD of the surface emitting laser 31 via the switch 432 and the switch 433. The current of the constant current source 424 at this time flows to the light emitting part LD of the surface emitting laser 31 as a compensation current in the first embodiment.

  As described above, when the voltage is driven during the rising period of the pulse data (A), the current of the constant current source 424 is caused to flow as a compensation current into the light emitting part LD of the surface emitting laser 31, so that the output current fluctuation of the operational amplifier 413 is changed. Therefore, transient voltage fluctuation due to load fluctuation when the switch 433 is turned on can be prevented.

  When light emission driving of the surface emitting laser 31 is started, a current flows through the light emitting portion LD of the surface emitting laser 31, so that the surface emitting laser 31 generates heat and the temperature (C) of the surface emitting laser 31 elapses over time. As it rises.

  When the rising period of the pulse data (A) has passed, the control circuit 35 turns on the switches 432 and 433 and turns off the switches 411, 415, 422, 431, and 434, as shown in FIG. Thereby, the current output from the constant current source 424 according to the holding voltage of the capacitor 423 is supplied to the light emitting unit LD of the surface emitting laser 31 through the switch 432 and the switch 433. As a result, after the rising edge of the pulse data (A), current driving by the current driving circuit 42 is performed.

  By this current driving, a constant current output from the constant current source 424 flows in the light emitting part LD of the surface emitting laser 31. As a result, the terminal voltage (B) of the surface emitting laser 31 gradually decreases. At this time, although the temperature (C) of the surface emitting laser 31 rises with time, a constant current flows through the surface emitting laser 31, and the light intensity (D) of the surface emitting laser 31 is caused by this current. Therefore, it is possible to prevent fluctuations in the light amount (D) of the surface emitting laser 31 as in the case of continuing voltage driving after the rise of the pulse data (A).

  Next, in the falling period of the pulse data (A), as shown in FIG. 20, the control circuit 35 turns on the switches 415, 431, and 434 and turns off the switches 411, 422, 432, and 433. Thereby, the bias voltage Vbias generated from the bias voltage source 417 is applied to the light emitting part LD of the surface emitting laser 31 via the switch 434 (voltage driving).

  Thus, when the laser is extinguished, the bias voltage Vbias, which is forward bias and lower than the laser oscillation threshold voltage, is applied in advance, so that the amplitude of the applied voltage at the time of modulation can be kept small. Can move quickly. Further, by continuously applying the bias voltage Vbias even after the falling period of the pulse data (A), as described above, the applied voltage amplitude can be kept small at the next rising edge.

  The circuit operations in the voltage drive mode, current drive mode, and laser extinction mode in the modulation period are as described above. Before entering the modulation period, the circuit operation in the APC mode that automatically controls the light amount of the surface emitting laser 31 is performed. Done.

  In this APC mode, as shown in FIG. 21, the control circuit 35 turns on the switches 411, 422, 432, 433 and turns off the switches 415, 431, 434. At this time, the detection voltage output from the light amount detection circuit 33 according to the light amount of the surface emitting laser 31 is supplied to the error amplification circuit 34. At this time, in the error amplifying circuit 34, the switch SWfb1 of ch1 is in the ON state, and the input detection voltage is given to the error amplifier 342 through the switch 341 as its inverting input.

  Then, the error amplifier 342 amplifies and outputs the difference between the detection voltage of the light amount detection circuit 33 and the reference voltage Vref. The output voltage of the error amplifier 342 is supplied to the drive control circuit 32-1. This output voltage is held in the capacitor 412 via the switch 411 in the drive control circuit 32-1, that is, the drive control circuit 32 shown in FIG. Then, the operational amplifier 413 controls the current of the constant current source 424 via the inverter 421 and the switch 422 based on the holding voltage of the capacitor 412.

  As a result, the current supplied from the constant current source 424 to the light emitting unit LD1 of the surface emitting laser 31 via the switches 432 and 433 changes, and as a result, the laser power of the ch1 light emitting unit LD1 is controlled. By this feedback control, the detection voltage of the light amount detection circuit 33 eventually converges with the reference voltage Vref. The series of controls described above is APC.

  Thereafter, when the switch SWfb1 of the error amplifying circuit 34, the switch 422 of the current driving circuit 42, and the switch 411 of the voltage driving circuit 41 are turned OFF, the respective control voltages at that time are held in the capacitor Cfb1, the capacitor 423, and the capacitor 412. . At this time, the voltages held in the capacitor Cfb1, the capacitor 423, and the capacitor 412 are respectively the output voltage of the error amplifier 342 in ch1, the control voltage for setting the drive current for the light emitting unit LD1, and the terminal voltage of the light emitting unit LD1 at that time. Corresponding voltage.

  By repeating the above operation continuously for the number of light emitting portions LD (36 in this example) of the surface emitting laser 31, all the control voltages of the drive control circuits 32-1 to 32-36 for 36 channels are obtained. It is held in 36 capacitors Cfb1 to Cfb36 connected between the inverting input terminal and the output terminal of the error amplifier 342. When the 36-channel APC is completed, the error amplification circuit 34 turns off the switch 341 and turns on the switch SWfb1 to prepare the control voltage at ch1 as the output voltage of the error amplifier 342 for the next APC.

  As described above, in the surface emitting laser driving apparatus according to the second embodiment, the voltage driving circuit 41 that drives the surface emitting laser 31 by voltage and the current driving circuit 42 that drives the surface emitting laser 31 by current are provided, and the voltage By switching the voltage drive by the drive circuit 41 and the current drive by the current drive circuit 42 based on the input data, it is possible to effectively combine the advantages of the voltage drive and the advantages of the current drive. Control can be realized.

  In particular, voltage driving is performed by the voltage driving circuit 41 during the rising period of the pulse data that is input data, and switching to current driving by the current driving circuit 42 is performed after the rising, so that the surface is instantaneously responded to the rising of the pulse data. The light emitting laser 31 can be driven to emit light, and fluctuations in the light amount of the surface emitting laser 31 can be prevented as in the case where voltage driving is continued after the rise of pulse data.

  By the way, as described above, the surface emitting laser 31 generates heat when a current flows, and the temperature rises according to the current. As the temperature of the surface emitting laser 31 rises, as is apparent from the timing waveform diagram of FIG. 17, when the surface emitting laser 31 is driven next, the light amount deviation caused by the temperature variation ΔT in the light amount (D). Will occur. In normal drive control, such a light amount deviation is of a level that does not cause a problem. However, in order to realize better drive control, it is preferable that there is no light amount deviation of this temperature variation ΔT. A light emitting element driving apparatus according to a third embodiment of the present invention described below is provided with a function of correcting the light amount deviation of the temperature variation ΔT.

[Third Embodiment]
FIG. 22 is a circuit diagram showing a configuration example of a drive control circuit in a light emitting element driving apparatus according to the third embodiment of the present invention, for example, a surface emitting laser driving apparatus. In FIG. Is shown. The only difference from the surface emitting laser driving apparatus according to the second embodiment is the configuration of the drive control circuit 32. Therefore, when a drive system is constructed, the configuration is basically the same as in FIG.

  In FIG. 22, each of the voltage drive circuit 41 and the current drive circuit 42 includes a correction circuit that corrects the light amount deviation of the temperature variation ΔT. Specifically, the correction circuit of the voltage drive circuit 41 includes an error amplifier 418 and a sample hold circuit 419. The error amplifier 418 is supplied with the terminal voltage (detection voltage) of the light emitting part LD of the surface emitting laser 31 as its non-inverted (+) input.

  Here, when a constant current flows through the light emitting portion LD of the surface emitting laser 31, the terminal voltage varies according to the temperature of the light emitting portion LD. Specifically, the terminal voltage decreases as the temperature of the element increases. Therefore, the temperature of the light emitting part LD can be monitored by detecting the terminal voltage of the light emitting part LD of the surface emitting laser 31.

  The method of monitoring the temperature of the light emitting unit LD is not limited to the method of detecting the terminal voltage of the light emitting unit LD, and for example, a temperature detecting means such as a thermistor is disposed in the vicinity of the surface emitting laser 31 to detect the temperature. It is also possible to adopt a method using the detection output of the means. However, the method of detecting the terminal voltage of the light emitting unit LD has an advantage that the temperature of the light emitting unit LD can be monitored more quickly and more accurately.

  The sample hold circuit 419 includes a sampling switch SWsh and a hold capacitor Ch, samples the terminal voltage of the light emitting unit LD with the sampling switch SWsh, and holds the sampling voltage in the hold capacitor Ch. The voltage held in the hold capacitor Ch becomes a reference voltage and is given to the error amplifier 418 as its inverting (−) input. The output terminal of the error amplifier 418 is connected to the open terminal of the capacitor 412.

  In the correction circuit of the voltage driving circuit 41, the sampling switch SWsh is turned on, for example, at the timing before entering the APC period described above, thereby sampling the terminal voltage of the light emitting unit LD. Thus, by sampling at the timing before the APC period starts, it is possible to sample a stable terminal voltage before the temperature of the surface emitting laser 31 rises. This sampling voltage is used for the subsequent correction processing.

  That is, the error amplifier 418 takes in the terminal voltage of the light emitting part LD of the surface emitting laser 31 as a non-inverted input, and sequentially compares it with the hold voltage of the hold capacitor Ch, that is, the reference voltage, and the error amplified voltage is the open end of the capacitor 412. To give. As a result, the holding voltage of the capacitor 412 is shifted by the error amplification voltage. That is, this error amplification voltage is superimposed on the holding voltage of the capacitor 412 as a correction value. The corrected voltage is applied to the light emitting unit LD when the surface emitting laser 31 is driven to emit light.

  On the other hand, the correction circuit of the current drive circuit 42 has a bias current source 425 whose one end is connected to the power supply Vcc, a terminal voltage of the light emitting part LD of the surface emitting laser 31 as a non-inverting input, and a bias voltage Vbias of the bias voltage source 417. An error amplifier 426 serving as an inverting input, and a sample hold circuit 427 that includes a capacitor 428 and a switch 429 and samples and holds the error amplification voltage of the error amplifier 426. The hold held by the capacitor 428 of the sample hold circuit 427 The bias current Ibias of the bias current source 425 is controlled according to the voltage. As a result, the bias current source 425 outputs a bias current Ibias corresponding to the bias voltage Vbias.

  The switch circuit 43 further includes a switch 435 connected between the other end of the bias current source 425 and the node N2. Under the control of the control circuit 35 (see FIG. 15), the switch 435 is turned on when the current is driven after the falling period of the input data in the input data off period, that is, the laser extinction period. At this time, the switch 432 is turned off under the control of the control circuit 35.

  As a result, during the current drive after the falling period of the input data has elapsed in the laser extinction period, the bias current Ibias from the bias current source 425 is replaced by the surface emitting laser 31 instead of the drive current from the constant current source 424. Supplied to the light emitting unit LD.

  Next, the circuit operation of the drive control circuit 32 having the above-described configuration provided with the correction circuit for correcting the light amount deviation of the temperature variation ΔT will be described with reference to the timing waveform diagram of FIG. In the timing waveform diagram of FIG. 23, (A) is input data (pulse data), (B) is a terminal voltage of the surface emitting laser 31, (C) is a temperature of the surface emitting laser 31, and (D) is a surface emitting laser 31. Each waveform of the amount of light is shown.

  Note that, in the light emitting period of the surface emitting laser 31, voltage driving is performed during the rising period of the input data, and thereafter switching to current driving is the same as in the case of the drive control circuit according to the second embodiment, but the third embodiment. In the embodiment, voltage driving is performed even during the falling period of the input data (A), and then current driving is performed by flowing a current equal to or less than the laser emission threshold current. During this laser extinction period, the correction circuit of the voltage drive circuit 41 operates, and an operation for correcting the light amount deviation of the temperature variation ΔT is performed.

  Specifically, the error amplifier 418 compares the terminal voltage of the light emitting part LD of the surface emitting laser 31 with the hold voltage of the hold capacitor Ch, and gives the error amplification voltage to the open end of the capacitor 412. Then, this error amplification voltage is superimposed on the holding voltage of the capacitor 412 as a correction value, and is applied to the light emitting portion LD of the surface emitting laser 31 as a corrected voltage as shown in FIG. As a result, the light amount deviation of the temperature variation ΔT is corrected, and the light amount (D) becomes constant even if the temperature variation ΔT remains.

  When the extinction period of the laser element is sufficiently long, there is no problem in correcting the rising voltage of the next pulse data. However, when the temperature is to be controlled in a high temperature region in laser xerography, the period during which the laser element is turned off is shortened. In such a case, since it is not possible to shift to constant current driving during a short light extinction period, the temperature fluctuation of the laser element cannot be detected correctly, and the rising voltage of the next pulse data cannot be corrected correctly. When such an extinguishing period is very short, temperature fluctuations during extinguishing can be ignored, so that the voltage at startup can be corrected with the laser temperature at the time of lighting immediately before the laser element is extinguished.

  Since the laser element is driven with a constant current when the laser is turned on, it can be known by detecting the change in the terminal voltage of the laser element as in the case of turning off the laser. However, since the current flowing through the laser element is different between when the light is turned off and when the light is turned on, the terminal potential of the laser element is different. Therefore, when the light is turned off, the fluctuation of the laser terminal voltage when the light is turned off and when the light is turned on, the fluctuation of the laser terminal voltage when the light is turned on may be output alternately.

  Specifically, the laser terminal voltage at the time of lighting in automatic light quantity control (APC) and the laser terminal voltage at the time of extinguishing at the laser temperature at this time are detected and held, and the laser is based on these. By subtracting the held lighting voltage at the time of modulation from the terminal voltage, and subtracting the held off voltage at the time of light extinction, it is possible to detect the temperature variation of the laser element at the time of turning on and off. Alternatively, by alternately outputting the difference voltage between the laser terminal voltage and the held lighting voltage at the time of lighting, and the difference voltage between the laser terminal voltage and the held lighting voltage at the time of turning off the light. It is possible to detect temperature fluctuations of the laser element when the light is on and when the light is off.

  In FIG. 22, in principle, one error amplifier 418 having one sample hold circuit for holding the reference voltage on the input side is shown. However, in order to realize the former, for example, the error amplifier 418 is similar to FIG. In order to realize the latter, the sample hold circuit for holding the reference voltage on the input side in one system has two systems, both on and off. There are two error amplifiers that can selectively output, on and off, and the output is wired-ORed and connected to the open end of the capacitor 412, and the connection point with the capacitor 412 is grounded through a resistor. The structure which performs is considered.

  In this example, the voltage at the time of automatic light control (APC) is held and used as the reference voltage for subtraction. However, if the correction is zero, the reference may be taken by APC or otherwise. It doesn't matter if you take it.

The above is the operation of the drive control circuit according to the third embodiment during the modulation period. The following points are different from the drive control circuit according to the second embodiment. That is, during the laser extinction period, the drive control circuit according to the second embodiment continuously applies the bias voltage Vbias during the extinction period, whereas the drive control circuit according to the third embodiment temporarily applies the bias voltage Vbias. A bias voltage Vbias is applied, but the bias current source 425 shown in FIG. 22 is then used to flow the bias current into the current mode so that the temperature can be predicted from the change in the terminal voltage of the laser element.

  Further, the laser device is turned off following the APC mode in the drive control circuit according to the second embodiment, and in FIG. 22, the error amplifier 426 using the laser terminal voltage as a non-inverting input and the bias voltage Vbias as an inverting input. At the same time, the capacitor 428 connected to the power supply Vcc side of the sample hold circuit 427 is charged with the control voltage at that time, and the switch 427 is turned on when the output voltage of the error amplifier 426 converges. Set to OFF state. As a result, the bias current source 425 is set with a bias current at which the laser terminal voltage substantially matches the bias voltage Vbias.

  As the bias current source 425, a PMOS transistor is assumed in the same way as the constant current source 424. Therefore, when the voltage applied to the control terminal corresponding to the gate of the PMOS transistor is increased, the current of the bias current source 425 decreases. When it is lowered, it increases. Thus, when the laser is turned off during the modulation period, the laser terminal voltage becomes the bias voltage Vbias during the fall period of the pulse data, and then the current mode can be quickly shifted to the minimum with the fluctuation of the terminal voltage.

  As a result, it is possible to shift to a state in which the temperature of the laser element can be monitored within a short time after the light is turned off, and the voltage at the next lighting can be corrected. As shown as the modulation period in FIG. 24, this is important in laser xerography, especially when the correction is made effective when the laser element is lit and driven with a short pulse with a very short period during highlighting. It is.

  As described above, in the drive control circuit 32 in the surface emitting laser drive device according to the third embodiment, the voltage drive circuit 41 and the current drive circuit 42 each include a correction circuit that corrects the light amount deviation of the temperature variation ΔT. Even if the temperature variation ΔT remains as the temperature of the surface emitting laser 31 rises, the light amount deviation caused by the temperature variation ΔT can be reliably corrected. There is no deviation of ΔT. Hereinafter, the driving accompanied with the correction of the temperature fluctuation is referred to as dynamic driving.

  In FIG. 24, in the APC period, the laser extinction period, and the modulation period, the amount of light (A), the terminal voltage (B), and the temperature (C) of the surface emitting laser 31 are the case of dynamic driving indicated by solid lines and the constant voltage driving indicated by dashed lines. The waveform which compared with the case of is shown. FIG. 25 shows waveforms of the light amount (A), the terminal voltage (B), and the temperature (C) of the surface emitting laser 31 with the modulation period enlarged. As is apparent from these waveform diagrams, since the light amount deviation caused by the temperature fluctuation ΔT can be reliably corrected by performing the dynamic drive indicated by the solid line, the light amount (A ) Is constant.

  In the third embodiment, the method of correcting the rising voltage of the pulse data has been described as the target for detecting the laser temperature from the laser terminal voltage and correcting the temperature dependence of the laser element. However, the correction target is the laser characteristic. Among them, all characteristics having temperature dependency are targeted, and specifically, it can be applied to correct the temperature dependency of threshold current and light emission efficiency (ΔP: light quantity / ΔI: drive current). It is.

  In each of the above embodiments, the case where a surface emitting laser is used as a light emitting element to be driven has been described as an example. However, the present invention is not limited to the application to driving a surface emitting laser, and an internal element such as an EL element is used. The present invention can be similarly applied to high-speed driving of all light-emitting elements having high resistance.

It is a block diagram which shows the basic concept of the light emitting element drive device which concerns on this invention. It is a circuit diagram which shows the voltage drive circuit using the emitter follower which concerns on a well-known technique. FIG. 3 is an equivalent circuit diagram of FIG. 2. FIG. 3 is an operation waveform diagram of FIG. 2. It is a circuit diagram which shows the voltage drive circuit based on the basic concept of this invention. FIG. 6 is an equivalent circuit diagram of FIG. 5. FIG. 6 is an operation waveform diagram of FIG. 5. It is a circuit diagram showing an example of composition of a drive system using a surface emitting laser drive device concerning a 1st embodiment of the present invention. It is a circuit diagram which shows the circuit structure of the drive control circuit for 1ch. It is a timing chart for demonstrating the circuit operation | movement of the surface emitting laser drive device which concerns on 1st Embodiment of this invention. It is a circuit diagram which shows the specific circuit structural example of the drive control circuit for 1ch. It is a circuit diagram which shows the modification of the concrete circuit structure of the drive control circuit for 1ch. It is a figure which shows the voltage source output impedance dependence of the printing speed at the time of assuming 2400 dpi, 256 gradations, and 36 beams. It is a timing waveform diagram which shows the waveform of each part in the case of voltage drive. It is a circuit diagram which shows the structural example of the drive system using the surface emitting laser drive device which concerns on 2nd Embodiment of this invention. It is a circuit diagram which shows the circuit structure of the drive control circuit for 1ch. It is a timing waveform diagram for explaining the operation of the surface emitting laser driving apparatus according to the second embodiment. It is an equivalent circuit diagram at the time of voltage driving. It is an equivalent circuit diagram at the time of current driving. It is an equivalent circuit diagram at the time of laser extinction. It is an equivalent circuit diagram at the time of APC mode. It is a circuit diagram which shows the circuit structure of the drive control circuit in the surface emitting laser drive device which concerns on 3rd Embodiment of this invention. FIG. 12 is a timing waveform diagram for explaining the operation of the drive control circuit according to the third embodiment. FIG. 6 is a timing waveform diagram comparing dynamic driving and constant voltage driving for the light amount (A), terminal voltage (B), and temperature (C) of a surface emitting laser in an APC period, a laser extinction period, and a modulation period. It is a timing waveform diagram showing each waveform of the light quantity (A), terminal voltage (B), and temperature (C) of the surface emitting laser showing the modulation period enlarged. It is the figure which compared the VI characteristic of an edge surface emitting laser and a surface emitting laser. It is a figure which shows the equivalent circuit of a surface emitting laser. It is a figure which shows the equivalent circuit of an edge emitting laser. It is a figure which shows the wiring of a surface emitting laser drive circuit. It is a figure which shows the wiring of an edge-emitting laser drive circuit. It is a wave form diagram of a surface emitting laser. It is a wave form diagram of an edge emitting laser. It is a circuit diagram which shows the circuit structure of the voltage drive type laser drive circuit which concerns on a prior art example.

Explanation of symbols

  11, 21, 31... Surface emitting laser, 12, 22, first voltage source, 13, 23, second voltage source, 24, 33, light quantity detection circuit, 32, 32-1 to 32-36, 233 233-1 to 233-36 drive control circuit 34 error control circuit 35 control circuit 41 voltage drive circuit 42 current drive circuit 43 switch circuit 231 342 418 426 error amplifier , 235, 235 ′, 413, operational amplifiers

Claims (5)

A light-emitting element driving device that drives a light-emitting element that emits light by flowing a direct current according to pulse data,
Voltage driving means for driving the light emitting element with voltage;
Current driving means for current-driving the light emitting element;
Switching means for switching between voltage driving by the voltage driving means and current driving by the current driving means based on the input data;
The voltage driving means includes bias voltage applying means for applying a bias voltage to the light emitting element,
The switching means switches to voltage driving by the voltage driving means during the rising period of the pulse data, and then switches to current driving by the current driving means, and then by the bias voltage applying means during the falling period of the pulse data. A light emitting element driving device characterized by switching to voltage driving . The switching means can simultaneously select voltage driving by the voltage driving means and current driving by the current driving means. When switching to voltage driving by the voltage driving means, the current driving by the current driving means is also selected at the same time. The light-emitting element driving device according to claim 1, wherein a current generated by the current driving is also supplied to the light-emitting element.   2. The light emitting element driving device according to claim 1, wherein a period of voltage driving by the voltage driving means is less than a minimum pulse width of the pulse data. The light-emitting element driving device according to claim 1, further comprising a correcting unit that corrects a temperature variation of the light-emitting element based on a terminal voltage of the light-emitting element. The light emitting element driving device according to claim 1, wherein a plurality of light emitting element driving devices are provided corresponding to the plurality of light emitting elements
Detecting means for detecting light amounts of the plurality of light emitting elements;
An error amplifying means for comparing the voltage corresponding to the detection result of the detecting means and a reference voltage and amplifying the error, and
Each of said light emitting element drive devices drives the said light emitting element based on the output of the said error amplification means. The light emitting element drive system characterized by the above-mentioned.

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